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标    签: 50Ohm

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传输线理论,为什么PCB传输线控制50Ohm

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Harbor翻译 Q: Why do most engineers use 50Ω pc-board transmission lines (sometimes to the extent of this value becoming a default for pc-board layout)? Why not 60Ωor 70Ω?—Tim Canales 为什么很多工程师用 50 Ohm PCB 传输线,有些时候这则成为 PCB 布线的默认设置。 为什么不是 60 Ohm 或者 70 Ohm? A: Given a fixed trace width, three factors heavily influence pc-board trace impedance decisions. First, the near-field EMI from a pc-board trace is proportional to the height of the trace above the nearest reference plane; less height means less radiation. Second, crosstalk varies dramatically with trace height; cutting the height in half reduces crosstalk by a factor of almost four. Third, lower heights generate lower impedances, which are less susceptible to capacitive loading. A:在线宽固定的情况下,有三个主要因素影响PCB的阻抗。第一,到PCB传输线最近的电磁干扰层的 影响正比于PCB传输线到最近的参考平面的距离,越小的距离就越小的辐射。第二,串扰也随传输线 的厚度有则明显的变化,减少一半的传输线厚度将减少传输线串扰。第三,越小的距离产生越小的阻 抗,这个有利于减少容性负载的影响 All three factors reward designers who place their traces as close as possible to the nearest reference plane. What stops you from pressing the trace height all the way down to zero is the fact that most chips cannot comfortably drive impedances less than about 50V.(Exceptions to this rule include Rambus, which drives 27Ω, and the old National BTL family, which drives 17Ω). 所有的三个因素鼓励设计者去设计传输线更加靠近参考平面。阻止传输线厚度降为0的主要原因是大 部分的芯片不可能很好的驱动小于50 Ohm的传输线,除了Rambus 27Ω,旧的国家BTL联盟17Ω It is not always best to use 50Ω.For example, an old NMOS 8080 processor operating at 100 kHz doesn’t have EMI, crosstalk, or capacitive-loading problems, and it can’t drive 50Ωanyway. For this processor, because very high impedance lines minimize the operating power, you should use the thinnest, highest-impedance lines you can make. 并不是所有的用50Ω做阻抗控制都是最好的。如:旧NMOS 8080处理器工作频率100Khz的时 候没有电磁干扰和串扰以及容性负载的问题,同时在任何时候他也不可能驱动50Ω的传输线.就这个 处理器而言,由于高的阻抗将减小驱动电源,故我们必须使用我们能制造出来的最薄的,最高的阻抗 的传输线. Purely mechanical considerations also apply. For example, in dense, multilayer boards with highly compressed interlayer spaces, the tiny lithography that 70Ωtraces require becomes difficult to fabricate. In such cases, you might have to go with 50Ωtraces, which permit a wider trace width, to get a manufacturability board. 同时我们也要考虑机械问题(制程问题)。如,在高密度多层板的高压合夹层空间下,70Ω的传输线 在现在微小印刷技术下很难被制造出来,在这种情况下,你也许会转而用50Ω的传输线,50Ω的传输 线允许用比70Ω更宽的线宽,从而使电路板存在可制造性。 What about coaxial-cable impedances? In the RF world, the considerations are unlike the pc-board problem, yet the RF industry has converged on a similar range of impedances for coaxial cables. According to IEC publication 78 (1967), 75Ω is a popular coaxial impedance standard because you can easily match it to Harbor翻译 several popular antenna configurations. It also defines a solid polyethylene-based 50Ω cable because, given a fixed outer-shield diameter and a fixed dielectric constant of about 2.2 (the value for solid polyethylene), 50Ω minimizes the skin-effect losses. 那同轴电缆线的阻抗又如何呢?在射频的世界里,考量的问题点与印刷电路板又有不同,至今射频工业 还是关注与相似阻抗的同轴电缆线,根据国际电工协会出版物78(1967), 75Ω是一个非常容易被接受的 同轴阻抗标准,因为你能更加容易的匹配几种比较流行的天线结构。它同时定义了50Ω的实心聚乙烯 线的结构,即给出了固定的外层屏蔽层的直径,同时给出了固定的介电常数2.2。因为在50Ω先可以减 小传输过程中的趋肤效应。 You can prove the optimality of 50Ω coaxial cable from basic physics. The skin-effect loss, L, (in decibels per unit length) of the cable is proportional to the total skin-effect resistance, R, (per unit length) divided by the characteristic impedance, Z0, of the cable. The total skin-effect resistance, R, is the sum of the shield resistance and center conductor resistances. The series skin-effect resistance of the coaxial shield, at high frequencies, varies inversely with its diameter d2. The series skin-effect resistance of the coaxial inner conductor, at high frequencies, varies inversely with its diameter d1. The total series resistance, R, therefore varies proportionally to (1/d2+1/d1). Combining these facts and given fixed values of d2 and the relative electric permittivity of the dielectric insulation, ER, you can minimize the skin effect loss, L, starting with the following equation: 同时你可以从物理学定理上证明50Ω同轴电缆线阻抗的优越性。电缆趋肤效应损失为L(每单位长度 上)正比于趋肤效应电阻R(每单位长度)除以电缆的特性阻抗Z。电缆总的趋肤电阻为外层屏蔽层 电阻加上内层传输线电阻之和。在高频下,屏蔽层的串接趋肤电阻反比与它的直径d2。内层传输线的 串联趋肤电阻反比与它的直径d1.则总的串接电阻正比于(1/d2+1/d1).结合上述因素,相互介电常数ER,屏 蔽层直径d2 为给定的固定值,你能下面这个公式来最小化趋肤效应的损失, In any elementary textbook on electromagnetic fields and waves, you can find the following formula for Z0 as a function of d2, d1, and ER: Substituting Equation 2 into Equation 1, multiplying numerator and denominator by d2, and rearranging terms: Equation 3 separates out the constant terms ((√ER/60 )(1/d2)) from the operative terms ((1+d2/d1)/ln(d2/d1))that control the position of the minimum. Close examination of Equation 3 reveals that the position of the minima is a function only of the ratio d2/d1 and not of either ER or the absolute diameter d2。 在任何初级电磁场电磁波课本里,你都能找到如下的公式:Z0 表示为d2, d1, 和ER:的公式 将公式2代入公式1,则可得公式三如下 Harbor翻译 从公式三分离常数项((√ER/60 )(1/d2)),变量((1+d2/d1)/ln(d2/d1))决定趋肤损失最小化的点,仔细检 查公式三,发现最小损失的点只与d2/d1 的比有关,与ER 和固定的d2都无关。 A plot of the operative terms from L, as a function of the argument d2/d1, shows a minimum at d2/d1=53.5911. Assuming a solid polyethylene insulation with a dielectric constant of 2.25 corresponding to a relative speed of 66% of the speed of light, the value d2/d153.5911 used in Equation 2 gives you a characteristic impedance of 51.1Ω. A long time ago, radio engineers decided to simply round off this optimal value of coaxial-cable impedance to a more convenient value of 50Ω. It turns out that the minimum in L is fairly broad and flat, so as long as you stay near 50Ω, it doesn’t much matter which impedance value you use. For example, if you produce a 75Ω cable with the same outer shield diameter and dielectric, the skin-effect loss increases by only About 12%. Different dielectrics used with the optimal d2/d1 ratio generate slightly different optimal impedances 将L作为自变量d2/d1的一个函数,一个可操作的结构显示最小损失点为d2/d1=53.5911.假设一个固体的 聚乙烯绝缘体的介电常数为2.25,相当于波的传播速度为光的66%。d2/d1=53.5911.用于公式二可得传 输线的特性阻抗为51.1Ω,很久以前,无线电工程师坚决的仅仅使同轴线的阻抗达到更方便的50Ω。 这不意味你就必须用50Ω。如,如果你设计了一个75Ω的传输线,此线有相同的外层屏蔽层直径和介 电常数,其趋肤损失仅增加了12%,不同的介电常数材料的可以优化d2/d1的值,从而产生优化的阻抗。

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