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    Microelectronic Circuits THE OXFORD SERIES IN ELECTRICAL AND COMPUTER ENGINEERING Adel S. Sedra, Series Editor Allen and Holberg, CMOS Analog Circuit Design, 3rd edition Bobrow, Elementary Linear Circuit Analysis, 2nd edition Bobrow, Fundamentals of Electrical Engineering, 2nd edition Campbell, Fabrication Engineering at the Micro- and Nanoscale, 4th edition Chen, Digital Signal Processing Chen, Linear System Theory and Design, 4th edition Chen, Signals and Systems, 3rd edition Comer, Digital Logic and State Machine Design, 3rd edition Comer, Microprocessor-Based System Design Cooper and McGillem, Probabilistic Methods of Signal and System Analysis, 3rd edition Dimitrijev, Principles of Semiconductor Device, 2nd edition Dimitrijev, Understanding Semiconductor Devices Fortney, Principles of Electronics: Analog & Digital Franco, Electric Circuits Fundamentals Ghausi, Electronic Devices and Circuits: Discrete and Integrated Guru and Hiziroğlu, Electric Machinery and Transformers, 3rd edition Houts, Signal Analysis in Linear Systems Jones, Introduction to Optical Fiber Communication Systems Krein, Elements of Power Electronics Kuo, Digital Control Systems, 2nd edition Lathi, Linear Systems and Signals, 2nd edition Lathi and Ding, Modern Digital and Analog Communication Systems, 4th edition Lathi, Signal Processing and Linear Systems Martin, Digital Integrated Circuit Design Miner, Lines and Electromagnetic Fields for Engineers Parhami, Computer Architecture Parhami, Computer Arithmetic, 2nd edition Roberts and Sedra, SPICE, 2nd edition Roberts, Taenzler, and Burns, An Introduction to Mixed-Signal IC Test and Measurement,   2nd edition Roulston, An Introduction to the Physics of Semiconductor Devices Sadiku, Elements of Electromagnetics, 6th edition Santina, Stubberud, and Hostetter, Digital Control System Design, 2nd edition Sarma, Introduction to Electrical Engineering Schaumann, Xiao, and Van Valkenburg, Design of Analog Filters, 3rd edition Schwarz and Oldham, Electrical Engineering: An Introduction, 2nd edition Sedra and Smith, Microelectronic Circuits, 7th edition Stefani, Shahian, Savant, and Hostetter, Design of Feedback Control Systems, 4th edition Tsividis/McAndrew, Operation and Modeling of the MOS Transistor, 3rd edition Van Valkenburg, Analog Filter Design Warner and Grung, Semiconductor Device Electronics Wolovich, Automatic Control Systems Yariv and Yeh, Photonics: Optical Electronics in Modern Communications, 6th edition Żak, Systems and Control SEVENTH EDITION Microelectronic Circuits Adel S. Sedra University of Waterloo Kenneth C. Smith University of Toronto New York Oxford OXFORD UNIVERSITY PRESS Oxford University Press is a department of the University of Oxford. It furthers the ­University’s objective of excellence in research, scholarship, and education by ­publishing worldwide. Oxford New York Auckland Cape Town Dar es Salaam Hong Kong Karachi Kuala Lumpur Madrid Melbourne Mexico City Nairobi New Delhi Shanghai Taipei Toronto With offices in Argentina Austria Brazil Chile Czech Republic France Greece Guatemala Hungary Italy Japan Poland Portugal Singapore South Korea Switzerland Thailand Turkey Ukraine Vietnam Copyright © 2015, 2010, 2004, 1998 by Oxford University Press; 1991, 1987 Holt, Rinehart, and Winston, Inc.; 1982 CBS College Publishing For titles covered by Section 112 of the US Higher Education Opportunity Act, please visit www.oup.com/us/he for the latest information about pricing and alternate formats. Published in the United States of America by Oxford University Press 198 Madison Avenue, New York, NY 10016 http://www.oup.com Oxford is a registered trade mark of Oxford University Press. All rights reserved. No part of this publication may be reproduced, stored in a retrieval system, or transmitted, in any form or by any means, electronic, mechanical, photocopying, recording, or otherwise, without the prior permission of Oxford University Press. Library of Congress Cataloging-in-Publication Data Sedra, Adel S., author. Microelectronic circuits / Adel S. Sedra, University of Waterloo, Kenneth C. Smith, University of Toronto. — Seventh edition.   pages cm. — (The Oxford series in electrical and computer engineering) Includes bibliographical references and index. ISBN 978–0–19–933913–6 1. Electronic circuits.  2. Integrated circuits.  I. Smith, Kenneth C. (Kenneth Carless), author.  II.  Title. TK7867.S39 2014 621.3815—dc232014033965 Multisim and National Instruments are trademarks of National Instruments. The Sedra/Smith, Microelectronic Circuits, Seventh Edition book is a product of Oxford University Press, not National Instruments Corporation or any of its affiliated companies, and Oxford University Press is solely responsible for the Sedra/Smith book and its content. Neither Oxford University Press, the Sedra/Smith book, nor any of the books and other goods and services offered by Oxford University Press are official publications of National Instruments Corporation or any of its affiliated companies, and they are not affiliated with, endorsed by, or sponsored by National Instruments Corporation or any of its affiliated companies. OrCad and PSpice are trademarks of Cadence Design Systems, Inc. The Sedra/Smith, Microelectronic Circuits, Seventh Edition book is a product of Oxford University Press, not Cadence Design Systems, Inc., or any of its affiliated companies, and Oxford University Press is solely responsible for the Sedra/Smith book and its content. Neither Oxford University Press, the Sedra/Smith book, nor any of the books and other goods and services offered by Oxford University Press are official publications of Cadence Design Systems, Inc. or any of its affiliated companies, and they are not affiliated with, endorsed by, or sponsored by Cadence Design Systems, Inc. or any of its affiliated companies. Cover Photo: This 3D IC system demonstrates the concept of wireless power delivery and communication through multiple layers of CMOS chips. The communication circuits were demonstrated in an IBM 45 nm SOI CMOS process. This technology is designed to serve a multi-Gb/s interconnect between cores spread across several IC layers for high-performance processors. (Photo Credit: The picture is courtesy of Professor David Wentzloff, Director of the Wireless Integrated Circuits Group at the University of Michigan, and was edited by Muhammad Faisal, Founder of Movellus Circuits Incorporated.) Printing number: 9 8 7 6 5 4 3 2 1 Printed in the United States of America on acid-free paper BRIEF TABLE OF CONTENTS Tables xvi “Expand-Your-Perspective” Notes  xvii Preface xix PART I  DEVICES AND BASIC CIRCUITS  2 1 Signals and Amplifiers  4 2 Operational Amplifiers  58 3 Semiconductors  134 4 Diodes  174 5 MOS Field-Effect Transistors (MOSFETs)  246 6 Bipolar Junction Transistors (BJTs)  304 7 Transistor Amplifiers  366 PART II  INTEGRATED-CIRCUIT AMPLIFIERS  506 8 Building Blocks of Integrated-Circuit Amplifiers  508 9 Differential and Multistage Amplifiers  594 10 Frequency Response  696 11 Feedback  806 12 Output Stages and Power Amplifiers  920 13 Operational Amplifier Circuits  994 PART III  DIGITAL INTEGRATED CIRCUITS  1086 14 CMOS Digital Logic Circuits  1088 15 Advanced Topics in Digital Integrated-Circuit Design  1166 16 Memory Circuits  1236 PART IV  FILTERS AND OSCILLATORS  1288 17 Filters and Tuned Amplifiers  1290 18 Signal Generators and Waveform-Shaping Circuits  1378 Appendices A–L Index  IN-1 v CONTENTS Tables xvi “Expand-Your-Perspective” Notes xvii Preface xix PART I  DEVICES AND BASIC CIRCUITS 2   1  Signals and Amplifiers  4 Introduction 5 1.1 Signals 6 1.2 Frequency Spectrum of Signals 9 1.3 Analog and Digital Signals 12 1.4 Amplifiers 15 1.4.1 Signal Amplification 15 1.4.2 Amplifier Circuit Symbol 16 1.4.3 Voltage Gain 17 1.4.4 Power Gain and Current Gain 17 1.4.5 Expressing Gain in Decibels 18 1.4.6 The Amplifier Power Supplies 18 1.4.7 Amplifier Saturation 21 1.4.8 Symbol Convention 22 1.5 Circuit Models for Amplifiers 23 1.5.1 Voltage Amplifiers 23 1.5.2 Cascaded Amplifiers 25 1.5.3 Other Amplifier Types 28 1.5.4 Relationships between the Four Amplifier Models 28 1.5.5 Determining Ri and Ro 29 1.5.6 Unilateral Models 29 1.6 Frequency Response of Amplifiers 33 1.6.1 Measuring the Amplifier Frequency Response 33 1.6.2 Amplifier Bandwidth 34 1.6.3 Evaluating the Frequency Response of Amplifiers 34 1.6.4 Single-Time-Constant Networks 35 1.6.5 Classification of Amplifiers Based on Frequency Response 41 Summary 44 Problems 45  2  Operational Amplifiers  58 Introduction 59 2.1 The Ideal Op Amp 60 2.1.1 The Op-Amp Terminals 60 2.1.2 Function and Characteristics of the Ideal Op Amp 61 2.1.3 Differential and Common-Mode Signals 63 2.2 The Inverting Configuration 64 2.2.1 The Closed-Loop Gain 65 2.2.2 Effect of the Finite Open-Loop Gain 67 2.2.3 Input and Output Resistances 68 2.2.4 An Important Application—The Weighted Summer 71 2.3 The Noninverting Configuration 73 2.3.1 The Closed-Loop Gain 73 2.3.2 Effect of Finite Open-Loop Gain 75 2.3.3 Input and Output Resistance 75 2.3.4 The Voltage Follower 75 2.4 Difference Amplifiers 77 2.4.1 A Single-Op-Amp Difference Amplifier 78 2.4.2 A Superior Circuit—The Instrumentation Amplifier 82 2.5 Integrators and Differentiators 87 2.5.1 The Inverting Configuration with General Impedances 87 2.5.2 The Inverting Integrator 89 2.5.3 The Op-Amp Differentiator 94 2.6 DC Imperfections 96 2.6.1 Offset Voltage 96 2.6.2 Input Bias and Offset Currents 100 2.6.3 Effect of VOS and IOS on the Operation of the Inverting Integrator 103 2.7 Effect of Finite Open-Loop Gain and Bandwidth on Circuit Performance 105 2.7.1 Frequency Dependence of the Open-Loop Gain 105 2.7.2 Frequency Response of Closed-Loop Amplifiers 107 vi Contents  vii  2.8 Large-Signal Operation of Op Amps 110 2.8.1 Output Voltage Saturation 110 2.8.2 Output Current Limits 110 2.8.3 Slew Rate 112 2.8.4 Full-Power Bandwidth 114 Summary 115 Problems 116   3 Semiconductors  134 Introduction 135 3.1 Intrinsic Semiconductors 136 3.2 Doped Semiconductors 139 3.3 Current Flow in Semiconductors 142 3.3.1 Drift Current 142 3.3.2 Diffusion Current 145 3.3.3 Relationship between D and μ  148 3.4 The pn Junction 148 3.4.1 Physical Structure 149 3.4.2 Operation with Open-Circuit Terminals 149 3.5 The pn Junction with an Applied Voltage 155 3.5.1 Qualitative Description of Junction Operation 155 3.5.2 The Current–Voltage Relationship of the Junction 158 3.5.3 Reverse Breakdown 162 3.6 Capacitive Effects in the pn Junction 164 3.6.1 Depletion or Junction Capacitance 164 3.6.2 Diffusion Capacitance 166 Summary 168 Problems 171   4 Diodes  174 Introduction 175 4.1 The Ideal Diode 176 4.1.1 Current–Voltage Characteristic 176 4.1.2 A Simple Application: The Rectifier 177 4.1.3 Another Application: Diode Logic Gates 180 4.2 Terminal Characteristics of Junction Diodes 184 4.2.1 The Forward-Bias Region 184 4.2.2 The Reverse-Bias Region 189 4.2.3 The Breakdown Region 190 4.3 Modeling the Diode Forward Characteristic 190 4.3.1 The Exponential Model 190 4.3.2 Graphical Analysis Using the Exponential Model 191 4.3.3 Iterative Analysis Using the Exponential Model 191 4.3.4 The Need for Rapid Analysis 192 4.3.5 The Constant-Voltage-Drop Model 193 4.3.6 The Ideal-Diode Model 194 4.3.7 The Small-Signal Model 195 4.3.8 Use of the Diode Forward Drop in Voltage Regulation 200 4.4 Operation in the Reverse Breakdown Region—Zener Diodes 202 4.4.1 Specifying and Modeling the Zener Diode 203 4.4.2 Use of the Zener as a Shunt Regulator 204 4.4.3 Temperature Effects 206 4.4.4 A Final Remark 207 4.5 Rectifier Circuits 207 4.5.1 The Half-Wave Rectifier 208 4.5.2 The Full-Wave Rectifier 210 4.5.3 The Bridge Rectifier 212 4.5.4 The Rectifier with a Filter Capacitor—The Peak Rectifier 213 4.5.5 Precision Half-Wave Rectifier—The Superdiode 219 4.6 Limiting and Clamping Circuits 221 4.6.1 Limiter Circuits 221 4.6.2 The Clamped Capacitor or DC Restorer 224 4.6.3 The Voltage Doubler 226 4.7 Special Diode Types 227 4.7.1 The Schottky-Barrier Diode (SBD) 227 4.7.2 Varactors 228 4.7.3 Photodiodes 228 4.7.4 Light-Emitting Diodes (LEDs) 228 Summary 229 Problems 230   5 MOS Field-Effect Transistors (MOSFETs) 246 Introduction 247 5.1 Device Structure and Physical Operation 248 5.1.1 Device Structure 248 5.1.2 Operation with Zero Gate Voltage 250 viii Contents 5.1.3 Creating a Channel for Current Flow 250 5.1.4 Applying a Small vDS 252 5.1.5 Operation as vDS Is Increased 256 5.1.6 Operation for vDS ≥ VOV: Channel Pinch-Off and Current Saturation 258 5.1.7 The p-Channel MOSFET 261 5.1.8 Complementary MOS or CMOS 263 5.1.9 Operating the MOS Transistor in the Subthreshold Region 264 5.2 Current–Voltage Characteristics 264 5.2.1 Circuit Symbol 264 5.2.2 The iD–vDS Characteristics 265 5.2.3 The iD–vGS Characteristic 267 5.2.4 Finite Output Resistance in Saturation 271 5.2.5 Characteristics of the p-Channel MOSFET 274 5.3 MOSFET Circuits at DC 276 5.4 The Body Effect and Other Topics 288 5.4.1 The Role of the Substrate—The Body Effect 288 5.4.2 Temperature Effects 289 5.4.3 Breakdown and Input Protection 289 5.4.4 Velocity Saturation 290 5.4.5 The Depletion-Type MOSFET 290 Summary 291 Problems 292   6 Bipolar Junction Transistors (BJTs) 304 Introduction 305 6.1 Device Structure and Physical Operation 306 6.1.1 Simplified Structure and Modes of Operation 306 6.1.2 Operation of the npn Transistor in the Active Mode 307 6.1.3 Structure of Actual Transistors 315 6.1.4 Operation in the Saturation Mode 316 6.1.5 The pnp Transistor 318 6.2 Current–Voltage Characteristics 320 6.2.1 Circuit Symbols and Conventions 320 6.2.2 G raphical Representation of Transistor Characteristics 325 6.2.3 D ependence of i C on the Collector Voltage—The Early Effect 326 6.2.4 A n Alternative Form of the Common- Emitter Characteristics 329 6.3 BJT Circuits at DC 333 6.4 Transistor Breakdown and Temperature Effects 351 6.4.1 Transistor Breakdown 351 6.4.2 Dependence of β on IC and Temperature 353 Summary 354 Problems 355   7  Transistor Amplifiers  366 Introduction 367 7.1 Basic Principles 368 7.1.1 The Basis for Amplifier Operation 368 7.1.2 Obtaining a Voltage Amplifier 369 7.1.3 The Voltage-Transfer Characteristic (VTC) 370 7.1.4 Obtaining Linear Amplification by Biasing the Transistor 371 7.1.5 The Small-Signal Voltage Gain 374 7.1.6 Determining the VTC by Graphical Analysis 380 7.1.7 Deciding on a Location for the Bias Point Q 381 7.2 Small-Signal Operation and Models 383 7.2.1 The MOSFET Case 383 7.2.2 The BJT Case 399 7.2.3 Summary Tables 420 7.3 Basic Configurations 423 7.3.1 The Three Basic Configurations 423 7.3.2 Characterizing Amplifiers 424 7.3.3 The Common-Source (CS) and Common-Emitter (CE) Amplifiers 426 7.3.4 The Common-Source (CommonEmitter) Amplifier with a Source (Emitter) Resistance 431 7.3.5 The Common-Gate (CG) and the Common-Base (CB) Amplifiers 439 7.3.6 The Source and Emitter Followers 442 7.3.7 Summary Tables and Comparisons 452 Contents  ix  7.3.8 When and How to Include the Transistor Output Resistance ro 453 7.4 Biasing 454 7.4.1 The MOSFET Case 455 7.4.2 The BJT Case 461 7.5 Discrete-Circuit Amplifiers 467 7.5.1 A Common-Source (CS) Amplifier 467 7.5.2 A Common-Emitter (CE) Amplifier 470 7.5.3 A Common-Emitter Amplifier with an Emitter Resistance Re 471 7.5.4 A Common-Base (CB) Amplifier 473 7.5.5 An Emitter Follower 475 7.5.6 The Amplifier Frequency Response 477 Summary 479 Problems 480 PART II  INTEGRATED-CIRCUIT AMPLIFIERS 506   8 Building Blocks of IntegratedCircuit Amplifiers  508 Introduction 509 8.1 IC Design Philosophy 510 8.2 IC Biasing—Current Sources, Current Mirrors, and Current-Steering Circuits 511 8.2.1 The Basic MOSFET Current Source 512 8.2.2 MOS Current-Steering Circuits 515 8.2.3 BJT Circuits 518 8.2.4 Small-Signal Operation of Current Mirrors 523 8.3 The Basic Gain Cell 525 8.3.1 The CS and CE Amplifiers with Current-Source Loads 525 8.3.2 The Intrinsic Gain 527 8.3.3 Effect of the Output Resistance of the Current-Source Load 530 8.3.4 Increasing the Gain of the Basic Cell 536 8.4 The Common-Gate and Common-Base Amplifiers 537 8.4.1 The CG Circuit 537 8.4.2 Output Resistance of a CS Amplifier with a Source Resistance 541 8.4.3 The Body Effect 542 8.4.4 The CB Circuit 543 8.4.5 Output Resistance of an Emitter- Degenerated CE Amplifier 546 8.5 The Cascode Amplifier 546 8.5.1 Cascoding 546 8.5.2 The MOS Cascode Amplifier 547 8.5.3 Distribution of Voltage Gain in a Cascode Amplifier 552 8.5.4 Double Cascoding 555 8.5.5 The Folded Cascode 555 8.5.6 The BJT Cascode 557 8.6 C urrent-Mirror Circuits with Improved Performance 559 8.6.1 Cascode MOS Mirrors 559 8.6.2 The Wilson Current Mirror 560 8.6.3 The Wilson MOS Mirror 563 8.6.4 The Widlar Current Souce 565 8.7 Some Useful Transistor Pairings 567 8.7.1 The CC–CE, CD–CS, and CD–CE Configurations 567 8.7.2 The Darlington Configuration 571 8.7.3 The CC–CB and CD–CG Configurations 572 Summary 575 Problems 576   9 Differential and Multistage Amplifiers 594 Introduction 595 9.1 The MOS Differential Pair 596 9.1.1 Operation with a Common-Mode Input Voltage 597 9.1.2 Operation with a Differential Input Voltage 601 9.1.3 Large-Signal Operation 602 9.1.4 Small-Signal Operation 607 9.1.5 The Differential Amplifier with Current-Source Loads 611 9.1.6 Cascode Differential Amplifier 612 9.2 The BJT Differential Pair 614 9.2.1 Basic Operation 614 9.2.2 Input Common-Mode Range 616 9.2.3 Large-Signal Operation 617 9.2.4 Small-Signal Operation 620 9.3 Common-Mode Rejection 627 9.3.1 The MOS Case 628 9.3.2 The BJT Case 634 9.4 DC Offset 637 x Contents 9.4.1 Input Offset Voltage of the MOS Differential Amplifier 637 9.4.2 Input Offset Voltage of the Bipolar Differential Amplifier 640 9.4.3 Input Bias and Offset Currents of the Bipolar Differential Amplifier 643 9.4.4 A Concluding Remark 644 9.5 The Differential Amplifier with a Current-Mirror Load 644 9.5.1 Differential to Single-Ended Conversion 644 9.5.2 The Current-Mirror-Loaded MOS Differential Pair 645 9.5.3 Differential Gain of the Current-Mirror-Loaded MOS Pair 647 9.5.4 The Bipolar Differential Pair with a Current-Mirror Load 651 9.5.5 Common-Mode Gain and CMRR 655 9.6 Multistage Amplifiers 659 9.6.1 A Two-Stage CMOS Op Amp 659 9.6.2 A Bipolar Op Amp 664 Summary 672 Problems 674   10  Frequency Response  696 Introduction 697 10.1 Low-Frequency Response of Discrete-Circuit CommonSource and Common-Emitter Amplifiers 699 10.1.1 The CS Amplifier 699 10.1.2 The Method of Short-Circuit Time-Constants 707 10.1.3 The CE Amplifier 707 10.2 Internal Capacitive Effects and the High-Frequency Model of the MOSFET and the BJT 711 10.2.1 The MOSFET 711 10.2.2 The BJT 717 10.3 High-Frequency Response of the CS and CE Amplifiers 722 10.3.1 The Common-Source Amplifier 722 10.3.2 The Common-Emitter Amplifier 728 10.3.3 Miller’s Theorem 732 10.3.4 Frequency Response of the CS Amplifier When Rsig Is Low 735 10.4 Useful Tools for the Analysis of the High-Frequency Response of Amplifiers 739 10.4.1 The High-Frequency Gain Function 739 10.4.2 Determining the 3-dB Frequency fH 740 10.4.3 The Method of Open-Circuit Time Constants 743 10.4.4 Application of the Method of Open-Circuit Time Constants to the CS Amplifier 744 10.4.5 Application of the Method of Open-Circuit Time Constants to the CE Amplifier 748 10.5 High-Frequency Response of the Common-Gate and Cascode Amplifiers 748 10.5.1 High-Frequency Response of the CG Amplifier 748 10.5.2 High-Frequency Response of the MOS Cascode Amplifier 754 10.5.3 High-Frequency Response of the Bipolar Cascode Amplifier 759 10.6 High-Frequency Response of the Source and Emitter Followers 760 10.6.1 The Source-Follower Case 761 10.6.2 The Emitter-Follower Case 767 10.7 High-Frequency Response of Differential Amplifiers 768 10.7.1 Analysis of the Resistively Loaded MOS Amplifier 768 10.7.2 Analysis of the Current-MirrorLoaded MOS Amplifier 772 10.8 Other Wideband Amplifier Configurations 778 10.8.1 Obtaining Wideband Amplification by Source and Emitter Degeneration 778 10.8.2 The CD–CS, CC–CE, and CD–CE Configurations 781 10.8.3 The CC–CB and CD–CG Configurations 786 Summary 788 Problems 789   11 Feedback  806 Introduction 807 11.1 The General Feedback Structure 808 11.1.1 Signal-Flow Diagram 808 11.1.2 The Closed-Loop Gain 809 Contents  xi  11.1.3 The Loop Gain 810 11.1.4 Summary 814 11.2 Some Properties of Negative Feedback 815 11.2.1 Gain Desensitivity 815 11.2.2 Bandwidth Extension 816 11.2.3 Interference Reduction 817 11.2.4 Reduction in Nonlinear Distortion 819 11.3 The Feedback Voltage Amplifier 820 11.3.1 The Series–Shunt Feedback Topology 820 11.3.2 Examples of Series–Shunt Feedback Amplifiers 821 11.3.3 Analysis of the Feedback Voltage Amplifier Utilizing the Loop Gain 823 11.3.4 A Final Remark 828 11.4 Systematic Analysis of Feedback Voltage Amplifiers 828 11.4.1 The Ideal Case 829 11.4.2 The Practical Case 831 11.5 Other Feedback Amplifier Types 840 11.5.1 Basic Principles 840 11.5.2 The Feedback Transconductance Amplifier (Series–Series) 844 11.5.3 The Feedback Transresistance Amplifier (Shunt–Shunt) 855 11.5.4 The Feedback Current Amplifier (Shunt–Series) 865 11.6 Summary of the Feedback Analysis Method 871 11.7 The Stability Problem 871 11.7.1 Transfer Function of the Feedback Amplifier 871 11.7.2 The Nyquist Plot 873 11.8 Effect of Feedback on the Amplifier Poles 875 11.8.1 Stability and Pole Location 875 11.8.2 Poles of the Feedback Amplifier 876 11.8.3 Amplifier with a Single-Pole Response 877 11.8.4 Amplifier with a Two-Pole Response 878 11.8.5 Amplifiers with Three or More Poles 883 11.9 Stability Study Using Bode Plots 885 11.9.1 Gain and Phase Margins 885 11.9.2 Effect of Phase Margin on Closed-Loop Response 886 11.9.3 An Alternative Approach for Investigating Stability 887 11.10 Frequency Compensation 889 11.10.1 Theory 889 11.10.2 Implementation 891 11.10.3 Miller Compensation and Pole Splitting 892 Summary 895 Problems 896   12 Output Stages and Power Amplifiers 920 Introduction 921 12.1 Classification of Output Stages 922 12.2 Class A Output Stage 923 12.2.1 Transfer Characteristic 924 12.2.2 Signal Waveforms 925 12.2.3 Power Dissipation 926 12.2.4 Power-Conversion Efficiency 928 12.3 Class B Output Stage 929 12.3.1 Circuit Operation 929 12.3.2 Transfer Characteristic 929 12.3.3 Power-Conversion Efficiency 930 12.3.4 Power Dissipation 931 12.3.5 Reducing Crossover Distortion 933 12.3.6 Single-Supply Operation 934 12.4 Class AB Output Stage 935 12.4.1 Circuit Operation 935 12.4.2 Output Resistance 937 12.5 Biasing the Class AB Circuit 940 12.5.1 Biasing Using Diodes 940 12.5.2 Biasing Using the VBE Multiplier 942 12.6 Variations on the Class AB Configuration 945 12.6.1 Use of Input Emitter Followers 945 12.6.2 Use of Compound Devices 946 12.6.3 Short-Circuit Protection 949 12.6.4 Thermal Shutdown 950 12.7 CMOS Class AB Output Stages 950 12.7.1 The Classical Configuration 950 12.7.2 An Alternative Circuit Utilizing Common-Source Transistors 953 12.8 IC Power Amplifiers 961 12.8.1 A Fixed-Gain IC Power Amplifier 962 xii Contents 12.8.2 T he Bridge Amplifier 966 12.9 Class D Power Amplifiers 967 12.10 Power Transistors 971 12.10.1 Packages and Heat Sinks 971 12.10.2 Power BJTs 972 12.10.3 Power MOSFETs 974 12.10.4 Thermal Considerations 976 Summary 982 Problems 983   13 Operational-Amplifier Circuits 994 Introduction 995 13.1 The Two-Stage CMOS Op Amp 996 13.1.1 The Circuit 997 13.1.2 Input Common-Mode Range and Output Swing 998 13.1.3 DC Voltage Gain 999 13.1.4 Common-Mode Rejection Ratio (CMRR) 1001 13.1.5 Frequency Response 1002 13.1.6 Slew Rate 1007 13.1.7 Power-Supply Rejection Ratio (PSRR) 1008 13.1.8 Design Trade-Offs 1009 13.1.9 A Bias Circuit for the Two-Stage CMOS Op Amp 1010 13.2 T he Folded-Cascode CMOS Op Amp 1016 13.2.1 The Circuit 1016 13.2.2 Input Common-Mode Range and Output Swing 1018 13.2.3 Voltage Gain 1020 13.2.4 Frequency Response 1021 13.2.5 Slew Rate 1022 13.2.6 Increasing the Input Common- Mode Range: Rail-to-Rail Input Operation 1024 13.2.7 Increasing the Output Voltage Range: The Wide-Swing Current Mirror 1026 13.3 The 741 BJT Op Amp 1028 13.3.1 The 741 Circuit 1028 13.3.2 DC Analysis 1032 13.3.3 Small-Signal Analysis 1038 13.3.4 Frequency Response 1051 13.3.5 Slew Rate 1053 13.4 Modern Techniques for the Design of BJT Op Amps 1054 13.4.1 Special Performance Requirements 1054 13.4.2 Bias Design 1056 13.4.3 Design of the Input Stage to Obtain Rail-to-Rail VICM 1058 13.4.4 Common-Mode Feedback to Control the DC Voltage at the Output of the Input Stage 1064 13.4.5 Output-Stage Design for Near Rail-to-Rail Output Swing 1069 13.4.6 Concluding Remark 1073 Summary 1073 Problems 1074 PART III  DIGITAL INTEGRATED CIRCUITS 1086   14  CMOS Digital Logic Circuits 1088 Introduction 1089 14.1 CMOS Logic-Gate Circuits 1090 14.1.1 Switch-Level Transistor Model 1090 14.1.2 The CMOS Inverter 1091 14.1.3 General Structure of CMOS Logic 1091 14.1.4 The Two-Input NOR Gate 1094 14.1.5 The Two-Input NAND Gate 1095 14.1.6 A Complex Gate 1096 14.1.7 Obtaining the PUN from the PDN and Vice Versa 1096 14.1.8 The Exclusive-OR Function 1097 14.1.9 Summary of the Synthesis Method 1098 14.2 Digital Logic Inverters 1100 14.2.1 The Voltage-Transfer Characteristic (VTC) 1100 14.2.2 Noise Margins 1101 14.2.3 The Ideal VTC 1103 14.2.4 Inverter Implementation 1103 14.3 The CMOS Inverter 1114 14.3.1 Circuit Operation 1114 14.3.2 The Voltage-Transfer Characteristic (VTC) 1117 14.3.3 The Situation When QN and QP Are Not Matched 1120 14.4 Dynamic Operation of the CMOS Inverter 1125 Contents  xiii  14.4.1 Propagation Delay 1125 14.4.2 Determining the Propagation Delay of the CMOS Inverter 1129 14.4.3 Determining the Equivalent Load Capacitance C 1136 14.5 Transistor Sizing 1139 14.5.1 Inverter Sizing 1139 14.5.2 Transistor Sizing in CMOS Logic Gates 1141 14.5.3 Effects of Fan-In and Fan-Out on Propagation Delay 1145 14.5.4 Driving a Large Capacitance 1146 14.6 Power Dissipation 1149 14.6.1 Sources of Power Dissipation 1149 14.6.2 Power–Delay and Energy–Delay Products 1152 Summary 1154 Problems 1156   15 Advanced Topics in Digital Integrated-Circuit Design  1166 Introduction 1167 15.1 Implications of Technology Scaling: Issues in Deep-Submicron Design 1168 15.1.1 Silicon Area 1169 15.1.2 Scaling Implications 1169 15.1.3 Velocity Saturation 1171 15.1.4 Subthreshold Conduction 1177 15.1.5 Temperature, Voltage, and Process Variations 1178 15.1.6 Wiring: The Interconnect 1178 15.2 Digital IC Technologies, Logic-Circuit Families, and Design Methodologies 1179 15.2.1 Digital IC Technologies and Logic-Circuit Families 1180 15.2.2 Styles for Digital System Design 1182 15.2.3 Design Abstraction and Computer Aids 1182 15.3 Pseudo-NMOS Logic Circuits 1183 15.3.1 The Pseudo-NMOS Inverter 1183 15.3.2 Static Characteristics 1184 15.3.3 Derivation of the VTC 1186 15.3.4 Dynamic Operation 1188 15.3.5 Design 1189 15.3.6 Gate Circuits 1189 15.3.7 Concluding Remarks 1190 15.4 Pass-Transistor Logic Circuits 1192 15.4.1 An Essential Design Requirement 1193 15.4.2 Operation with NMOS Transistors as Switches 1194 15.4.3 Restoring the Value of VOH to VDD 1198 15.4.4 The Use of CMOS Transmission Gates as Switches 1199 15.4.5 Examples of Pass-Transistor Logic Circuits 1206 15.4.6 A Final Remark 1208 15.5 Dynamic MOS Logic Circuits 1208 15.5.1 The Basic Principle 1209 15.5.2 Nonideal Effects 1212 15.5.3 Domino CMOS Logic 1216 15.5.4 Concluding Remarks 1217 15.6 Bipolar and BiCMOS Logic Circuits 1217 15.6.1 Emitter-Coupled Logic (ECL) 1218 15.6.2 BiCMOS Digital Circuits 1223 Summary 1226 Problems 1227   16  Memory Circuits  1236 Introduction 1237 16.1 Latches and Flip-Flops 1238 16.1.1 The Latch 1238 16.1.2 The SR Flip-Flop 1240 16.1.3 CMOS Implementation of SR Flip-Flops 1241 16.1.4 A Simpler CMOS Implementation of the Clocked SR Flip-Flop 1247 16.1.5 D Flip-Flop Circuits 1247 16.2 Semiconductor Memories: Types and Architectures 1249 16.2.1 Memory-Chip Organization 1250 16.2.2 Memory-Chip Timing 1252 16.3 Random-Access Memory (RAM) Cells 1253 16.3.1 Static Memory (SRAM) Cell 1253 16.3.2 Dynamic Memory (DRAM) Cell 1260 16.4 Sense Amplifiers and Address Decoders 1262 16.4.1 The Sense Amplifier 1263 xiv Contents 16.4.2 The Row-Address Decoder 1271 16.4.3 The Column-Address Decoder 1273 16.4.4 Pulse-Generation Circuits 1274 16.5 Read-Only Memory (ROM) 1276 16.5.1 A MOS ROM 1276 16.5.2 Mask Programmable ROMs 1278 16.5.3 Programmable ROMs (PROMs, EPROMs, and Flash) 1279 16.6 CMOS Image Sensors 1281 Summary 1282 Problems 1283 PART IV  FILTERS AND OSCILLATORS 1288   17  Filters and Tuned Amplifiers 1290 Introduction 1291 17.1 Filter Transmission, Types, and Specification 1292 17.1.1 Filter Transmission 1292 17.1.2 Filter Types 1293 17.1.3 Filter Specification 1293 17.2 The Filter Transfer Function 1296 17.3 Butterworth and Chebyshev Filters 1300 17.3.1 The Butterworth Filter 1300 17.3.2 The Chebyshev Filter 1304 17.4 First-Order and Second-Order Filter Functions 1307 17.4.1 First-Order Filters 1308 17.4.2 Second-Order Filter Functions 1311 17.5 The Second-Order LCR Resonator 1316 17.5.1 The Resonator Natural Modes 1316 17.5.2 Realization of Transmission Zeros 1317 17.5.3 Realization of the Low-Pass Function 1317 17.5.4 Realization of the High-Pass Function 1319 17.5.5 Realization of the Bandpass Function 1319 17.5.6 Realization of the Notch Functions 1319 17.5.7 Realization of the All-Pass Function 1321 17.6 Second-Order Active Filters Based on Inductor Replacement 1322 17.6.1 The Antoniou InductanceSimulation Circuit 1322 17.6.2 The Op Amp–RC Resonator 1323 17.6.3 Realization of the Various Filter Types 1325 17.6.4 The All-Pass Circuit 1325 17.7 Second-Order Active Filters Based on the Two-Integrator-Loop Topology 1330 17.7.1 Derivation of the Two-Integrator- Loop Biquad 1330 17.7.2 Circuit Implementation 1332 17.7.3 An Alternative Two-Integrator- Loop Biquad Circuit 1334 17.7.4 Final Remarks 1335 17.8 Single-Amplifier Biquadratic Active Filters 1336 17.8.1 Synthesis of the Feedback Loop 1336 17.8.2 Injecting the Input Signal 1339 17.8.3 Generation of Equivalent Feedback Loops 1341 17.9 Sensitivity 1344 17.10 Transconductance-C Filters 1347 17.10.1 Methods for IC Filter Implementation 1347 17.10.2 Transconductors 1348 17.10.3 Basic Building Blocks 1349 17.10.4 Second-Order Gm−C Filter 1351 17.11 Switched-Capacitor Filters 1354 17.11.1 The Basic Principle 1354 17.11.2 Practical Circuits 1356 17.11.3 Final Remarks 1359 17.12 Tuned Amplifiers 1359 17.12.1 The Basic Principle 1360 17.12.2 Inductor Losses 1362 17.12.3 Use of Transformers 1363 17.12.4 Amplifiers with Multiple Tuned Circuits 1365 17.12.5 The Cascode and the CC–CB Cascade 1366 17.12.6 Synchronous Tuning and Stagger Tuning 1367 Summary 1368 Problems 1369   18 Signal Generators and Waveform-Shaping Circuits 1378 Introduction 1379 18.1 Basic Principles of Sinusoidal Oscillators 1380 Contents  xv  18.1.1 The Oscillator Feedback Loop 1380 18.1.2 The Oscillation Criterion 1381 18.1.3 Analysis of Oscillator Circuits 1382 18.1.4 Nonlinear Amplitude Control 1385 18.1.5 A Popular Limiter Circuit for Amplitude Control 1386 18.2 Op Amp–RC Oscillator Circuits 1388 18.2.1 The Wien-Bridge Oscillator 1388 18.2.2 The Phase-Shift Oscillator 1391 18.2.3 The Quadrature Oscillator 1392 18.2.4 The Active-Filter-Tuned Oscillator 1394 18.2.5 A Final Remark 1396 18.3 LC and Crystal Oscillators 1396 18.3.1 The Colpitts and Hartley Oscillators 1396 18.3.2 The Cross-Coupled LC Oscillator 1400 18.3.3 Crystal Oscillators 1402 18.4 Bistable Multivibrators 1404 18.4.1 The Feedback Loop 1405 18.4.2 Transfer Characteristic of the Bistable Circuit 1406 18.4.3 Triggering the Bistable Circuit 1407 18.4.4 The Bistable Circuit as a Memory Element 1407 18.4.5 A Bistable Circuit with Noninverting Transfer Characteristic 1408 18.4.6 Application of the Bistable Circuit as a Comparator 1409 18.4.7 Making the Output Levels More Precise 1411 18.5 Generation of Square and Triangular Waveforms Using Astable Multivibrators 1412 18.5.1 Operation of the Astable Multivibrator 1413 18.5.2 Generation of Triangular Waveforms 1415 18.6 Generation of a Standardized Pulse: The Monostable Multivibrator 1417 18.7 Integrated-Circuit Timers 1419 18.7.1 The 555 Circuit 1419 18.7.2 Implementing a Monostable Multivibrator Using the 555 IC 1420 18.7.3 An Astable Multivibrator Using the 555 IC 1420 18.8 Nonlinear Waveform-Shaping Circuits 1424 18.8.1 The Breakpoint Method 1424 18.8.2 The Nonlinear-Amplification Method 1426 Summary 1428 Problems 1428 Appendices A. VLSI Fabrication Technology (on website) A-1 B. SPICE Device Models and Design and Simulation Examples Using PSpice® and MultisimTM (on website)  B-1 C. Two-Port Network Parameters (on website) C-1 D. Some Useful Network Theorems (on website) D-1 E. Single-Time-Constant Circuits (on website) E-1 F. s-Domain Analysis: Poles, Zeros, and Bode Plots (on website)  F-1 G. Comparison of the MOSFET and the BJT (on website, also Table G.3 in text)  G-1 H. Design of Stagger-Tuned Amplifiers (on website) H-1 I. Bibliography (on website)  I-1 J. Standard Resistance Values and Unit Prefixes J-1 K. Typical Parameter Values for IC Devices Fabricated in CMOS and Bipolar Processes  K-1 L. Answers to Selected Problems (on website) L-1 Index  IN-1 TABLES FOR REFERENCE AND STUDY Table 1.1 The Four Amplifier Types  28 Table 1.2 Frequency Response of STC Networks  36 Table 2.1 Characteristics of the Ideal Op Amp  62 Table 3.1 Summary of Important Semiconductor Equations  169 Table 5.1 Regions of Operation of the NMOS Transistor  266 Table 5.2 Regions of Operation of the PMOS Transistor  275 Table 6.1 BJT Modes of Operation  307 Table 6.2 Summary of the BJT Current–Voltage Relationships in the Active Mode 322 Table 6.3 Simplified Models for the Operation of the BJT in DC Circuits  334 Table 7.1 Systematic Procedure for the Analysis of Transistor Amplifier Circuits 421 Table 7.2 Small-Signal Models of the MOSFET  421 Table 7.3 Small-Signal Models of the BJT  422 Table 7.4 Characteristics of MOSFET Amplifiers  452 Table 7.5 Characteristics of BJT Amplifiers  453 Table 8.1 Gain Distribution in the MOS Cascode Amplifier for Various Values of RL 554 Table 10.1 The MOSFET High-Frequency Model  716 Table 10.2 The BJT High-Frequency Model  722 Table 11.1 Summary of the Parameters and Formulas for the Ideal Feedback-Amplifier Structure of Fig. 11.1  815 Table 11.2 Summary of Relationships for the Four Feedback- Amplifier Topologies 872 Table 13.1 DC Collector Currents of the 741 Circuit (μA)  1038 Table 14.1 Important Parameters of the VTC of the Logic Inverter 1102 Table 14.2 Summary of Important Characteristics of the CMOS Logic Inverter 1155 Table 15.1 Implications of Device and Voltage Scaling  1170 Table 15.2 Regions of Operation of the Pseudo-NMOS Inverter  1187 Table 17.1 Design Data for the Circuits Based on Inductance Simulation (Fig 17.22)  1328 Table 17.2 Design Data for the Tow-Thomas Biquad Circuit in Fig 17.26  1335 Table G.3 Comparison of the MOSFET and the BJT  G-1 Table J.1 Standard Resistance Values  J-1 Table J.2 SI Unit Prefixes  J-2 Table J.3 Meter Conversion Factors  J-2 Table K.1 Typical Values of CMOS Device Parameters  K-1 Table K.2 Typical Parameter Values for BJTs  K-1 xvi “EXPAND-YOUR-PERSPECTIVE” NOTES Chapter 1: Analog vs. Digital Circuit Engineers  15 Chapter 1: Bode Plots  37 Chapter 2: Integrated Instrumentation Amplifiers  85 Chapter 2: Early Op Amps and Analog Computation  88 Chapter 3: LCDs, the Face of Electronics  139 Chapter 4: The Earliest Semiconductor Diode  219 Chapter 4: From Indication to Illumination  229 Chapter 5: The First Field-Effect Devices  248 Chapter 5: Gordon Moore—His Law  288 Chapter 6: The Invention of the BJT  320 Chapter 7: Shockley and Silicon Valley  405 Chapter 7: Lee de Forest—a Father of the Electronics Age  454 Chapter 8: Solid Circuits with “Flying Wires”  511 Chapter 8: The Integrated Circuit 525 Chapter 9: The Long-Tailed Pair  612 Chapter 9:  The International Solid-State Circuits Conference (ISSCC) 659 Chapter 10: John Milton Miller—Capacitance Multiplication  735 Chapter 10: RFID—Identification at a Distance  772 Chapter 11: Feedback—Historical Note  823 Chapter 11: Harry Nyquist—A Diverse Electronics Fundamentalist 875 Chapter 12: Early Power-Op-Amp Product  962 Chapter 12: Hans Camenzind—the Inventor of the Class D Amplifier 968 Chapter 13: The Genie of Analog  996 Chapter 13: The Creator of the μA741—David Fullagar  1031 Chapter 14: Frank Marion Wanless—the Inventor of CMOS  1117 Chapter 14: Federico Faggin—a Pioneer in Microprocessor Electronics 1141 Chapter 15: The Invisible Computer  1182 Chapter 15: Grand-Scale Graphics  1213 Chapter 16: Flip-Flop Fact  1240 Chapter 16: Blinding Flash  1282 Chapter 17: A Brief History of Analog Filters  1295 Chapter 17: Early Filter Pioneers—Cauer and Darlington  1348 Chapter 18: The Wien-Bridge Oscillator  1390 Chapter 18: Oscillator Pioneers  1400 xvii PREFACE Microelectronic Circuits, Seventh Edition, is intended as a text for the core courses in electronic circuits taught to majors in electrical and computer engineering. It should also prove useful to engineers and other professionals wishing to update their knowledge through self-study. As was the case with the first six editions, the objective of this book is to develop in the reader the ability to analyze and design electronic circuits, both analog and digital, discrete and integrated. While the application of integrated circuits is covered, emphasis is placed on transistor circuit design. This is done because of our belief that even if the majority of those studying this book were not to pursue a career in IC design, knowledge of what is inside the IC package would enable intelligent and innovative application of such chips. Furthermore, with the advances in VLSI technology and design methodology, IC design itself has become accessible to an increasing number of engineers. Prerequisites The prerequisite for studying the material in this book is a first course in circuit analysis. As a review, some linear circuits material is included here in the appendices: specifically, two-port network parameters in Appendix C; some useful network theorems in Appendix D; single-time-constant circuits in Appendix E; and s-domain analysis in Appendix F. In addition, a number of relevant circuit analysis problems are included at the beginning of the end-of-chapter problems section of Chapter 1. No prior knowledge of physical electronics is assumed. All required semiconductor device physics is included, and Appendix A provides a brief description of IC fabrication. All these appendices can be found on the book’s website. Emphasis on Design It has been our philosophy that circuit design is best taught by pointing out the various tradeoffs available in selecting a circuit configuration and in selecting component values for a given configuration. The emphasis on design has been retained in this edition. In addition to design examples, and design-oriented exercises and end-of-chapter problems (indicated with a D), the book includes on its website an extensive appendix (Appendix B) where a large number of simulation and design examples are presented. These emphasize the use of SPICE, the most valuable circuit-design aid. xix xx Preface New to the Seventh Edition While maintaining the philosophy and pedagogical approach of the first six editions, several changes have been made to both organization and coverage. Our goal in making structural changes has been to increase modularity and thus flexibility for the instructor, without causing disturbance to courses currently using the sixth edition. Changes in coverage are necessitated by the continuing advances in technology which make some topics of greater relevance and others of less interest. As well, advances in IC process technology require that the numbers used in the examples, exercises and end-of-chapter problems be updated to reflect the parameters of newer generations of IC technologies (e.g., some problems utilize the parameters of the 65-nm CMOS process). This ensures that students are acquiring a real-world perspective on technology. To improve presentation, a number of chapters and sections have been rewritten for greater clarity. Specific, noteworthy changes are: 1. New End-of-Chapter Problems and a New Instructor’s Solutions Manual.  The number of the end-of-chapter problems has increased by about 50. Of the resulting 1532 problems, 176 are entirely new and 790 have new data. The new Instructor’s Solutions Manual is written by Adel Sedra. 2. Expand-Your-Perspective Notes.  This is a new feature providing historical and application perspectives. About two such notes are included in each chapter. Most are focused on notable circuit engineers and key inventions. 3. Greater Flexibility in Presenting the MOSFET and the BJT. Two short and completely parallel chapters present the MOSFET (Chapter 5) and the BJT (Chapter 6). Here the focus is on the device structure and its physical operation, its current-voltage characteristics, and its application in dc circuits. The order of coverage of these two chapters is entirely at the instructor’s discretion as they have been written to be completely independent of each other. 4. A Unified Treatment of Transistor Amplifiers. The heart of a first course in electronics is the study of transistor amplifiers. The seventh edition provides a new approach to this subject: A new Chapter 7 begins with the basic principles that underlie the operation of a transistor of either type as an amplifier, and presents such concepts as small-signal operation and modeling. This is followed by the classical configurations of transistor amplifiers, biasing methods, and practical discrete-circuit amplifiers. The combined presentation emphasizes the unity of the basic principles while allowing for separate treatment of the two device types where this is warranted. Very importantly, we are able to compare the two devices and to draw conclusions about their unique areas of application. 5. Improved Presentation of Cascoding. Chapter 8 dealing with the basic building blocks of IC amplifiers has been rewritten to improve presentation. Specifically, the development of cascoding and the key circuit building blocks, the cascode amplifier and the cascode current source, is now much clearer. 6. Clearer and Simplified Study of Feedback. The feedback chapter has been rewritten to improve, simplify and clarify the presentation of this key subject. 7. Streamlined Presentation of Frequency Response. While keeping the treatment of frequency response all together, the chapter has been rewritten to streamline its flow, and simplify and clarify the presentation. 8. Updated Treatment of Output Stages and Power Amplifiers. Here, we have updated the material on MOS power transistors and added a new section on the increasingly important class-D switching power amplifier. 9. A More Contemporary Approach to Operational Amplifier Circuits. While maintaining coverage of some of the enduring features and subcircuits of the classical 741 op amp, its total coverage is somewhat reduced to make room for modern IC op amp design techniques. Preface  xxi  10. Better Organized and Modernized Coverage of Digital IC Design. Significant improvements have been made to the brief but comprehensive coverage of digital IC design in Part III. These include a better motivated study of CMOS logic circuits (Chapter 14) which now begins with logic gate circuits. The material on logic circuit technologies and design methodologies as well as the advanced topic of technology scaling and its implications have been moved to Chapter 15. This modularly structured chapter now deals with a selection of advanced and somewhat specialized topics. Since bipolar is hardly ever used in new digital design, coverage of ECL has been significantly reduced. Similarly, BiCMOS has become somewhat of a specialty topic and its coverage has been correspondingly reduced. Nevertheless, the complete material on both ECL and BiCMOS is now available on the book’s website. Finally, we have added a new section on image sensors to Chapter 16 (Memory Circuits). 11. Increased Emphasis on Integrated-Circuit Filters and Oscillators. A section on a popular approach to integrated-circuit filter design, namely, Transconductance-C filters, has been added to Chapter 17. To make room for this new material, the subsection on stagger-tuned amplifiers has been removed and placed in Appendix H, on the website. The cross-coupled LC oscillator, popular in IC design, has been added to Chapter 18. The section on precision diode circuits has been removed but is still made available on the website. 12. A Useful and Insightful Comparison of the MOSFET and the BJT. This is now included in Appendix G, available on the website. The Book’s Website A Companion Website for the book has been set up at www.oup.com/us/sedrasmith. Its content will change frequently to reflect new developments. The following material is available on the website: 1. Data sheets for hundreds of useful devices to help in laboratory experiments as well as in design projects. 2. Links to industrial and academic websites of interest. 3. A message center to communicate with the authors and with Oxford University Press. 4. Links to the student versions of both Cadence PSpice® and National Instruments Multisim™. 5. The input files for all the PSpice® and Multisim™ examples of Appendix B. 6. Step-by-step guidance to help with the simulation examples and the end-of-chapter problems identified with a SIM icon. 7. Bonus text material of specialized topics which are either not covered or covered briefly in the current edition of the textbook. These include: •  Junction Field-Effect Transistors (JFETs) •  Gallium Arsenide (GaAs) Devices and Circuits •  Transistor-Transistor Logic (TTL) Circuits •  Emitter-Coupled Logic (ECL) Circuits •  BiCMOS Circuits •  Precision Rectifier Circuits 8. Appendices for the Book: •  Appendix A: VLSI Fabrication Technology • Appendix B: SPICE Device Models and Design and Simulation Examples Using PSpice® and Multisim™ •  Appendix C: Two-Port Network Parameters •  Appendix D: Some Useful Network Theorems •  Appendix E: Single-Time-Constant Circuits •  Appendix F: s-domain Analysis: Poles, Zeros, and Bode Plots •  Appendix G: Comparison of the MOSFET and the BJT xxii Preface •  Appendix H: Design of Stagger-Tuned Amplifiers •  Appendix I: Bibliography •  Appendix L: Answers to Selected Problems Exercises and End-of-Chapter Problems Over 475 Exercises are integrated throughout the text. The answer to each exercise is given below the exercise so students can check their understanding of the material as they read. Solving these exercises should enable the reader to gauge his or her grasp of the preceding material. In addition, more than 1530 end-of-chapter Problems, 65% of which are new or revised in this edition, are provided. The problems are keyed to the individual chapter sections and their degree of difficulty is indicated by a rating system: difficult problems are marked with an asterisk (*); more difficult problems with two asterisks (**); and very difficult (and/or time consuming) problems with three asterisks (***). We must admit, however, that this classification is by no means exact. Our rating no doubt depended to some degree on our thinking (and mood!) at the time a particular problem was created. Answers to sample problems are given in Appendix L (on the website), so students have a checkpoint to tell if they are working out the problems correctly. Complete solutions for all exercises and problems are included in the Instructor’s Solutions Manual, which is available from the publisher to those instructors who adopt the book. As in the previous six editions, many examples are included. The examples, and indeed most of the problems and exercises, are based on real circuits and anticipate the applications encountered in designing real-life circuits. This edition continues the use of numbered solution steps in the figures for many examples, as an attempt to recreate the dynamics of the classroom. Course Organization The book contains sufficient material for a sequence of two single-semester courses, each of 40-50 lecture hours. The modular organization of the book provides considerable flexibility for course design. In the following, we suggest content for a sequence of two classical or standard courses. We also describe some variations on the content of these two courses and specify supplemental material for a possible third course. The First Course The first course is based on Part I of the book, that is, Chapters 1–7. It can be taught, most simply by starting at the beginning of Chapter 1 and concluding with the end of Chapter 7. However, as guidance to instructors who wish to follow a different order of presentation or a somewhat modified coverage, or to deal with situations where time might be constrained, we offer the following remarks: The core of the first course is the study of the two transistor types, Chapters 5 and 6, in whatever order the instructor wishes, and transistor amplifiers in Chapter 7. These three chapters must be covered in full. Another important part of the first course is the study of diodes (Chapter 4). Here, however, if time does not permit, some of the applications in the later part of the chapter can be skipped. We have found it highly motivational to cover op amps (Chapter 2) near the beginning of the course. This provides the students with the opportunity to work with a practical integrated circuit and to experiment with non-trivial circuits. Coverage of Chapter 1, at least of the amplifier sections, should prove helpful. Here the sections on signals can be either covered in class or assigned as reading material. Section 1.6 on frequency response is needed if the frequency-response of op-amp circuits is to be studied; otherwise this section can be delayed to the second course. Preface  xxiii  Finally, if the students have not taken a course on physical electronics, Chapter 3 needs to be covered. Otherwise, it can be used as review material or skipped altogether. The Second Course The main subject of the second course is integrated-circuit amplifiers and is based on Part II of the book, that is, Chapters 8-13. Here also, the course can be taught most simply by beginning with Chapter 8 and concluding with Chapter 13. However, this being a second course, considerable flexibility in coverage is possible to satisfy particular curriculum designs and/or to deal with time constraints. First, however, we note that the core material is presented in Chapters 8-11 and these four chapters must be covered, though not necessarily in their entirety. For instance, some of the sections near the end of a chapter and identified by the “advanced material” icon can be skipped, usually with no loss of continuity. Beyond the required chapters, (8-11), the instructor has many possibilities for the remainder of the course. These include one or both of the two remaining chapters in Part II, namely, Output Stages and Power Amplifier (Chapter 12), and Op-Amp Circuits (Chapter 13). Another possibility, is to include an introduction to digital integrated circuits by covering Chapter 14, and if time permits, selected topics of Chapters 15 and 16. Yet another possibility for the remainder of the second course is selected topics from the filters chapter (17) and/or the oscillators chapter (18). A Digitally Oriented First Course A digitally-oriented first course can include the following: Chapter 1 (without Section 1.6), Chapter 2, Chapter 3 (if the students have not had any exposure to physical electronics), Chapter 4 (perhaps without some of the later applications sections), Chapter 5, selected topics from Chapter 7 emphasizing the basics of the application of the MOSFET as an amplifier, Chapter 14, and selected topics from Chapters 15 and 16. Such a course would be particularly suited for Computer Engineering students. Supplemental Material/Third Course Depending on the selection of topics for the first and second courses, some material will remain and can be used for part of a third course or as supplemental material to support student design projects. These can include Chapter 12 (Output Stages and Power Amplifiers), Chapter 13 (Op-Amp Circuits), Chapter 17 (Filters) and Chapter 18 (Oscillators), which can be used to support a third course on analog circuits. These can also include Chapters 14, 15 and 16 which can be used for a portion of a senior-level course on digital IC design. The Accompanying Laboratory Courses in electronic circuits are usually accompanied by laboratory experiments. To support the laboratory component for courses using this book, Professor Vincent Gaudet of the University of Waterloo has, in collaboration with K.C. Smith, authored a laboratory manual. Laboratory Explorations, together with an Instructor’s Manual, is available from Oxford University Press. Another innovative laboratory instruction system, designed to accompany this book, has been recently developed. Specifically, Illuster Technologies Inc. has developed a digitally controlled lab platform, AELabs. The platform is realized on printed circuit boards using surface mount devices. A wide variety of circuits can be configured on this platform through a custom graphical user interface. This allows students to conduct many experiments relatively quickly. More information is available from Illuster (see link on the Companion Website). xxiv Preface An Outline for the Reader Part I, Devices and Basic Circuits, includes the most fundamental and essential topics for the study of electronic circuits. At the same time, it constitutes a complete package for a first course on the subject. Chapter 1. The book starts with an introduction to the basic concepts of electronics in Chapter 1. Signals, their frequency spectra, and their analog and digital forms are presented. Amplifiers are introduced as circuit building blocks and their various types and models are studied. This chapter also establishes some of the terminology and conventions used throughout the text. Chapter 2. Chapter 2 deals with operational amplifiers, their terminal characteristics, simple applications, and practical limitations. We chose to discuss the op amp as a circuit building block at this early stage simply because it is easy to deal with and because the student can experiment with op-amp circuits that perform nontrivial tasks with relative ease and with a sense of accomplishment. We have found this approach to be highly motivating to the student. We should point out, however, that part or all of this chapter can be skipped and studied at a later stage (for instance, in conjunction with Chapter 9, Chapter 11, and/or Chapter 13) with no loss of continuity. Chapter 3. Chapter 3 provides an overview of semiconductor concepts at a level sufficient for understanding the operation of diodes and transistors in later chapters. Coverage of this material is useful in particular for students who have had no prior exposure to device physics. Even those with such a background would find a review of Chapter 3 beneficial as a refresher. The instructor can choose to cover this material in class or assign it for outside reading. Chapter 4. The first electronic device, the diode, is studied in Chapter 4. The diode terminal characteristics, the circuit models that are used to represent it, and its circuit applications are presented. Depending on the time available in the course, some of the diode applications (e.g. Section 4.6) can be skipped. Also, the brief description of special diode types (Section 4.7) can be left for the student to read. Chapters 5 and 6. The foundation of electronic circuits is established by the study of the two transistor types in use today: the MOS transistor in Chapter 5 and the bipolar transistor in Chapter 6. These two chapters have been written to be completely independent of one another and thus can be studied in either order, as desired. Furthermore, the two chapters have the same structure, making it easier and faster to study the second device, as well as to draw comparisons between the two device types. Each of Chapters 5 and 6 begins with a study of the device structure and its physical operation, leading to a description of its terminal characteristics. Then, to allow the student to become very familiar with the operation of the transistor as a circuit element, a large number of examples are presented of dc circuits utilizing the device. The last section of each of Chapters 5 and 6 deals with second-order effects that are included for completeness, but that can be skipped if time does not permit detailed coverage. Chapter 7. The heart of a first course in electronics is the study of transistor amplifiers. Chapter 7 (new to this edition) presents a unified treatment of the subject. It begins with the basic principles that underlie the operation of a transistor, of either type, as an amplifier, and proceeds to present the important concepts of small-signal operation and modeling. This is followed by a study of the basic configurations of single-transistor amplifiers. After a presentation of dc biasing methods, the chapter concludes with practical examples of discrete-circuit amplifiers. The combined presentation emphasizes the unity of the basic principles while allowing for separate treatment of the two device types where this is warranted. Very importantly, we are able to compare the two devices and to draw conclusions about their unique areas of application. After the study of Part I, the reader will be fully prepared to study either integrated-circuit amplifiers in Part II, or digital integrated circuits in Part III. Part II, Integrated-Circuit Amplifiers, is devoted to the study of practical amplifier circuits that can be fabricated in the integrated-circuit (IC) form. Its six chapters constitute a coherent treatment of IC amplifier design and can thus serve as a second course in electronic circuits. MOS and Bipolar. Throughout Part II, both MOS and bipolar circuits are presented side-by-side. Because the MOSFET is by far the dominant device, its circuits are presented first. Bipolar circuits are discussed to the same depth but occasionally more briefly. Preface  xxv  Chapter 8. Beginning with a brief introduction to the philosophy of IC design, Chapter 8 presents the basic circuit building blocks that are used in the design of IC amplifiers. These include current mirrors, current sources, gain cells, and cascode amplifiers. Chapter 9. The most important IC building block, the differential pair, is the main topic of Chapter 9. The last section of Chapter 9 is devoted to the study of multistage amplifiers. Chapter 10. Chapter 10 presents a comprehensive treatment of the important subject of amplifier frequency response. Here, Sections 10.1, 10.2, and 10.3 contain essential material; Section 10.4 provides an in-depth treatment of very useful new tools; and Sections 10.5 to 10.8 present the frequency response analysis of a variety of amplifier configurations that can be studied as and when needed. A selection of the latter sections can be made depending on the time available and the instructor’s preference. Chapter 11. The fourth of the essential topics of Part II, feedback, is the subject of Chapter 11. Both the theory of negative feedback and its application in the design of practical feedback amplifiers are presented. We also discuss the stability problem in feedback amplifiers and treat frequency compensation in some detail. Chapter 12. In Chapter 12 we switch gears from dealing with small-signal amplifiers to those that are required to handle large signals and large amounts of power. Here we study the different amplifier classes—A, B, and AB—and their realization in bipolar and CMOS technologies. We also consider power BJTs and power MOSFETs, and study representative IC power amplifiers. A brief study of the increasingly popular Class D amplifier is also presented. Depending on the availability of time, some of the later sections can be skipped in a first reading. Chapter 13. Finally, Chapter 13 brings together all the topics of Part II in an important application; namely, the design of operational amplifier circuits. We study both CMOS and bipolar op amps. In the latter category, besides the classical and still timely 741 circuit, we present modern techniques for the design of low-voltage op amps (Section 13.4). Part III, Digital Integrated Circuits, provides a brief but nonetheless comprehensive and sufficiently detailed study of digital IC design. Our treatment is almost self-contained, requiring for the most part only a thorough understanding of the MOSFET material presented in Chapter 5. Thus, Part III can be studied right after Chapter 5. The only exceptions to this are the last section in Chapter 15 which requires knowledge of the BJT (Chapter 6). Also, knowledge of the MOSFET internal capacitances (Section 10.2.2) will be needed. Chapter 14. Chapter 14 is the foundation of Part III. It begins with the motivating topic of CMOS logic-gate circuits. Then, following a detailed study of digital logic inverters, we concentrate on the CMOS inverter; its static and dynamic characteristics and its design. Transistor sizing and power dissipation round out the topics of Chapter 14. The material covered in this chapter is the minimum needed to learn something meaningful about digital circuits. Chapter 15. Chapter 15 has a modular structure and presents six topics of somewhat advanced nature. It begins with a presentation of Moore’s law and the technology scaling that has made the multi-billion-transistor chip possible. This is followed by an overview of digital IC technologies, and the design methodologies that make the design of super-complex digital ICs possible. Four different logic-circuit types are then presented. Only the last of these includes bipolar transistors. Chapter 16. Digital circuits can be broadly divided into logic and memory circuits. The latter is the subject of Chapter 16. Part IV, Filters and Oscillators, is intentionally oriented toward applications and systems. The two topics illustrate powerfully and dramatically the application of both negative and positive feedback. Chapter 17. Chapter 17 deals with the design of filters, which are important building blocks of communication and instrumentation systems. A comprehensive, design-oriented treatment of the subject is presented. The material provided should allow the reader to perform a complete filter design, starting from specification and ending with a complete circuit realization. A wealth of design tables is included. Chapter 18. Chapter 18 deals with circuits for the generation of signals with a variety of waveforms: sinusoidal, square, and triangular. We also present circuits for the nonlinear shaping of waveforms. xxvi Preface Appendices. The twelve appendices contain much useful background and supplementary material. We wish to draw the reader’s attention in particular to the first two: Appendix A provides a concise introduction to the important topic of IC fabrication technology including IC layout. Appendix B provides SPICE device models as well as a large number of design and simulation examples in PSpice® and Multisim™. The examples are keyed to the book chapters. These Appendices and a great deal more material on these simulation examples can be found on the Companion Website. Ancillaries A complete set of ancillary materials is available with this text to support your course. For the Instructor The Ancillary Resource Center (ARC) at www.oup-arc.com/sedrasmith is a convenient destination for all the instructor resources that accompany Microelectronic Circuits. Accessed online through individual user accounts, the ARC provides instructors with access to up-to-date ancillaries at any time while guaranteeing the security of grade-significant resources. The ARC replaces the Instructor’s Resource CD that accompanied the sixth edition. On the ARC, you will find: •  An electronic version of the Instructor’s Solutions Manual. • PowerPoint-based figure slides that feature all the images and summary tables from the text, with their captions, so they can easily be displayed and explained in class. • Detailed instructor’s support for the SPICE circuit simulations in Multisim™ and PSpice®. The Instructor’s Solutions Manual (ISBN 978-0-19-933915-0), written by Adel Sedra, contains detailed solutions to all in-text exercises and end-of-chapter problems found in Microelectronic Circuits. The Instructor’s Solutions Manual for Laboratory Explorations to Accompany Microelectronic Circuits (ISBN 978-0-19-933926-6) contains detailed solutions to all the exercises and problems found in this student’s laboratory guide. For the Student and Instructor A Companion Website at www.oup.com/us/sedrasmith features permanently cached versions of device datasheets, so students can design their own circuits in class. The website also contains SPICE circuit simulation examples and lessons. Bonus text topics and the Appendices are also featured on the website. The Laboratory Explorations to Accompany Microelectronic Circuits (ISBN 978-0-19-933925-9) invites students to explore the realm of real-world engineering through practical, hands-on experiments. Keyed to sections in the text and taking a “learn-by-doing” approach, it presents labs that focus on the development of practical engineering skills and design practices. Acknowledgments Many of the changes in this seventh edition were made in response to feedback received from instructors who adopted the sixth edition. We are grateful to all those who took the time to write to us. In addition, many of the reviewers provided detailed commentary on the sixth edition and suggested a number of the changes that we have incorporated in this edition. They are listed later; to all of them, we extend our sincere thanks. Adel Sedra is also grateful for the feedback received from the students who have taken his electronics courses over the past number of years at the University of Waterloo. Preface  xxvii  A number of individuals made significant contributions to this edition. Vincent Gaudet of the University of Waterloo contributed to Part III as well as co-authoring the laboratory manual. WaiTung Ng of the University of Toronto contributed to Chapter 12 and updated Appendix A (of which he is the original author). Muhammad Faisal of the University of Michigan updated Appendix B, which he helped create for the sixth edition; helped in obtaining the cover photo, and has over a number of years been the source of many good ideas. Olivier Trescases and his students at the University of Toronto pioneered the laboratory system described elsewhere in the Preface. Jennifer Rodrigues typed all the revisions, as she did for a number of the previous editions, with tremendous skill and good humour. Chris Schroeder was of great assistance to Adel Sedra with local logistics. Laura Fujino assisted in many ways and in particular with the “Expand-Your-Perspective” notes. To all of these friends and colleagues we say thank you. Over the recent years we have benefited greatly from discussions with a number of colleagues and friends. In particular we are very grateful to the following: James Barby, University of Waterloo; David Nairn, University of Waterloo; Anthony Chan Carusone, University of Toronto; David Johns, University of Toronto; Ken Martin, University of Toronto; Khoman Phang, University of Toronto; Gordon Roberts, McGill University; Ali Sheikholeslami, University of Toronto; and Amir Yazdani, Ryerson University. The cover photograph shows a 3D IC system, which demonstrates the concept of wireless power delivery and communication through multiple layers of CMOS chips. The communication circuits were demonstrated in an IBM 45 nm SOI CMOS process. This technology is designed to serve a multi-Gb/s interconnect between cores spread across several IC layers for high-performance processors. We are grateful to Professor David Wentzloff, Director of the Wireless Integrated Circuits Group at the University of Michigan, who allowed us to use this image, and to Muhammad Faisal, Founder of Movellus Circuits Incorporated, who edited the image. A large number of people at Oxford University Press contributed to the development of this edition and its various ancillaries. We would like to specifically mention Marketing Manager David Jurman, Marketing Director Frank Mortimer, Higher Ed Sales Director Bill Marting, Copywriter Kristin Maffei, Art Director Michele Laseau, Production Manager Lisa Grzan, Team Leader Amy Whitmer, and Senior Production Editor Jane Lee. We wish to extend special thanks to our Publisher at Oxford University Press, John Challice, and the Editorial Director, Patrick Lynch. Both have always shown great interest in this book and have provided considerable guidance and support throughout the process of preparing this edition. The Senior Acquisitions Editor, Nancy Blaine, and Associate Editor, Christine Mahon, have done a truly outstanding job. It has been a pleasure to work with both of them, both as professionals and as highly thoughtful individuals; we owe them much gratitude. On the production side, Barbara Mathieu, Senior Production Editor, has been superb: her attention to detail and emphasis on quality is without par. Finally, we wish to thank our families for their support and understanding, and to thank all the students and instructors who have valued this book throughout its history. Adel S. Sedra Kenneth C. (KC) Smith Waterloo, Ontario, Canada August 2014 xxviii Preface Reviewers of Seventh Edition Junseok Chae, Arizona State University, Tempe, AZ Liang Dong, Baylor University, Waco, TX Muhammad Faisal, University of Michigan, Ann Arbor, MI Patrick Fay, University of Notre Dame, Notre Dame, IN Vincent Gaudet, University of Waterloo, Waterloo, Canada Elmer A Grubbs, Northern Arizona University, Flagstaff, AZ Serhiy Levkov, New Jersey Institute of Technology, Newark, NJ Leda Lunardi, North Carolina State University, Raleigh, NC Phyllis R. Nelson, California State Polytechnic University, Pomona, CA Robert W. Newcomb, University of Maryland, College Park, MD Toshikazu Nishida, University of Florida, Gainesville, FL Matthew Swabey, Purdue University, West Lafayette, IN Khalid Hasan Tantawi, University of Alabama, Huntsville, AL Farid M. Tranjan, University of North Carolina, Charlotte, NC Mustapha C.E. Yagoub, University of Ottawa, Ottawa, Canada Justin Jackson, Weber State University, Ogden, UT John Mankowski, Texas Tech University, Lubbock, TX Chris Mi, University of Michigan, Dearborn, MI Reviewers of Prior Editions Maurice Aburdene, Bucknell University, Lewisburg, PA Michael Bartz, University of Memphis, TN Elizabeth Brauer, Northern Arizona University, Flagstaff, AZ Martin Brooke, Duke University, Durham, NC Patrick L. Chapman, University of Illinois, Urbana– Champaign, IL Yun Chiu, University of Illinois, Urbana–Champaign, IL Roy H. Cornely, New Jersey Institute of Technology, Newark, NJ Norman Cox, Missouri University of Science and Technology, Rolla, MO Dale L. Critchlow, University of Vermont, Burlingon, VT Robert Bruce Darling, University of Washington, Seattle, WA Artice Davis, San Jose State University, CA John Davis, University of Texas, Austin, TX Christopher DeMarco, University of Wisconsin, Madison, WI Robert Engelken, Arkansas State University, Jonesboro, AR Ethan Farquhar, University of Tennessee, Knoxville, TN Eby G. Friedman, University of Rochester, NY Paul M. Furth, New Mexico State University, Las Cruces, NM Rhett T. George, Jr., Duke University, Durham, NC Roobik Gharabagi, St. Louis University, MO George Giakos, University of Akron, OH John Gilmer, Wilkes University, Wilkes-Barre, PA Michael Green, University of California, Irvine, CA Steven de Haas, California State University, Sacramento, CA Anas Hamoui, McGill University, Montreal, Canada Reza Hashemian, Northern Illinois University, DeKalb, IL William Harrell, Clemson University, SC Reid Harrison, University of Utah, Salt Lake City, UT Ward J. Helms, University of Washington, Seattle, WA Richard Hornsey, York University, Toronto, Canada Timothy Horiuchi, University of Maryland, College Park, MD Hsiung Hsu, The Ohio State University, Columbus, OH Robert Irvine, California State Polytechnic University, Pomona, CA Mohammed Ismail, The Ohio State University, Columbus, OH Paul Israelsen, Utah State University, Logan UT Steve Jantzi, Broadcom, CA Zhenhua Jiang, University of Miami, FL Marian Kazimierczuk, Wright State University, Dayton, OH John Khoury, Columbia University, New York, NY Jacob B. Khurgin, The Johns Hopkins University, Baltimore, MD Seongsin M. Kim, University of Alabama, Tuscaloosa, AL Roger King, University of Toledo, OH Clark Kinnaird, Southern Methodist University, Dallas, TX Robert J. Krueger, University of Wisconsin, Milwaukee, WI Joy Laskar, Georgia Institute of Technology, Atlanta, GA Tsu-Jae King Liu, University of California, Berkeley, CA Yicheng Lu, Rutgers University, Piscataway, NJ David Luke, University of New Brunswick, Fredericton, Canada Thomas Matthews, California State University, Sacramento, CA Un-Ku Moon, Oregon State University, Corvallis, OR Bahram Nabet, Drexel University, Philadelphia, PA Dipankar Nagchoudhuri, Indian Institute of Technology, Delhi, India David Nairn, University of Waterloo, Waterloo, Canada Joseph H. Nevin, University of Cincinnati, OH Ken Noren, University of Idaho, Moscow, ID Brita Olson, California Polytechnic University, Pomona, CA Martin Peckerar, University of Maryland, College Park, MD Khoman Phang, University of Toronto, Canada Mahmudur Rahman, Santa Clara University, CA Rabin Raut, Concordia University, Montreal, Canada John A. Ringo, Washington State University, Pullman, WA Zvi S. Roth, Florida Atlantic University, Boca Raton, FL Mulukutla Sarma, Northeastern University, Boston, MA John Scalzo, Louisiana State University, Baton Rouge, LA Norman Scheinberg, City College, New York, NY Pierre Schmidt, Florida International University, Miami, FL Richard Schreier, Analog Devices, Toronto, Canada Dipankar Sengupta, Royal Melbourne Institute of Technology, Australia Ali Sheikholeslami, University of Toronto, Canada Kuang Sheng, Rutgers University, Piscataway, NJ Michael L. Simpson, University of Tennessee, Knoxville, TN Karl A. Spuhl, Washington University in St. Louis, MO Charles Sullivan, Dartmouth College, Hanover, NH Andrew Szeto, San Diego State University, CA Joel Therrien, University of Massachusetts, Lowell, MA Len Trombetta, University of Houston, TX Daniel van der Weide, University of Delaware, Newark, DE Gregory M. Wierzba, Michigan State University, East Lansing, MI Donna Yu, North Carolina State University, Raleigh, NC Jiann-Shiun Yuan, University of Central Florida, Orlando, FL Sandra Yost, University of Detroit, Mercy, MI Alex Zaslavsky, Brown University, Providence, RI Jianhua (David) Zhang, University of Illinois, Urbana– Champaign, IL Microelectronic Circuits PART I Devices and Basic Circuits CHAPTER 1 Signals and Amplifiers 4 CHAPTER 2 Operational Amplifiers 58 CHAPTER 3 Semiconductors 134 CHAPTER 4 Diodes 174 CHAPTER 5 MOS Field-Effect Transistors (MOSFETs) 246 CHAPTER 6 Bipolar Junction Transistors (BJTs) 304 CHAPTER 7 Transistor Amplifiers 366 P art I, Devices and Basic Circuits, includes the most fundamental and essential topics for the study of electronic circuits. At the same time, it constitutes a complete package for a first course on the subject. The heart of Part I is the study of the three basic semiconductor devices: the diode (Chapter 4), the MOS transistor (Chapter 5), and the bipolar transistor (Chapter 6). In each case, we study the device operation, its characterization, and its basic circuit applications. Chapter 7 then follows with a study of the most fundamental application of the two transistor types; namely, their use in amplifier design. This side-by-side study of MOSFET and BJT amplifiers allows us to see similarities between these amplifiers and to compare them, which in turn highlights the distinct areas of applicability of each, as well as showing the unity of the basic principles that underlie the use of transistors as amplifiers. For those who have not had a prior course on device physics, Chapter 3 provides an overview of semiconductor concepts at a level sufficient for the study of electronic circuits. A review of Chapter 3 should prove useful even for those with prior knowledge of semiconductors. Since the purpose of electronic circuits is the processing of signals, it is essential to understand signals, their characterization in the time and frequency domains, and their analog and digital representations. The basis for such understanding is provided in Chapter 1, which also introduces the most common signal-processing function, amplification, and the characterization and types of amplifiers. Besides diodes and transistors, the basic electronic devices, the op amp is studied in Part I. Although not an electronic device in the most fundamental sense, the op amp is commercially available as an integrated circuit (IC) package and has well-defined terminal characteristics. Thus, even though the op amp’s internal circuit is complex, typically incorporating 20 or more transistors, its almost-ideal terminal behavior makes it possible to treat the op amp as a circuit element and to use it in the design of powerful circuits, as we do in Chapter 2, without any knowledge of its internal construction. We should mention, however, that the study of op amps can be delayed until a later point, and Chapter 2 can be skipped with no loss of continuity. The foundation of this book, and of any electronics course, is the study of the two transistor types in use today: the MOS transistor in Chapter 5 and the bipolar transistor in Chapter 6. These two chapters have been written to be completely independent of each other and thus can be studied in either order, as desired. After the study of Part I, the reader will be fully prepared to undertake the study of either integrated-circuit amplifiers in Part II or digital integrated circuits in Part III. 3 CHAPTER 1 Signals and Amplifiers Introduction 5 1.1 Signals 6 1.2 Frequency Spectrum of Signals 9 1.3 Analog and Digital Signals 12 1.4 Amplifiers 15 1.5 Circuit Models for Amplifiers 23 1.6 Frequency Response of Amplifiers 33 Summary 44 Problems 45 IN THIS CHAPTER YOU WILL LEARN 1. That electronic circuits process signals, and thus understanding electrical signals is essential to appreciating the material in this book. 2. The The´ venin and Norton representations of signal sources. 3. The representation of a signal as the sum of sine waves. 4. The analog and digital representations of a signal. 5. The most basic and pervasive signal-processing function: signal amplification, and correspondingly, the signal amplifier. 6. How amplifiers are characterized (modeled) as circuit building blocks independent of their internal circuitry. 7. How the frequency response of an amplifier is measured, and how it is calculated, especially in the simple but common case of a single-time-constant (STC) type response. Introduction The subject of this book is modern electronics, a field that has come to be known as microelectronics. Microelectronics refers to the integrated-circuit (IC) technology that at the time of this writing is capable of producing circuits that contain billions of components in a small piece of silicon (known as a silicon chip) whose area is on the order of 100 mm2. One such microelectronic circuit, for example, is a complete digital computer, which accordingly is known as a microcomputer or, more generally, a microprocessor. The microelectronic circuits you will learn to design in this book are used in almost every device we encounter in our daily lives: in the appliances we use in our homes; in the vehicles and transportation systems we use to travel; in the cell phones we use to communicate; in the medical equipment we need to care for our health; in the computers we use to do our work; and in the audio and video systems, the radio and TV sets, and the multitude of other digital devices we use to entertain ourselves. Indeed, it is difficult to conceive of modern life without microelectronic circuits. In this book we shall study electronic devices that can be used singly (in the design of discrete circuits) or as components of an integrated-circuit (IC) chip. We shall study the design and analysis of interconnections of these devices, which form discrete and integrated 5 6 Chapter 1 Signals and Amplifiers circuits of varying complexity and perform a wide variety of functions. We shall also learn about available IC chips and their application in the design of electronic systems. The purpose of this first chapter is to introduce some basic concepts and terminology. In particular, we shall learn about signals and about one of the most important signal-processing functions electronic circuits are designed to perform, namely, signal amplification. We shall then look at circuit representations or models for linear amplifiers. These models will be employed in subsequent chapters in the design and analysis of actual amplifier circuits. In addition to motivating the study of electronics, this chapter serves as a bridge between the study of linear circuits and that of the subject of this book: the design and analysis of electronic circuits. 1.1 Signals Signals contain information about a variety of things and activities in our physical world. Examples abound: Information about the weather is contained in signals that represent the air temperature, pressure, wind speed, etc. The voice of a radio announcer reading the news into a microphone provides an acoustic signal that contains information about world affairs. To monitor the status of a nuclear reactor, instruments are used to measure a multitude of relevant parameters, each instrument producing a signal. To extract required information from a set of signals, the observer (be it a human or a machine) invariably needs to process the signals in some predetermined manner. This signal processing is usually most conveniently performed by electronic systems. For this to be possible, however, the signal must first be converted into an electrical signal, that is, a voltage or a current. This process is accomplished by devices known as transducers. A variety of transducers exist, each suitable for one of the various forms of physical signals. For instance, the sound waves generated by a human can be converted into electrical signals by using a microphone, which is in effect a pressure transducer. It is not our purpose here to study transducers; rather, we shall assume that the signals of interest already exist in the electrical domain and represent them by one of the two equivalent forms shown in Fig. 1.1. In Fig. 1.1(a) the signal is represented by a voltage source vs(t) having a source resistance Rs. In the alternate representation of Fig. 1.1(b) the signal is represented by a current source is(t) having a source resistance Rs. Although the two representations are equivalent, that in Fig. 1.1(a) (known as the The´venin form) is preferred when Rs is low. The representation of Fig. 1.1(b) (known as the Norton form) is preferred when Rs is high. The reader will come to appreciate this point later in this chapter when we study the different types of amplifiers. For the time being, it is important to be familiar with The´venin’s and Norton’s theorems (for a Rs vs(t) ϩ Ϫ (a) is(t) Rs (b) Figure 1.1 Two alternative representations of a signal source: (a) the The´venin form; (b) the Norton form. 1.1 Signals 7 brief review, see Appendix D) and to note that for the two representations in Fig. 1.1 to be equivalent, their parameters are related by vs(t) = Rsis(t) Example 1.1 The output resistance of a signal source, although inevitable, is an imperfection that limits the ability of the source to deliver its full signal strength to a load. To see this point more clearly, consider the signal source when connected to a load resistance RL as shown in Fig. 1.2. For the case in which the source is represented by its The´venin equivalent form, find the voltage vo that appears across RL, and hence the condition that Rs must satisfy for vo to be close to the value of vs. Repeat for the Norton-represented source; in this case finding the current io that flows through RL and hence the condition that Rs must satisfy for io to be close to the value of is. Rs vs ϩ Ϫ (a) Figure 1.2 Circuits for Example 1.1. ϩ RL vo is Ϫ io Rs RL (b) Solution For the The´venin-represented signal source shown in Fig. 1.2(a), the output voltage vo that appears across the load resistance RL can be found from the ratio of the voltage divider formed by Rs and RL, vo = vs RL RL + Rs From this equation we see that for vo vs the source resistance Rs must be much lower than the load resistance RL, Rs RL Thus, for a source represented by its The´venin equivalent, ideally Rs = 0, and as Rs is increased, relative to the load resistance RL with which this source is intended to operate, the voltage vo that appears across the load becomes smaller, not a desirable outcome. 8 Chapter 1 Signals and Amplifiers Example 1.1 continued Next, we consider the Norton-represented signal source in Fig. 1.2(b). To obtain the current io that flows through the load resistance RL, we utilize the ratio of the current divider formed by Rs and RL, io = is Rs Rs + RL From this relationship we see that for io is the source resistance Rs must be much larger than RL, Rs RL Thus for a signal source represented by its Norton equivalent, ideally Rs = ∞, and as Rs is reduced, relative to the load resistance RL with which this source is intended to operate, the current io that flows through the load becomes smaller, not a desirable outcome. Finally, we note that although circuit designers cannot usually do much about the value of Rs, they may have to devise a circuit solution that minimizes or eliminates the loss of signal strength that results when the source is connected to the load. EXERCISES 1.1 For the signal-source representations shown in Figs. 1.1(a) and 1.1(b), what are the open-circuit output voltages that would be observed? If, for each, the output terminals are short-circuited (i.e., wired together), what current would flow? For the representations to be equivalent, what must the relationship be between vs, is, and Rs? Ans. For (a), voc = vs(t); for (b), voc = Rsis(t); for (a), isc = vs(t)/Rs; for (b), isc = is(t); for equivalency, vs(t) = Rsis(t) 1.2 A signal source has an open-circuit voltage of 10 mV and a short-circuit current of 10 μA. What is the source resistance? Ans. 1 k 1.3 A signal source that is most conveniently represented by its The´venin equivalent has vs = 10 mV and Rs = 1 k . If the source feeds a load resistance RL, find the voltage vo that appears across the load for RL = 100 k , 10 k , 1 k , and 100 . Also, find the lowest permissible value of RL for which the output voltage is at least 80% of the source voltage. Ans. 9.9 mV; 9.1 mV; 5 mV; 0.9 mV; 4 k 1.4 A signal source that is most conveniently represented by its Norton equivalent form has is = 10 μA and Rs = 100 k . If the source feeds a load resistance RL, find the current io that flows through the load for RL = 1 k , 10 k , 100 k , and 1 M . Also, find the largest permissible value of RL for which the load current is at least 80% of the source current. Ans. 9.9 μA; 9.1 μA; 5 μA; 0.9 μA; 25 k 1.2 Frequency Spectrum of Signals 9 Figure 1.3 An arbitrary voltage signal vs(t). From the discussion above, it should be apparent that a signal is a time-varying quantity that can be represented by a graph such as that shown in Fig. 1.3. In fact, the information content of the signal is represented by the changes in its magnitude as time progresses; that is, the information is contained in the “wiggles” in the signal waveform. In general, such waveforms are difficult to characterize mathematically. In other words, it is not easy to describe succinctly an arbitrary-looking waveform such as that of Fig. 1.3. Of course, such a description is of great importance for the purpose of designing appropriate signal-processing circuits that perform desired functions on the given signal. An effective approach to signal characterization is studied in the next section. 1.2 Frequency Spectrum of Signals An extremely useful characterization of a signal, and for that matter of any arbitrary function of time, is in terms of its frequency spectrum. Such a description of signals is obtained through the mathematical tools of Fourier series and Fourier transform.1 We are not interested here in the details of these transformations; suffice it to say that they provide the means for representing a voltage signal vs(t) or a current signal is(t) as the sum of sine-wave signals of different frequencies and amplitudes. This makes the sine wave a very important signal in the analysis, design, and testing of electronic circuits. Therefore, we shall briefly review the properties of the sinusoid. Figure 1.4 shows a sine-wave voltage signal va(t), va(t) = Va sin ωt (1.1) where Va denotes the peak value or amplitude in volts and ω denotes the angular frequency in radians per second; that is, ω = 2πf rad/s, where f is the frequency in hertz, f = 1/T Hz, and T is the period in seconds. The sine-wave signal is completely characterized by its peak value Va, its frequency ω, and its phase with respect to an arbitrary reference time. In the case depicted in Fig. 1.4, the time 1The reader who has not yet studied these topics should not be alarmed. No detailed application of this material will be made until Chapter 10. Nevertheless, a general understanding of Section 1.2 should be very helpful in studying early parts of this book. 10 Chapter 1 Signals and Amplifiers Figure 1.4 Sine-wave voltage signal of amplitude Va and frequency f = 1/T Hz. The angular frequency ω = 2π f rad/s. Figure 1.5 A symmetrical square-wave signal of amplitude V . origin has been chosen so that the phase angle is 0. It should be mentioned that it is common to express the amplitude of a sine-wave signal√in terms of its root-mean-square (rms) value, which is equal t√o the peak value divided by 2. Thus the rms value of the sinusoid va(t) of Fig. 1.4 is Va/ 2. For instance, when we speak of the w√all power supply in our homes as being 120 V, we mean that it has a sine waveform of 120 2 volts peak value. Returning now to the representation of signals as the sum of sinusoids, we note that the Fourier series is utilized to accomplish this task for the special case of a signal that is a periodic function of time. On the other hand, the Fourier transform is more general and can be used to obtain the frequency spectrum of a signal whose waveform is an arbitrary function of time. The Fourier series allows us to express a given periodic function of time as the sum of an infinite number of sinusoids whose frequencies are harmonically related. For instance, the symmetrical square-wave signal in Fig. 1.5 can be expressed as 4V 1 1 v(t) = π (sin ω0t + 3 sin 3ω0t + 5 sin 5ω0t + · · · ) (1.2) where V is the amplitude of the square wave and ω0 = 2π /T (T is the period of the square wave) is called the fundamental frequency. Note that because the amplitudes of the harmonics progressively decrease, the infinite series can be truncated, with the truncated series providing an approximation to the square waveform. The sinusoidal components in the series of Eq. (1.2) constitute the frequency spectrum of the square-wave signal. Such a spectrum can be graphically represented as in Fig. 1.6, where the horizontal axis represents the angular frequency ω in radians per second. 1.2 Frequency Spectrum of Signals 11 Figure 1.6 The frequency spectrum (also known as the line spectrum) of the periodic square wave of Fig. 1.5. Figure 1.7 The frequency spectrum of an arbitrary waveform such as that in Fig. 1.3. The Fourier transform can be applied to a nonperiodic function of time, such as that depicted in Fig. 1.3, and provides its frequency spectrum as a continuous function of frequency, as indicated in Fig. 1.7. Unlike the case of periodic signals, where the spectrum consists of discrete frequencies (at ω0 and its harmonics), the spectrum of a nonperiodic signal contains in general all possible frequencies. Nevertheless, the essential parts of the spectra of practical signals are usually confined to relatively short segments of the frequency (ω) axis—an observation that is very useful in the processing of such signals. For instance, the spectrum of audible sounds such as speech and music extends from about 20 Hz to about 20 kHz—a frequency range known as the audio band. Here we should note that although some musical tones have frequencies above 20 kHz, the human ear is incapable of hearing frequencies that are much above 20 kHz. As another example, analog video signals have their spectra in the range of 0 MHz to 4.5 MHz. We conclude this section by noting that a signal can be represented either by the manner in which its waveform varies with time, as for the voltage signal va(t) shown in Fig. 1.3, or in terms of its frequency spectrum, as in Fig. 1.7. The two alternative representations are known as the time-domain representation and the frequency-domain representation, respectively. The frequency-domain representation of va(t) will be denoted by the symbol Va(ω). 12 Chapter 1 Signals and Amplifiers EXERCISES 1.5 Find the frequencies f and ω of a sine-wave signal with a period of 1 ms. Ans. f = 1000 Hz; ω = 2π × 103 rad/s 1.6 What is the period T of sine waveforms characterized by frequencies of (a) f = 60 Hz? (b) f = 10−3 Hz? (c) f = 1 MHz? Ans. 16.7 ms; 1000 s; 1 μs 1.7 The UHF (ultra high frequency) television broadcast band begins with channel 14 and extends from 470 MHz to 806 MHz. If 6 MHz is allocated for each channel, how many channels can this band accommodate? Ans. 56; channels 14 to 69 1.8 When the square-wave signal of Fig. 1.5, whose Fourier series is given in Eq. (1.2), is applied to a resistor, the total power dissipated may be calculated directly using the relationship P = 1/T T 0 (v 2 /R) dt or indirectly by summing the contribution of each of the harmonic components, that is, P = P1 + P3 + P5 + . . . , which may be found directly from rms values. Verify that the two approaches are equivalent. What fraction of the energy of a square wave is in its fundamental? In its first five harmonics? In its first seven? First nine? In what number of harmonics is 90% of the energy? (Note that in counting harmonics, the fundamental at ω0 is the first, the one at 2ω0 is the second, etc.) Ans. 0.81; 0.93; 0.95; 0.96; 3 1.3 Analog and Digital Signals The voltage signal depicted in Fig. 1.3 is called an analog signal. The name derives from the fact that such a signal is analogous to the physical signal that it represents. The magnitude of an analog signal can take on any value; that is, the amplitude of an analog signal exhibits a continuous variation over its range of activity. The vast majority of signals in the world around us are analog. Electronic circuits that process such signals are known as analog circuits. A variety of analog circuits will be studied in this book. An alternative form of signal representation is that of a sequence of numbers, each number representing the signal magnitude at an instant of time. The resulting signal is called a digital signal. To see how a signal can be represented in this form—that is, how signals can be converted from analog to digital form—consider Fig. 1.8(a). Here the curve represents a voltage signal, identical to that in Fig. 1.3. At equal intervals along the time axis, we have marked the time instants t0, t1, t2, and so on. At each of these time instants, the magnitude of the signal is measured, a process known as sampling. Figure 1.8(b) shows a representation of the signal of Fig. 1.8(a) in terms of its samples. The signal of Fig. 1.8(b) is defined only at the sampling instants; it no longer is a continuous function of time; rather, it is a discrete-time signal. However, since the magnitude of each sample can take any value in a continuous range, the signal in Fig. 1.8(b) is still an analog signal. Now if we represent the magnitude of each of the signal samples in Fig. 1.8(b) by a number having a finite number of digits, then the signal amplitude will no longer be continuous; rather, 1.3 Analog and Digital Signals 13 (a) Figure 1.8 Sampling the continuous-time analog signal in (a) results in the discrete-time signal in (b). it is said to be quantized, discretized, or digitized. The resulting digital signal then is simply a sequence of numbers that represent the magnitudes of the successive signal samples. The choice of number system to represent the signal samples affects the type of digital signal produced and has a profound effect on the complexity of the digital circuits required to process the signals. It turns out that the binary number system results in the simplest possible digital signals and circuits. In a binary system, each digit in the number takes on one of only two possible values, denoted 0 and 1. Correspondingly, the digital signals in binary systems need have only two voltage levels, which can be labeled low and high. As an example, in some of the digital circuits studied in this book, the levels are 0 V and +5 V. Figure 1.9 shows the time variation of such a digital signal. Observe that the waveform is a pulse train with 0 V representing a 0 signal, or logic 0, and +5 V representing logic 1. If we use N binary digits (bits) to represent each sample of the analog signal, then the digitized sample value can be expressed as D = b020 + b121 + b222 + · · · + bN−12N−1 (1.3) where b0, b1, . . . , bN−1, denote the N bits and have values of 0 or 1. Here bit b0 is the least significant bit (LSB), and bit bN−1 is the most significant bit (MSB). Conventionally, this binary number is written as bN−1 bN−2 . . . b0. We observe that such a representation quantizes the analog sample into one of 2N levels. Obviously the greater the number of bits (i.e., the larger the N), the closer the digital word D approximates the magnitude of the analog sample. That is, increasing the number of bits reduces the quantization error and increases the resolution of the 14 Chapter 1 Signals and Amplifiers v (t) (V) ϩ5 0 Logic values 1 0 1 1 0 1 0 0 Time, t Figure 1.9 Variation of a particular binary digital signal with time. Analog input ϩ vA Ϫ A/D converter b0 b1 bNϪ1 Digital output Figure 1.10 Block-diagram representation of the analog-to-digital converter (ADC). analog-to-digital conversion. This improvement is, however, usually obtained at the expense of more complex and hence more costly circuit implementations. It is not our purpose here to delve into this topic any deeper; we merely want the reader to appreciate the nature of analog and digital signals. Nevertheless, it is an opportune time to introduce a very important circuit building block of modern electronic systems: the analog-to-digital converter (A/D or ADC) shown in block form in Fig. 1.10. The ADC accepts at its input the samples of an analog signal and provides for each input sample the corresponding N-bit digital representation (according to Eq. 1.3) at its N output terminals. Thus although the voltage at the input might be, say, 6.51 V, at each of the output terminals (say, at the ith terminal), the voltage will be either low (0 V) or high (5 V) if bi is supposed to be 0 or 1, respectively. The dual circuit of the ADC is the digital-to-analog converter (D/A or DAC). It converts an N-bit digital input to an analog output voltage. Once the signal is in digital form, it can be processed using digital circuits. Of course digital circuits can deal also with signals that do not have an analog origin, such as the signals that represent the various instructions of a digital computer. Since digital circuits deal exclusively with binary signals, their design is simpler than that of analog circuits. Furthermore, digital systems can be designed using a relatively few different kinds of digital circuit blocks. However, a large number (e.g., hundreds of thousands or even millions) of each of these blocks are usually needed. Thus the design of digital circuits poses its own set of challenges to the designer but provides reliable and economic implementations of a great variety of signal-processing functions, many of which are not possible with analog circuits. At the present time, more and more of the signal-processing functions are being performed digitally. Examples around us abound: from the digital watch and the calculator to digital audio systems, digital cameras, and digital television. Moreover, some long-standing 1.4 Amplifiers 15 analog systems such as the telephone communication system are now almost entirely digital. And we should not forget the most important of all digital systems, the digital computer. The basic building blocks of digital systems are logic circuits and memory circuits. We shall study both in this book, beginning in Chapter 14. One final remark: Although the digital processing of signals is at present all-pervasive, there remain many signal-processing functions that are best performed by analog circuits. Indeed, many electronic systems include both analog and digital parts. It follows that a good electronics engineer must be proficient in the design of both analog and digital circuits, or mixed-signal or mixed-mode design as it is currently known. Such is the aim of this book. EXERCISE 1.9 Consider a 4-bit digital word D = b3b2b1b0 (see Eq. 1.3) used to represent an analog signal vA that varies between 0 V and +15 V. (a) Give D corresponding to vA = 0 V, 1 V, 2 V, and 15 V. (b) What change in vA causes a change from 0 to 1 in (i) b0, (ii) b1, (iii) b2, and (iv) b3? (c) If vA = 5.2 V, what do you expect D to be? What is the resulting error in representation? Ans. (a) 0000, 0001, 0010, 1111; (b) +1 V, +2 V, +4 V, +8 V; (c) 0101, –4% ANALOG VS. DIGITAL CIRCUIT ENGINEERS: As digital became the preferred implementation of more and more signal-processing functions, the need arose for greater numbers of digital circuit design engineers. Yet despite predictions made periodically that the demand for analog circuit design engineers would lessen, this has not been the case. Rather, the demand for analog engineers has, if anything, increased. What is true, however, is that the skill level required of analog engineers has risen. Not only are they asked to design circuits of greater sophistication and tighter specifications, but they also have to do this using technologies that are optimized for digital (and not analog) circuits. This is dictated by economics, as digital usually constitutes the larger part of most systems. 1.4 Amplifiers In this section, we shall introduce the most fundamental signal-processing function, one that is employed in some form in almost every electronic system, namely, signal amplification. We shall study the amplifier as a circuit building block; that is, we shall consider its external characteristics and leave the design of its internal circuit to later chapters. 1.4.1 Signal Amplification From a conceptual point of view the simplest signal-processing task is that of signal amplification. The need for amplification arises because transducers provide signals that 16 Chapter 1 Signals and Amplifiers are said to be “weak,” that is, in the microvolt (μV) or millivolt (mV) range and possessing little energy. Such signals are too small for reliable processing, and processing is much easier if the signal magnitude is made larger. The functional block that accomplishes this task is the signal amplifier. It is appropriate at this point to discuss the need for linearity in amplifiers. Care must be exercised in the amplification of a signal, so that the information contained in the signal is not changed and no new information is introduced. Thus when we feed the signal shown in Fig. 1.3 to an amplifier, we want the output signal of the amplifier to be an exact replica of that at the input, except of course for having larger magnitude. In other words, the “wiggles” in the output waveform must be identical to those in the input waveform. Any change in waveform is considered to be distortion and is obviously undesirable. An amplifier that preserves the details of the signal waveform is characterized by the relationship vo(t) = Avi(t) (1.4) where vi and vo are the input and output signals, respectively, and A is a constant representing the magnitude of amplification, known as amplifier gain. Equation (1.4) is a linear relationship; hence the amplifier it describes is a linear amplifier. It should be easy to see that if the relationship between vo and vi contains higher powers of vi, then the waveform of vo will no longer be identical to that of vi. The amplifier is then said to exhibit nonlinear distortion. The amplifiers discussed so far are primarily intended to operate on very small input signals. Their purpose is to make the signal magnitude larger, and therefore they are thought of as voltage amplifiers. The preamplifier in the home stereo system is an example of a voltage amplifier. At this time we wish to mention another type of amplifier, namely, the power amplifier. Such an amplifier may provide only a modest amount of voltage gain but substantial current gain. Thus while absorbing little power from the input signal source to which it is connected, often a preamplifier, it delivers large amounts of power to its load. An example is found in the power amplifier of the home stereo system, whose purpose is to provide sufficient power to drive the loudspeaker, which is the amplifier load. Here we should note that the loudspeaker is the output transducer of the stereo system; it converts the electric output signal of the system into an acoustic signal. A further appreciation of the need for linearity can be acquired by reflecting on the power amplifier. A linear power amplifier causes both soft and loud music passages to be reproduced without distortion. 1.4.2 Amplifier Circuit Symbol The signal amplifier is obviously a two-port circuit. Its function is conveniently represented by the circuit symbol of Fig. 1.11(a). This symbol clearly distinguishes the input and output ports and indicates the direction of signal flow. Thus, in subsequent diagrams it will not be necessary to label the two ports “input” and “output.” For generality we have shown the amplifier to have two input terminals that are distinct from the two output terminals. A more common situation is illustrated in Fig. 1.11(b), where a common terminal exists between the input and output ports of the amplifier. This common terminal is used as a reference point and is called the circuit ground. 1.4 Amplifiers 17 (a) Figure 1.11 (a) Circuit symbol for amplifier. (b) An amplifier with a common terminal (ground) between the input and output ports. (a) Figure 1.12 (a) A voltage amplifier fed with a signal vI (t) and connected to a load resistance RL. (b) Transfer characteristic of a linear voltage amplifier with voltage gain Av . 1.4.3 Voltage Gain A linear amplifier accepts an input signal vI(t) and provides at the output, across a load resistance RL (see Fig. 1.12(a)), an output signal vO(t) that is a magnified replica of vI (t). The voltage gain of the amplifier is defined by Voltage gain (Av ) = vO vI (1.5) Fig. 1.12(b) shows the transfer characteristic of a linear amplifier. If we apply to the input of this amplifier a sinusoidal voltage of amplitude Vˆ , we obtain at the output a sinusoid of amplitude AvVˆ . 1.4.4 Power Gain and Current Gain An amplifier increases the signal power, an important feature that distinguishes an amplifier from a transformer. In the case of a transformer, although the voltage delivered to the load could be greater than the voltage feeding the input side (the primary), the power delivered to the load (from the secondary side of the transformer) is less than or at most equal to the 18 Chapter 1 Signals and Amplifiers power supplied by the signal source. On the other hand, an amplifier provides the load with power greater than that obtained from the signal source. That is, amplifiers have power gain. The power gain of the amplifier in Fig. 1.12(a) is defined as Power gain (Ap) ≡ load power (PL) input power(PI ) = vOiO vI iI (1.6) (1.7) where iO is the current that the amplifier delivers to the load (RL), iO = vO/RL, and iI is the current the amplifier draws from the signal source. The current gain of the amplifier is defined as Current gain (Ai) ≡ iO iI (1.8) From Eqs. (1.5) to (1.8) we note that Ap = Av Ai (1.9) 1.4.5 Expressing Gain in Decibels The amplifier gains defined above are ratios of similarly dimensioned quantities. Thus they will be expressed either as dimensionless numbers or, for emphasis, as V/V for the voltage gain, A/A for the current gain, and W/W for the power gain. Alternatively, for a number of reasons, some of them historic, electronics engineers express amplifier gain with a logarithmic measure. Specifically the voltage gain Av can be expressed as Voltage gain in decibels = 20 log |Av| dB and the current gain Ai can be expressed as Current gain in decibels = 20 log |Ai| dB Since power is related to voltage (or current) squared, the power gain Ap can be expressed in decibels as Power gain in decibels = 10 log Ap dB The absolute values of the voltage and current gains are used because in some cases Av or Ai will be a negative number. A negative gain Av simply means that there is a 180° phase difference between input and output signals; it does not imply that the amplifier is attenuating the signal. On the other hand, an amplifier whose voltage gain is, say, −20 dB is in fact attenuating the input signal by a factor of 10 (i.e., Av = 0.1 V/V). 1.4.6 The Amplifier Power Supplies Since the power delivered to the load is greater than the power drawn from the signal source, the question arises as to the source of this additional power. The answer is found by observing that amplifiers need dc power supplies for their operation. These dc sources supply the extra power delivered to the load as well as any power that might be dissipated in the internal circuit 1.4 Amplifiers 19 ICC VCC VCC ICC IEE IEE VEE VEE (a) (b) Figure 1.13 An amplifier that requires two dc supplies (shown as batteries) for operation. of the amplifier (such power is converted to heat). In Fig. 1.12(a) we have not explicitly shown these dc sources. Figure 1.13(a) shows an amplifier that requires two dc sources: one positive of value VCC and one negative of value VEE. The amplifier has two terminals, labeled V + and V −, for connection to the dc supplies. For the amplifier to operate, the terminal labeled V + has to be connected to the positive side of a dc source whose voltage is VCC and whose negative side is connected to the circuit ground. Also, the terminal labeled V − has to be connected to the negative side of a dc source whose voltage is VEE and whose positive side is connected to the circuit ground. Now, if the current drawn from the positive supply is denoted ICC and that from the negative supply is IEE (see Fig. 1.13a), then the dc power delivered to the amplifier is Pdc = VCC ICC + VEE IEE If the power dissipated in the amplifier circuit is denoted Pdissipated, the power-balance equation for the amplifier can be written as Pdc + PI = PL + Pdissipated where PI is the power drawn from the signal source and PL is the power delivered to the load. Since the power drawn from the signal source is usually small, the amplifier power efficiency is defined as η ≡ PL × 100 Pdc (1.10) The power efficiency is an important performance parameter for amplifiers that handle large amounts of power. Such amplifiers, called power amplifiers, are used, for example, as output amplifiers of stereo systems. In order to simplify circuit diagrams, we shall adopt the convention illustrated in Fig. 1.13(b). Here the V + terminal is shown connected to an arrowhead pointing upward and the V − terminal to an arrowhead pointing downward. The corresponding voltage is indicated next to each arrowhead. Note that in many cases we will not explicitly show the connections 20 Chapter 1 Signals and Amplifiers of the amplifier to the dc power sources. Finally, we note that some amplifiers require only one power supply. Example 1.2 Consider an amplifier operating from ±10-V power supplies. It is fed with a sinusoidal voltage having 1 V peak and delivers a sinusoidal voltage output of 9 V peak to a 1-k load. The amplifier draws a current of 9.5 mA from each of its two power supplies. The input current of the amplifier is found to be sinusoidal with 0.1 mA peak. Find the voltage gain, the current gain, the power gain, the power drawn from the dc supplies, the power dissipated in the amplifier, and the amplifier efficiency. Solution or or or Av = 9 1 = 9 V/V Av = 20 log 9 = 19.1 dB Iˆo = 9V 1k = 9 mA Ai = Iˆo Iˆi = 9 0.1 = 90 A/A Ai = 20 log 90 = 39.1 dB 99 PL = Vorms Iorms = √ 2 √ 2 = 40.5 mW 1 0.1 PI = Virms Iirms = √ 2 √ 2 = 0.05 mW Ap = PL PI = 40.5 0.05 = 810 W/W Ap = 10 log 810 = 29.1 dB Pdc = 10 × 9.5 + 10 × 9.5 = 190 mW Pdissipated = Pdc + PI − PL = 190 + 0.05 − 40.5 = 149.6 mW η = PL × 100 = 21.3% Pdc From the above example we observe that the amplifier converts some of the dc power it draws from the power supplies to signal power that it delivers to the load. 1.4 Amplifiers 21 1.4.7 Amplifier Saturation Practically speaking, the amplifier transfer characteristic remains linear over only a limited range of input and output voltages. For an amplifier operated from two power supplies the output voltage cannot exceed a specified positive limit and cannot decrease below a specified negative limit. The resulting transfer characteristic is shown in Fig. 1.14, with the positive and negative saturation levels denoted L+ and L−, respectively. Each of the two saturation levels is usually within a fraction of a volt of the voltage of the corresponding power supply. Obviously, in order to avoid distorting the output signal waveform, the input signal swing must be kept within the linear range of operation, L− Av ≤ vI ≤ L+ Av In Fig. 1.14, which shows two input waveforms and the corresponding output waveforms, the peaks of the larger waveform have been clipped off because of amplifier saturation. Figure 1.14 An amplifier transfer characteristic that is linear except for output saturation. 22 Chapter 1 Signals and Amplifiers iC ic Ic iC IC 0 t Figure 1.15 Symbol convention employed throughout the book. 1.4.8 Symbol Convention At this point, we draw the reader’s attention to the terminology we shall employ throughout the book. To illustrate the terminology, Fig. 1.15 shows the waveform of a current iC(t) that is flowing through a branch in a particular circuit. The current iC(t) consists of a dc component IC on which is superimposed a sinusoidal component ic(t) whose peak amplitude is Ic. Observe that at a time t, the total instantaneous current iC(t) is the sum of the dc current IC and the signal current ic(t), iC(t) = IC + ic(t) (1.11) where the signal current is given by ic(t) = Ic sin ωt Thus, we state some conventions: Total instantaneous quantities are denoted by a lowercase symbol with uppercase subscript(s), for example, iC(t), vDS(t). Direct-current (dc) quantities are denoted by an uppercase symbol with uppercase subscript(s), for example, IC, VDS. Incremental signal quantities are denoted by a lowercase symbol with lowercase subscript(s), for example, ic(t), vgs(t). If the signal is a sine wave, then its amplitude is denoted by an uppercase symbol with lowercase subscript(s), for example, Ic, Vgs. Finally, although not shown in Fig. 1.15, dc power supplies are denoted by an uppercase letter with a double-letter uppercase subscript, for example, VCC, VDD. A similar notation is used for the dc current drawn from the power supply, for example, ICC, IDD. EXERCISES 1.10 An amplifier has a voltage gain of 100 V/V and a current gain of 1000 A/A. Express the voltage and current gains in decibels and find the power gain. Ans. 40 dB; 60 dB; 50 dB 1.5 Circuit Models for Amplifiers 23 1.11 An amplifier operating from a single 15-V supply provides a 12-V peak-to-peak sine-wave signal to a 1-k load and draws negligible input current from the signal source. The dc current drawn from the 15-V supply is 8 mA. What is the power dissipated in the amplifier, and what is the amplifier efficiency? Ans. 102 mW; 15% 1.5 Circuit Models for Amplifiers A substantial part of this book is concerned with the design of amplifier circuits that use transistors of various types. Such circuits will vary in complexity from those using a single transistor to those with 20 or more devices. In order to be able to apply the resulting amplifier circuit as a building block in a system, one must be able to characterize, or model, its terminal behavior. In this section, we study simple but effective amplifier models. These models apply irrespective of the complexity of the internal circuit of the amplifier. The values of the model parameters can be found either by analyzing the amplifier circuit or by performing measurements at the amplifier terminals. 1.5.1 Voltage Amplifiers Figure 1.16(a) shows a circuit model for the voltage amplifier. The model consists of a voltage-controlled voltage source having a gain factor Avo, an input resistance Ri that accounts for the fact that the amplifier draws an input current from the signal source, and an output resistance Ro that accounts for the change in output voltage as the amplifier is called upon to supply output current to a load. To be specific, we show in Fig. 1.16(b) the amplifier model fed with a signal voltage source vs having a resistance Rs and connected at the output to a load resistance RL. The nonzero output resistance Ro causes only a fraction of Av ovi to appear across the output. Using the voltage-divider rule we obtain Thus the voltage gain is given by vo = Av ovi RL RL + Ro Av ≡ vo vi = Av o RL RL + Ro (1.12) It follows that in order not to lose gain in coupling the amplifier output to a load, the output resistance Ro should be much smaller than the load resistance RL. In other words, for a given RL one must design the amplifier so that its Ro is much smaller than RL. Furthermore, there are applications in which RL is known to vary over a certain range. In order to keep the output voltage vo as constant as possible, the amplifier is designed with Ro much smaller than the lowest value of RL. An ideal voltage amplifier is one with Ro = 0. Equation (1.12) indicates also that for RL = ∞, Av = Av o. Thus Av o is the voltage gain of the unloaded amplifier, or the open-circuit voltage gain. It should also be clear that in specifying the voltage gain of an amplifier, one must also specify the value of load resistance at which this gain is measured or 24 Chapter 1 Signals and Amplifiers ϩ vo Ϫ (a) ii vs io ϩ vo Ϫ (b) Figure 1.16 (a) Circuit model for the voltage amplifier. (b) The voltage amplifier with input signal source and load. calculated. If a load resistance is not specified, it is normally assumed that the given voltage gain is the open-circuit gain Av o. The finite input resistance Ri introduces another voltage-divider action at the input, with the result that only a fraction of the source signal vs actually reaches the input terminals of the amplifier; that is, vi = vs Ri Ri + Rs (1.13) It follows that in order not to lose a significant portion of the input signal in coupling the signal source to the amplifier input, the amplifier must be designed to have an input resistance Ri much greater than the resistance of the signal source, Ri Rs. Furthermore, there are applications in which the source resistance is known to vary over a certain range. To minimize the effect of this variation on the value of the signal that appears at the input of the amplifier, the design ensures that Ri is much greater than the largest value of Rs. An ideal voltage amplifier is one with Ri = ∞. In this ideal case both the current gain and power gain become infinite. The overall voltage gain (vo/vs) can be found by combining Eqs. (1.12) and (1.13), vo vs = Av o Ri Ri + Rs RL RL + Ro There are situations in which one is interested not in voltage gain but only in a significant power gain. For instance, the source signal can have a respectable voltage but a source resistance that is much greater than the load resistance. Connecting the source directly to the load would result in significant signal attenuation. In such a case, one requires an amplifier with a high input resistance (much greater than the source resistance) and a low output resistance (much smaller than the load resistance) but with a modest voltage gain (or even unity gain). 1.5 Circuit Models for Amplifiers 25 Such an amplifier is referred to as a buffer amplifier. We shall encounter buffer amplifiers often throughout this book. EXERCISES 1.12 A transducer characterized by a voltage of 1 V rms and a resistance of 1 M is available to drive a 10- load. If connected directly, what voltage and power levels result at the load? If a unity-gain (i.e., Av o = 1) buffer amplifier with 1-M input resistance and 10- output resistance is interposed between source and load, what do the output voltage and power levels become? For the new arrangement, find the voltage gain from source to load, and the power gain (both expressed in decibels). Ans. 10 μV rms; 10−11 W; 0.25 V; 6.25 mW; −12 dB; 44 dB 1.13 The output voltage of a voltage amplifier has been found to decrease by 20% when a load resistance of 1 k is connected. What is the value of the amplifier output resistance? Ans. 250 1.14 An amplifier with a voltage gain of +40 dB, an input resistance of 10 k , and an output resistance of 1 k is used to drive a 1-k load. What is the value of Av o? Find the value of the power gain in decibels. Ans. 100 V/V; 44 dB 1.5.2 Cascaded Amplifiers To meet given amplifier specifications, we often need to design the amplifier as a cascade of two or more stages. The stages are usually not identical; rather, each is designed to serve a specific purpose. For instance, in order to provide the overall amplifier with a large input resistance, the first stage is usually required to have a large input resistance. Also, in order to equip the overall amplifier with a low output resistance, the final stage in the cascade is usually designed to have a low output resistance. To illustrate the analysis and design of cascaded amplifiers, we consider a practical example. Example 1.3 Figure 1.17 depicts an amplifier composed of a cascade of three stages. The amplifier is fed by a signal source with a source resistance of 100 k and delivers its output into a load resistance of 100 . The first stage has a relatively high input resistance and a modest gain factor of 10. The second stage has a higher gain factor but lower input resistance. Finally, the last, or output, stage has unity gain but a low output resistance. We wish to evaluate the overall voltage gain, that is, vL/vs, the current gain, and the power gain. 26 Chapter 1 Signals and Amplifiers Example 1.3 continued Source 100 k⍀ Stage 1 1 k⍀ ϩ ϩ ϩ Ϫ 1 M⍀ ϩ Ϫ Ϫ Ϫ Stage 2 1 k⍀ Stage 3 10 ⍀ ϩ ϩ Ϫ 10 k⍀ ϩ Ϫ Ϫ Load ϩ 100 ⍀ Ϫ Figure 1.17 Three-stage amplifier for Example 1.3. Solution The fraction of source signal appearing at the input terminals of the amplifier is obtained using the voltage-divider rule at the input, as follows: vi1 = 1M = 0.909 V/V vs 1 M + 100 k The voltage gain of the first stage is obtained by considering the input resistance of the second stage to be the load of the first stage; that is, Av 1 ≡ v i2 v i1 = 100 k 10 100 k +1 k = 9.9 V/V Similarly, the voltage gain of the second stage is obtained by considering the input resistance of the third stage to be the load of the second stage, Av 2 ≡ v i3 v i2 = 100 10 10 k k +1 k = 90.9 V/V Finally, the voltage gain of the output stage is as follows: Av 3 ≡ vL v i3 = 100 1 100 + 10 = 0.909 V/V The total gain of the three stages in cascade can now be found from Av ≡ vL v i1 = Av 1Av 2Av 3 = 818 V/V or 58.3 dB. To find the voltage gain from source to load, we multiply Av by the factor representing the loss of gain at the input; that is, vL vs = vL v i1 v i1 vs = Av v i1 vs = 818 × 0.909 = 743.6 V/V or 57.4 dB. 1.5 Circuit Models for Amplifiers 27 The current gain is found as follows: Ai ≡ io ii = v L /100 vi1/1 M = 104 × Av = 8.18 × 106 A/A or 138.3 dB. The power gain is found from Ap ≡ PL PI = v L io v i1 ii = Av Ai = 818 × 8.18 × 106 = 66.9 × 108 W/W or 98.3 dB. Note that 1 Ap(dB) = 2 [Av(dB) + Ai(dB)] A few comments on the cascade amplifier in the above example are in order. To avoid losing signal strength at the amplifier input where the signal is usually very small, the first stage is designed to have a relatively large input resistance (1 M ), which is much larger than the source resistance. The trade-off appears to be a moderate voltage gain (10 V/V). The second stage does not need to have such a high input resistance; rather, here we need to realize the bulk of the required voltage gain. The third and final, or output, stage is not asked to provide any voltage gain; rather, it functions as a buffer amplifier, providing a relatively large input resistance and a low output resistance, much lower than RL. It is this stage that enables connecting the amplifier to the 100- load. These points can be made more concrete by solving the following exercises. In so doing, observe that in finding the gain of an amplifier stage in a cascade amplifier, the loading effect of the succeeding amplifier stage must be taken into account as we have done in the above example. EXERCISES 1.15 What would the overall voltage gain of the cascade amplifier in Example 1.3 be without stage 3 (i.e., with the load resistance connected to the output of the second stage)? Ans. 81.8 V/V; a decrease by a factor of 9. 1.16 For the cascade amplifier of Example 1.3, let vs be 1 mV. Find vi1, vi2, vi3, and vL. Ans. 0.91 mV; 9 mV; 818 mV; 744 mV 1.17 (a) Model the three-stage amplifier of Example 1.3 (without the source and load), using the voltage amplifier model of Fig. 1.16(a). What are the values of Ri, Av o, and Ro? (b) If RL varies in the range 10 to 1000 , find the corresponding range of the overall voltage gain, v o /v s . Ans. 1 M , 900 V/V, 10 ; 409 V/V to 810 V/V 28 Chapter 1 Signals and Amplifiers 1.5.3 Other Amplifier Types In the design of an electronic system, the signal of interest—whether at the system input, at an intermediate stage, or at the output—can be either a voltage or a current. For instance, some transducers have very high output resistances and can be more appropriately modeled as current sources. Similarly, there are applications in which the output current rather than the voltage is of interest. Thus, although it is the most popular, the voltage amplifier considered above is just one of four possible amplifier types. The other three are the current amplifier, the transconductance amplifier, and the transresistance amplifier. Table 1.1 shows the four amplifier types, their circuit models, the definition of their gain parameters, and the ideal values of their input and output resistances. 1.5.4 Relationships between the Four Amplifier Models Although for a given amplifier a particular one of the four models in Table 1.1 is most preferable, any of the four can be used to model any amplifier. In fact, simple relationships can be derived to relate the parameters of the various models. For instance, the open-circuit Table 1.1 The Four Amplifier Types Type Voltage Amplifier Circuit Model Gain Parameter Ideal Characteristics Open-Circuit Voltage Gain Av o ≡ vo vi (V/V) io =0 Ri = ∞ Ro = 0 Current Amplifier Short-Circuit Current Gain Ais ≡ io ii (A/A) vo =0 Ri = 0 Ro = ∞ Transconductance Amplifier Short-Circuit Transconductance Gm ≡ io vi (A/V) vo =0 Ri = ∞ Ro = ∞ Transresistance Amplifier Open-Circuit Transresistance Rm ≡ vo ii (V/A) io =0 Ri = 0 Ro = 0 1.5 Circuit Models for Amplifiers 29 voltage gain Av o can be related to the short-circuit current gain Ais as follows: The open-circuit output voltage given by the voltage amplifier model of Table 1.1 is Avovi. The current amplifier model in the same table gives an open-circuit output voltage of AisiiRo. Equating these two values and noting that ii = vi/Ri gives Av o = Ais Ro Ri (1.14) Similarly, we can show that Av o = GmRo (1.15) and Av o = Rm Ri (1.16) The expressions in Eqs. (1.14) to (1.16) can be used to relate any two of the gain parameters Av o, Ais, Gm, and Rm. 1.5.5 Determining Ri and Ro From the amplifier circuit models given in Table 1.1, we observe that the input resistance Ri of the amplifier can be determined by applying an input voltage vi and measuring (or calculating) the input current ii; that is, Ri = vi/ii. The output resistance is found as the ratio of the open-circuit output voltage to the short-circuit output current. Alternatively, the output resistance can be found by eliminating the input signal source (then ii and vi will both be zero) and applying a voltage signal vx to the output of the amplifier, as shown in Fig. 1.18. If we denote the current drawn from vx into the output terminals as ix (note that ix is opposite in direction to io), then Ro = vx/ix. Although these techniques are conceptually correct, in actual practice more refined methods are employed in measuring Ri and Ro. 1.5.6 Unilateral Models The amplifier models considered above are unilateral; that is, signal flow is unidirectional, from input to output. Most real amplifiers show some reverse transmission, which is usually undesirable but must nonetheless be modeled. We shall not pursue this point further at this time except to mention that more complete models for linear two-port networks are given in Appendix C. Also, in later chapters, we will find it necessary in certain cases to augment the models of Table 1.1 to take into account the nonunilateral nature of some transistor amplifiers. ix ϩ Ϫ vx Ro ϵ vx ix Figure 1.18 Determining the output resistance. 30 Chapter 1 Signals and Amplifiers Example 1.4 The bipolar junction transistor (BJT), which will be studied in Chapter 6, is a three-terminal device that when powered up by a dc source (battery) and operated with small signals can be modeled by the linear circuit shown in Fig. 1.19(a). The three terminals are the base (B), the emitter (E), and the collector (C). The heart of the model is a transconductance amplifier represented by an input resistance between B and E (denoted rπ ), a short-circuit transconductance gm, and an output resistance ro. Rs B C B ϩ vbe rp Ϫ gmvbe C vs ϩ Ϫ ro ϩ vbe rp Ϫ E ϩ gmvbe ro RL vo Ϫ E (a) (b) B ib ϩ vbe rp Ϫ C ro bib E (c) Figure 1.19 (a) Small-signal circuit model for a bipolar junction transistor (BJT). (b) The BJT connected as an amplifier with the emitter as a common terminal between input and output (called a common-emitter amplifier). (c) An alternative small-signal circuit model for the BJT. (a) With the emitter used as a common terminal between input and output, Fig. 1.19(b) shows a transistor amplifier known as a common-emitter or grounded-emitter circuit. Derive an expression for the voltage gain vo/vs, and evaluate its magnitude for the case Rs = 5 k , rπ = 2.5 k , gm = 40 mA/V, ro = 100 k , and RL = 5 k . What would the gain value be if the effect of ro were neglected? (b) An alternative model for the transistor in which a current amplifier rather than a transconductance amplifier is utilized is shown in Fig. 1.19(c). What must the short-circuit current gain β be? Give both an expression and a value. 1.5 Circuit Models for Amplifiers 31 Solution (a) Refer to Fig. 1.19(b). We use the voltage-divider rule to determine the fraction of input signal that appears at the amplifier input as v be = vs rπ rπ + Rs (1.17) Next we determine the output voltage vo by multiplying the current (gmvbe) by the resistance (RL ro), vo = −gmvbe(RL ro) (1.18) Substituting for vbe from Eq. (1.17) yields the voltage-gain expression vo vs =− rπ rπ + Rs gm (RL ro) (1.19) Observe that the gain is negative, indicating that this amplifier is inverting. For the given component values, vo = − 2.5 × 40 × (5 100) vs 2.5 + 5 = −63.5 V/V Neglecting the effect of ro, we obtain vo − 2.5 × 40 × 5 vs 2.5 + 5 = −66.7 V/V which is quite close to the value obtained including ro. This is not surprising, since ro RL. (b) For the model in Fig. 1.19(c) to be equivalent to that in Fig. 1.19(a), But ib = vbe/rπ ; thus, For the values given, βib = gmvbe β = gmrπ β = 40 mA/V × 2.5 k = 100 A/A 32 Chapter 1 Signals and Amplifiers EXERCISES 1.18 Consider a current amplifier having the model shown in the second row of Table 1.1. Let the amplifier be fed with a signal current-source is having a resistance Rs, and let the output be connected to a load resistance RL. Show that the overall current gain is given by io is = Ais Rs Rs + Ri Ro Ro + RL 1.19 Consider the transconductance amplifier whose model is shown in the third row of Table 1.1. Let a voltage signal source vs with a source resistance Rs be connected to the input and a load resistance RL be connected to the output. Show that the overall voltage gain is given by vo vs = Gm Ri Ri + Rs (Ro RL ) 1.20 Consider a transresistance amplifier having the model shown in the fourth row of Table 1.1. Let the amplifier be fed with a signal current source is having a resistance Rs, and let the output be connected to a load resistance RL. Show that the overall gain is given by vo is = Rm Rs Rs + Ri RL RL + Ro 1.21 Find the input resistance between terminals B and G in the circuit shown in Fig. E1.21. The voltage vx is a test voltage with the input resistance Rin defined as Rin ≡ vx/ix. ix Rin Ans. Rin = rπ + (β + 1)Re Figure E1.21 1.6 Frequency Response of Amplifiers 33 1.6 Frequency Response of Amplifiers2 From Section 1.2 we know that the input signal to an amplifier can always be expressed as the sum of sinusoidal signals. It follows that an important characterization of an amplifier is in terms of its response to input sinusoids of different frequencies. Such a characterization of amplifier performance is known as the amplifier frequency response. 1.6.1 Measuring the Amplifier Frequency Response We shall introduce the subject of amplifier frequency response by showing how it can be measured. Figure 1.20 depicts a linear voltage amplifier fed at its input with a sine-wave signal of amplitude Vi and frequency ω. As the figure indicates, the signal measured at the amplifier output also is sinusoidal with exactly the same frequency ω. This is an important point to note: Whenever a sine-wave signal is applied to a linear circuit, the resulting output is sinusoidal with the same frequency as the input. In fact, the sine wave is the only signal that does not change shape as it passes through a linear circuit. Observe, however, that the output sinusoid will in general have a different amplitude and will be shifted in phase relative to the input. The ratio of the amplitude of the output sinusoid (Vo) to the amplitude of the input sinusoid (Vi) is the magnitude of the amplifier gain (or transmission) at the test frequency ω. Also, the angle φ is the phase of the amplifier transmission at the test frequency ω. If we denote the amplifier transmission, or transfer function as it is more commonly known, by T (ω), then |T (ω)| = Vo Vi ∠T (ω) = φ The response of the amplifier to a sinusoid of frequency ω is completely described by |T (ω)| and ∠T (ω). Now, to obtain the complete frequency response of the amplifier we simply change the frequency of the input sinusoid and measure the new value for |T | and ∠T . The end result will be a table and/or graph of gain magnitude [|T (ω)|] versus frequency and a table and/or graph of phase angle [∠T (ω)] versus frequency. These two plots together constitute the frequency response of the amplifier; the first is known as the magnitude or amplitude vi ϭ Vi sin ␻ t ϩ Ϫ Linear amplifier ϩ vo ϭ Vo sin (␻t ϩ ␾ ) Ϫ Figure 1.20 Measuring the frequency response of a linear amplifier: At the test frequency, the amplifier gain is characterized by its magnitude (Vo/Vi) and phase φ. 2Except for its use in the study of the frequency response of op-amp circuits in Sections 2.5 and 2.7, the material in this section will not be needed in a substantial manner until Chapter 10. 34 Chapter 1 Signals and Amplifiers Figure 1.21 Typical magnitude response of an amplifier: |T (ω)| is the magnitude of the amplifier transfer function—that is, the ratio of the output Vo(ω) to the input Vi(ω). response, and the second is the phase response. Finally, we should mention that it is a common practice to express the magnitude of transmission in decibels and thus plot 20 log |T (ω)| versus frequency. 1.6.2 Amplifier Bandwidth Figure 1.21 shows the magnitude response of an amplifier. It indicates that the gain is almost constant over a wide frequency range, roughly between ω1 and ω2. Signals whose frequencies are below ω1 or above ω2 will experience lower gain, with the gain decreasing as we move farther away from ω1 and ω2. The band of frequencies over which the gain of the amplifier is almost constant, to within a certain number of decibels (usually 3 dB), is called the amplifier bandwidth. Normally the amplifier is designed so that its bandwidth coincides with the spectrum of the signals it is required to amplify. If this were not the case, the amplifier would distort the frequency spectrum of the input signal, with different components of the input signal being amplified by different amounts. 1.6.3 Evaluating the Frequency Response of Amplifiers Above, we described the method used to measure the frequency response of an amplifier. We now briefly discuss the method for analytically obtaining an expression for the frequency response. What we are about to say is just a preview of this important subject, whose detailed study is in Chapter 10. To evaluate the frequency response of an amplifier, one has to analyze the amplifier equivalent circuit model, taking into account all reactive components.3 Circuit analysis proceeds in the usual fashion but with inductances and capacitances represented by their reactances. An inductance L has a reactance or impedance jωL, and a capacitance C has a reactance or impedance 1/jωC or, equivalently, a susceptance or admittance jωC. Thus in a frequency-domain analysis we deal with impedances and/or admittances. The result of the 3Note that in the models considered in previous sections no reactive components were included. These were simplified models and cannot be used alone to predict the amplifier frequency response. 1.6 Frequency Response of Amplifiers 35 analysis is the amplifier transfer function T (ω) T (ω) = Vo(ω) Vi(ω) where Vi(ω) and Vo(ω) denote the input and output signals, respectively. T (ω) is generally a complex function whose magnitude |T (ω)| gives the magnitude of transmission or the magnitude response of the amplifier. The phase of T (ω) gives the phase response of the amplifier. In the analysis of a circuit to determine its frequency response, the algebraic manipulations can be considerably simplified by using the complex frequency variable s. In terms of s, the impedance of an inductance L is sL and that of a capacitance C is 1/sC. Replacing the reactive elements with their impedances and performing standard circuit analysis, we obtain the transfer function T (s) as T (s) ≡ Vo(s) Vi(s) Subsequently, we replace s by jω to determine the transfer function for physical frequencies, T (jω). Note that T ( jω) is the same function we called T (ω) above4; the additional j is included in order to emphasize that T ( jω) is obtained from T (s) by replacing s with jω. 1.6.4 Single-Time-Constant Networks In analyzing amplifier circuits to determine their frequency response, one is greatly aided by knowledge of the frequency-response characteristics of single-time-constant (STC) networks. An STC network is one that is composed of, or can be reduced to, one reactive component (inductance or capacitance) and one resistance. Examples are shown in Fig. 1.22. An STC network formed of an inductance L and a resistance R has a time constant τ = L/R. The time constant τ of an STC network composed of a capacitance C and a resistance R is given by τ = CR. Appendix E presents a study of STC networks and their responses to sinusoidal, step, and pulse inputs. Knowledge of this material will be needed at various points throughout this book, and the reader will be encouraged to refer to the appendix. At this point we need in particular the frequency-response results; we will, in fact, briefly discuss this important topic now. R Vi ϩ Ϫ C (a) ϩ Vo Vi ϩ Ϫ Ϫ C ϩ R Vo Ϫ (b) Figure 1.22 Two examples of STC networks: (a) a low-pass network and (b) a high-pass network. 4At this stage, we are using s simply as a shorthand for jω. We shall not require detailed knowledge of s-plane concepts until Chapter 10. A brief review of s-plane analysis is presented in Appendix F. 36 Chapter 1 Signals and Amplifiers Most STC networks can be classified into two categories,5 low pass (LP) and high pass (HP), with each of the two categories displaying distinctly different signal responses. As an example, the STC network shown in Fig. 1.22(a) is of the low-pass type and that in Fig. 1.22(b) is of the high-pass type. To see the reasoning behind this classification, observe that the transfer function of each of these two circuits can be expressed as a voltage-divider ratio, with the divider composed of a resistor and a capacitor. Now, recalling how the impedance of a capacitor varies with frequency (Z = 1/jωC), it is easy to see that the transmission of the circuit in Fig. 1.22(a) will decrease with frequency and approach zero as ω approaches ∞. Thus the circuit of Fig. 1.22(a) acts as a low-pass filter6; it passes low-frequency, sine-wave inputs with little or no attenuation (at ω = 0, the transmission is unity) and attenuates high-frequency input sinusoids. The circuit of Fig. 1.22(b) does the opposite; its transmission is unity at ω = ∞ and decreases as ω is reduced, reaching 0 for ω = 0. The latter circuit, therefore, performs as a high-pass filter. Table 1.2 provides a summary of the frequency-response results for STC networks of both types.7 Also, sketches of the magnitude and phase responses are given in Figs. 1.23 and 1.24. These frequency-response diagrams are known as Bode plots, and the 3-dB frequency (ω0) is also known as the corner frequency, break frequency, or pole frequency. The reader is urged to become familiar with this information and to consult Appendix E if further clarifications are needed. In particular, it is important to develop a facility for the rapid Table 1.2 Frequency Response of STC Networks Transfer Function T (s) Transfer Function (for physical frequencies) T ( jω) Magnitude Response |T ( jω)| Phase Response ∠T ( jω) Transmission at ω = 0 (dc) Transmission at ω = ∞ 3-dB Frequency Bode Plots Low-Pass (LP) High-Pass (HP) K 1 + (s/ω0) K 1 + j(ω/ω0) |K | 1 + (ω/ω0)2 Ks s + ω0 K 1 − j(ω0/ω) |K | 1 + (ω0/ω)2 − tan−1(ω/ω0) tan−1 (ω0 /ω) K 0 0 K ω0 = 1/τ ; τ ≡ time constant τ = CR or L/R in Fig. 1.23 in Fig. 1.24 5An important exception is the all-pass STC network studied in Chapter 17. 6A filter is a circuit that passes signals in a specified frequency band (the filter passband) and stops or severely attenuates (filters out) signals in another frequency band (the filter stopband). Filters will be studied in Chapter 17. 7The transfer functions in Table 1.2 are given in general form. For the circuits of Fig. 1.22, K = 1 and ω0 = 1/CR. 1.6 Frequency Response of Amplifiers 37 (a) (b) Figure 1.23 (a) Magnitude and (b) phase response of STC networks of the low-pass type. determination of the time constant τ of an STC circuit. The process is very simple: Set the independent voltage or current source to zero; “grab hold” of the two terminals of the reactive element (capacitor C or inductor L); and determine the equivalent resistance R that appears between these two terminals. The time constant is then CR or L/R. BODE PLOTS: In the 1930s, while working at Bell Labs, Hendrik Bode devised a simple but accurate method for using linearized asymptotic responses to graph gain and phase shift against frequency on a logarithmic scale. Such gain and phase presentations, together called Bode plots, have enormous importance in the design and analysis of the frequency-dependent behavior of systems large and small. 38 Chapter 1 Signals and Amplifiers (a) (b) Figure 1.24 (a) Magnitude and (b) phase response of STC networks of the high-pass type. Example 1.5 Figure 1.25 shows a voltage amplifier having an input resistance Ri, an input capacitance Ci, a gain factor μ, and an output resistance Ro. The amplifier is fed with a voltage source Vs having a source resistance Rs, and a load of resistance RL is connected to the output. Rs Ro Vs ϩ Ϫ ϩ ϩ Vi Ri Ci ϩ Ϫ ␮Vi RL Vo Ϫ Ϫ Figure 1.25 Circuit for Example 1.5. 1.6 Frequency Response of Amplifiers 39 (a) Derive an expression for the amplifier voltage gain Vo/Vs as a function of frequency. From this find expressions for the dc gain and the 3-dB frequency. (b) Calculate the values of the dc gain, the 3-dB frequency, and the frequency at which the gain becomes 0 dB (i.e., unity) for the case Rs = 20 k , Ri = 100 k , Ci = 60 pF, μ = 144 V/V, Ro = 200 RL = 1 k . (c) Find vo(t) for each of the following inputs: (i) vi = 0.1 sin 102t, V (ii) vi = 0.1 sin 105t, V (iii) vi = 0.1 sin 106t, V (iv) vi = 0.1 sin 108t, V , and Solution (a) Utilizing the voltage-divider rule, we can express Vi in terms of Vs as follows Vi = Vs Zi Zi + Rs where Zi is the amplifier input impedance. Since Zi is composed of two parallel elements, it is obviously easier to work in terms of Yi = 1/Zi. Toward that end we divide the numerator and denominator by Zi, thus obtaining Thus, Vi = Vs 1 1 + RsYi = Vs 1 + Rs 1 [(1/Ri ) + sCi ] Vi = 1 Vs 1 + (Rs/Ri) + sCiRs This expression can be put in the standard form for a low-pass STC network (see the top line of Table 1.2) by extracting [1 + (Rs/Ri)] from the denominator; thus we have Vi = 1 1 Vs 1 + (Rs/Ri) 1 + sCi[(RsRi)/(Rs + Ri)] (1.20) At the output side of the amplifier we can use the voltage-divider rule to write Vo = μVi RL RL + Ro This equation can be combined with Eq. (1.20) to obtain the amplifier transfer function as Vo = μ 1 1 1 Vs 1 + (Rs/Ri) 1 + (Ro/RL) 1 + sCi[(RsRi)/(Rs + Ri)] (1.21) 40 Chapter 1 Signals and Amplifiers Example 1.5 continued We note that only the last factor in this expression is new (compared with the expression derived in the last section). This factor is a result of the input capacitance Ci, with the time constant being τ = Ci Rs Ri Rs + Ri = Ci(Rs Ri) (1.22) We could have obtained this result by inspection: From Fig. 1.25 we see that the input circuit is an STC network and that its time constant can be found by reducing Vs to zero, with the result that the resistance seen by Ci is Ri in parallel with Rs. The transfer function in Eq. (1.21) is of the form K/(1 + (s/ωo)), which corresponds to a low-pass STC network. The dc gain is found as K ≡ Vo (s = 0) = μ 1 1 Vs 1 + (Rs/Ri) 1 + (Ro/RL) (1.23) The 3-dB frequency ω0 can be found from 1 1 ω0 = τ = Ci(Rs Ri) (1.24) Since the frequency response of this amplifier is of the low-pass STC type, the Bode plots for the gain magnitude and phase will take the form shown in Fig. 1.23, where K is given by Eq. (1.23) and ω0 is given by Eq. (1.24). (b) Substituting the numerical values given into Eq. (1.23) results in K = 144 1 + 1 (20/100) 1 + 1 (200/1000) = 100 V/V Thus the amplifier has a dc gain of 40 dB. Substituting the numerical values into Eq. (1.24) gives the 3-dB frequency 1 ω0 = 60 pF × (20 k //100 k ) = 1 = 106 rad/s 60 × 10−12 × (20 × 100/(20 + 100)) × 103 Thus, f0 = 106 2π = 159.2 kHz Since the gain falls off at the rate of –20 dB/decade, starting at ω0 (see Fig. 1.23a) the gain will reach 0 dB in two decades (a factor of 100); thus we have Unity-gain frequency = 100 × ω0 = 108 rad/s or 15.92 MHz 1.6 Frequency Response of Amplifiers 41 (c) To find vo(t) we need to determine the gain magnitude and phase at 102, 105, 106, and 108 rad/s. This can be done either approximately utilizing the Bode plots of Fig. 1.23 or exactly utilizing the expression for the amplifier transfer function, T ( jω) ≡ Vo ( jω) = 100 Vs 1 + j(ω/106) We shall do both: (i) For ω = 102 rad/s, which is (ω0/104), the Bode plots of Fig. 1.23 suggest that |T | = K = 100 and φ = 0°. The transfer function expression gives |T | 100 and φ = − tan−1 10−4 0°. Thus, vo(t) = 10 sin 102t, V (ii) For ω = 105 rad/s, which is (ω0/10), the Bode plots of Fig. 1.23 suggest that |T | K = 100 and φ = −5.7°. The transfer function expression gives |T | = 99.5 and φ = − tan−1 0.1 = −5.7°. Thus, vo(t) = 9.95 sin(105t − 5.7°), V (iii) For ω = 106 rad/s = ω0, |T | = √ 100/ 2 = 70.7 V/V or 37 dB and φ = −45°. Thus, vo(t) = 7.07 sin(106t − 45°), V (iv) For ω = 108 rad/s, which is (100 ω0), the Bode plots suggest that |T | = 1 and φ = −90°. The transfer function expression gives |T | 1 and φ = − tan−1 100 = −89.4°. Thus, vo(t) = 0.1 sin(108t − 89.4°), V 1.6.5 Classification of Amplifiers Based on Frequency Response Amplifiers can be classified based on the shape of their magnitude-response curve. Figure 1.26 shows typical frequency-response curves for various amplifier types. In Fig. 1.26(a) the gain remains constant over a wide frequency range, but falls off at low and high frequencies. This type of frequency response is common in audio amplifiers. As will be shown in later chapters, internal capacitances in the device (a transistor) cause the falloff of gain at high frequencies, just as Ci did in the circuit of Example 1.5. On the other hand, the falloff of gain at low frequencies is usually caused by coupling capacitors used to connect one amplifier stage to another, as indicated in Fig. 1.27. This practice is usually adopted to simplify the design process of the different stages. The coupling capacitors are usually chosen quite large (a fraction of a microfarad to a few tens of microfarads) so that their reactance (impedance) is small at the frequencies of interest. Nevertheless, at sufficiently low frequencies the reactance of a coupling capacitor will become large enough to cause part of the signal being coupled to appear as a voltage drop across the coupling capacitor, thus not reaching the subsequent stage. Coupling capacitors will thus cause loss of gain at low 42 Chapter 1 Signals and Amplifiers (b) (c) Figure 1.26 Frequency response for (a) a capacitively coupled amplifier, (b) a direct-coupled amplifier, and (c) a tuned or bandpass amplifier. s Figure 1.27 Use of a capacitor to couple amplifier stages. frequencies and cause the gain to be zero at dc. This is not at all surprising, since from Fig. 1.27 we observe that the coupling capacitor, acting together with the input resistance of the subsequent stage, forms a high-pass STC circuit. It is the frequency response of this high-pass circuit that accounts for the shape of the amplifier frequency response in Fig. 1.26(a) at the low-frequency end. There are many applications in which it is important that the amplifier maintain its gain at low frequencies down to dc. Furthermore, monolithic integrated-circuit (IC) technology does not allow the fabrication of large coupling capacitors. Thus IC amplifiers are usually designed as directly coupled or dc amplifiers (as opposed to capacitively coupled, or ac amplifiers). 1.6 Frequency Response of Amplifiers 43 Figure 1.26(b) shows the frequency response of a dc amplifier. Such a frequency response characterizes what is referred to as a low-pass amplifier. In a number of applications, such as in the design of radio and TV receivers, the need arises for an amplifier whose frequency response peaks around a certain frequency (called the center frequency) and falls off on both sides of this frequency, as shown in Fig. 1.26(c). Amplifiers with such a response are called tuned amplifiers, bandpass amplifiers, or bandpass filters. A tuned amplifier forms the heart of the front-end or tuner of a communication receiver; by adjusting its center frequency to coincide with the frequency of a desired communications channel (e.g., a radio station), the signal of this particular channel can be received while those of other channels are attenuated or filtered out. EXERCISES 1.22 Consider a voltage amplifier having a frequency response of the low-pass STC type with a dc gain of 60 dB and a 3-dB frequency of 1000 Hz. Find the gain in dB at f = 10 Hz, 10 kHz, 100 kHz, and 1 MHz. Ans. 60 dB; 40 dB; 20 dB; 0 dB D1.23 Consider a transconductance amplifier having the model shown in Table 1.1 with Ri = 5 k , Ro = 50 k , and Gm = 10 mA/V. If the amplifier load consists of a resistance RL in parallel with a capacitance CL, convince yourself that the voltage transfer function realized, Vo/Vi, is of the low-pass STC type. What is the lowest value that RL can have while a dc gain of at least 40 dB is obtained? With this value of RL connected, find the highest value that CL can have while a 3-dB bandwidth of at least 100 kHz is obtained. Ans. 12.5 k ; 159.2 pF D1.24 Consider the situation illustrated in Fig. 1.27. Let the output resistance of the first voltage amplifier be 1 k and the input resistance of the second voltage amplifier (including the resistor shown) be 9 k . The resulting equivalent circuit is shown in Fig. E1.24. Convince yourself that V2/Vs is a high-pass STC function. What is the smallest value for C that will ensure that the 3-dB frequency is not higher than 100 Hz? Ans. 0.16 μF Rs ϭ 1 k⍀ C Vs ϩ Ϫ ϩ Ri ϭ 9 k⍀ V2 Ϫ Figure E1.24 44 Chapter 1 Signals and Amplifiers Summary An electrical signal source can be represented in either the The´venin form (a voltage source vs in series with a source resistance Rs) or the Norton form (a current source is in parallel with a source resistance Rs). The The´venin voltage vs is the open-circuit voltage between the source terminals; the Norton current is is equal to the short-circuit current between the source terminals. For the two representations to be equivalent, vs and Rsis must be equal. A signal can be represented either by its waveform versus time or as the sum of sinusoids. The latter representation is known as the frequency spectrum of the signal. The sine-wave signal is completely characterize√d by its peak value (or rms value, which is the peak/ 2), its frequency (ω in rad/s or f in Hz; ω = 2πf and f = 1/T , where T is the period in seconds), and its phase with respect to an arbitrary reference time. Analog signals have magnitudes that can assume any value. Electronic circuits that process analog signals are called analog circuits. Sampling the magnitude of an analog signal at discrete instants of time and representing each signal sample by a number results in a digital signal. Digital signals are processed by digital circuits. The simplest digital signals are obtained when the binary system is used. An individual digital signal then assumes one of only two possible values: low and high (say, 0 V and +5 V), corresponding to logic 0 and logic 1, respectively. An analog-to-digital converter (ADC) provides at its output the digits of the binary number representing the analog signal sample applied to its input. The output digital signal can then be processed using digital circuits. Refer to Fig. 1.10 and Eq. (1.3). The transfer characteristic, vO versus vI , of a linear amplifier is a straight line with a slope equal to the voltage gain. Refer to Fig. 1.12. Amplifiers increase the signal power and thus require dc power supplies for their operation. The amplifier voltage gain can be expressed as a ratio Av in V/V or in decibels, 20 log |Av|, dB. Similarly, for current gain: Ai A/A or 20 log |Ai|, dB. For power gain: Ap W/W or 10 log Ap, dB. Depending on the signal to be amplified (voltage or current) and on the desired form of output signal (voltage or current), there are four basic amplifier types: voltage, current, transconductance, and transresistance amplifiers. For the circuit models and ideal characteristics of these four amplifier types, refer to Table 1.1. A given amplifier can be modeled by any one of the four models, in which case their parameters are related by the formulas in Eqs. (1.14) to (1.16). A sinusoid is the only signal whose waveform is unchanged through a linear circuit. Sinusoidal signals are used to measure the frequency response of amplifiers. The transfer function T (s) ≡ Vo(s)/Vi(s) of a voltage amplifier can be determined from circuit analysis. Substituting s = jω gives T ( jω), whose magnitude |T ( jω)| is the magnitude response, and whose phase φ(ω) is the phase response, of the amplifier. Amplifiers are classified according to the shape of their frequency response, |T ( jω)|. Refer to Fig. 1.26. Single-time-constant (STC) networks are those networks that are composed of, or can be reduced to, one reactive component (L or C) and one resistance (R). The time constant τ is either L/R or CR. STC networks can be classified into two categories: low pass (LP) and high pass (HP). LP networks pass dc and low frequencies and attenuate high frequencies. The opposite is true for HP networks. The gain of an LP (HP) STC circuit drops by 3 dB below the zero-frequency (infinite-frequency) value at a frequency ω0 = 1/τ . At high frequencies (low frequencies) the gain falls off at the rate of 6 dB/octave or 20 dB/decade. Refer to Table 1.2 on page 36 and Figs. 1.23 and 1.24. Further details are given in Appendix E. PROBLEMS Circuit Basics As a review of the basics of circuit analysis and in order for the readers to gauge their preparedness for the study of electronic circuits, this section presents a number of relevant circuit analysis problems. For a summary of The´venin’s and Norton’s theorems, refer to Appendix D. The problems are grouped in appropriate categories. Resistors and Ohm’s Law 1.1 Ohm’s law relates V , I, and R for a resistor. For each of the situations following, find the missing item: (a) R = 1 k , V = 5 V (b) V = 5 V, I = 1 mA (c) R = 10 k , I = 0.1 mA (d) R = 100 , V = 1 V Note: Volts, milliamps, and kilohms constitute a consistent set of units. 1.2 Measurements taken on various resistors are shown below. For each, calculate the power dissipated in the resistor and the power rating necessary for safe operation using standard components with power ratings of 1/8 W, 1/4 W, 1/2 W, 1 W, or 2 W: (a) 1 k conducting 20 mA (b) 1 k conducting 40 mA (c) 100 k conducting 1 mA (d) 10 k conducting 4 mA (e) 1 k dropping 20 V (f) 1 k dropping 11 V 1.3 Ohm’s law and the power law for a resistor relate V , I, R, and P, making only two variables independent. For each pair identified below, find the other two: (a) R = 1 k , I = 5 mA (b) V = 5 V, I = 1 mA (c) V = 10 V, P = 100 mW (d) I = 0.1 mA, P = 1 mW (e) R = 1 k , P = 1 W Combining Resistors 1.4 You are given three resistors whose values are 10 k , 20 k , and 40 k . How many different resistances can you create using series and parallel combinations of these three? List them in value order, lowest first. Be thorough and organized. (Hint: In your search, first consider all parallel combinations, then consider series combinations, and then consider series-parallel combinations, of which there are two kinds.) 1.5 In the analysis and test of electronic circuits, it is often useful to connect one resistor in parallel with another to obtain a nonstandard value, one which is smaller than the smaller of the two resistors. Often, particularly during circuit testing, one resistor is already installed, in which case the second, when connected in parallel, is said to “shunt” the first. If the original resistor is 10 k , what is the value of the shunting resistor needed to reduce the combined value by 1%, 5%, 10%, and 50%? What is the result of shunting a 10-k resistor by 1 M ? By 100 k ? By 10 k ? Voltage Dividers 1.6 Figure P1.6(a) shows a two-resistor voltage divider. Its function is to generate a voltage VO (smaller than the power-supply voltage VDD) at its output node X. The circuit looking back at node X is equivalent to that shown in Fig. P1.6(b). Observe that this is the The´venin equivalent of the voltage-divider circuit. Find expressions for VO and RO. VDD R1 X VO R2 RO X VO RO (a) (b) Figure P1.6 1.7 A two-resistor voltage divider employing a 2-k and a 3-k resistor is connected to a 5-V ground-referenced power supply to provide a 2-V voltage. Sketch the circuit. Assuming exact-valued resistors, what output voltage (measured to ground) and equivalent output resistance result? If the resistors used are not ideal but have a ±5% manufacturing tolerance, what are the extreme output voltages and resistances that can result? = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem 46 Chapter 1 Signals and Amplifiers CHAPTER 1 PROBLEMS D 1.8 You are given three resistors, each of 10 k , and a 9-V battery whose negative terminal is connected to ground. With a voltage divider using some or all of your resistors, how many positive-voltage sources of magnitude less than 9 V can you design? List them in order, smallest first. What is the output resistance (i.e., the The´venin resistance) of each? D *1.9 Two resistors, with nominal values of 4.7 k and 10 k , are used in a voltage divider with a +15-V supply to create a nominal +5-V output. Assuming the resistor values to be exact, what is the actual output voltage produced? Which resistor must be shunted (paralleled) by what third resistor to create a voltage-divider output of 5.00 V? If an output resistance of exactly 3.33 k is also required, what do you suggest? this problem. What is the value of the resistor required in each case? What is the input resistance of the current divider in each case? D 1.13 A particular electronic signal source generates currents in the range 0 mA to 0.5 mA under the condition that its load voltage not exceed 1 V. For loads causing more than 1 V to appear across the generator, the output current is no longer assured but will be reduced by some unknown amount. This circuit limitation, occurring, for example, at the peak of a sine-wave signal, will lead to undesirable signal distortion that must be avoided. If a 10-k load is to be connected, what must be done? What is the name of the circuit you must use? How many resistors are needed? What is (are) the(ir) value(s)? What is the range of current through the load? Current Dividers 1.10 Current dividers play an important role in circuit design. Therefore it is important to develop a facility for dealing with current dividers in circuit analysis. Figure P1.10 shows a two-resistor current divider fed with an ideal current source I. Show that I1 = R2 R1 + R2 I I2 = R1 R1 + R2 I and find the voltage V that develops across the current divider. I1 I2 ϩ I R1 R2 V Ϫ Figure P1.10 D 1.11 Design a simple current divider that will reduce the current provided to a 10-k load to one-third of that available from the source. D 1.12 A designer searches for a simple circuit to provide one-fifth of a signal current I to a load resistance R. Suggest a solution using one resistor. What must its value be? What is the input resistance of the resulting current divider? For a particular value R, the designer discovers that the otherwise-best-available resistor is 10% too high. Suggest two circuit topologies using one additional resistor that will solve The´ venin Equivalent Circuits 1.14 For the circuit in Fig. P1.14, find the The´venin equivalent circuit between terminals (a) 1 and 2, (b) 2 and 3, and (c) 1 and 3. 1 1 kΩ 1.5 V 2 1 kΩ 3 Figure P1.14 1.15 Through repeated application of The´venin’s theorem, find the The´venin equivalent of the circuit in Fig. P1.15 between node 4 and ground, and hence find the current that flows through a load resistance of 3 k connected between node 4 and ground. 1 20 kΩ 2 20 kΩ 3 20 kΩ 4 10 V 20 kΩ 20 kΩ 20 kΩ Figure P1.15 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 1 PROBLEMS Problems 47 Circuit Analysis 1.16 For the circuit shown in Fig. P1.16, find the current in each of the three resistors and the voltage (with respect to ground) at their common node using two methods: (a) Loop Equations: Define branch currents I1 and I2 in R1 and R2, respectively; write two equations; and solve them. (b) Node Equation: Define the node voltage V at the common node; write a single equation; and solve it. Which method do you prefer? Why? a much easier approach is possible: Find the The´venin equivalent of the circuit to the left of node 1 and the The´venin equivalent of the circuit to the right of node 2. Then solve the resulting simplified circuit. *1.18 For the circuit in Fig. P1.18, find the equivalent resistance to ground, Req. To do this, apply a voltage Vx between terminal X and ground and find the current drawn from Vx. Note that you can use particular special properties of the circuit to get the result directly! Now, if R4 is raised to 1.2 k , what does Req become? ϩ5 V R2 5 k⍀ ϩ10 V R1 10 k⍀ R3 2 k⍀ R1 1 kV R2 1 kV X R5 1 kV Req R3 1 kV R4 1 kV Figure P1.16 1.17 The circuit shown in Fig. P1.17 represents the equivalent circuit of an unbalanced bridge. It is required to calculate the current in the detector branch (R5) and the voltage across it. Although this can be done by using loop and node equations, Figure P1.18 1.19 Derive an expression for vo/vs for the circuit shown in Fig. P1.19. ϩ10 V R1 1 k⍀ R3 9.1 k⍀ 1 R5 2 2 k⍀ R2 1.2 k⍀ R4 11 k⍀ Figure P1.17 Rs vs ϩ Ϫ Figure P1.19 ϩ vp rp Ϫ ro gmvp ϩ RL vo Ϫ = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem 48 Chapter 1 Signals and Amplifiers CHAPTER 1 PROBLEMS AC Circuits 1.20 The periodicity of recurrent waveforms, such as sine waves or square waves, can be completely specified using only one of three possible parameters: radian frequency, ω, in radians per second (rad/s); (conventional) frequency, f , in hertz (Hz); or period T , in seconds (s). As well, each of the parameters can be specified numerically in one of several ways: using letter prefixes associated with the basic units, using scientific notation, or using some combination of both. Thus, for example, a particular period may be specified as 100 ns, 0.1 μs, 10−1 μs, 105 ps, or 1 × 10−7 s. (For the definition of the various prefixes used in electronics, see Appendix J.) For each of the measures listed below, express the trio of terms in scientific notation associated with the basic unit (e.g., 10−7 s rather than 10−1μs). (a) T = 10−4 ms (b) f = 1 GHz (c) ω = 6.28 × 102 rad/s (d) T = 10 s (e) f = 60 Hz (f) ω = 1 krad/s (g) f = 1900 MHz 1.21 Find the complex impedance, Z, of each of the following basic circuit elements at 60 Hz, 100 kHz, and 1 GHz: (a) R = 1 k (b) C = 10 nF (c) C = 10 pF (d) L = 10 mH (e) L = 1 μH sources, calculate the internal resistance, Rs; the Norton current, is; and the The´venin voltage, vs: (a) voc = 1 V, isc = 0.1 mA (b) voc = 0.1 V, isc = 1 μA 1.24 A particular signal source produces an output of 40 mV when loaded by a 100-k resistor and 10 mV when loaded by a 10-k resistor. Calculate the The´venin voltage, Norton current, and source resistance. 1.25 A temperature sensor is specified to provide 2 mV/°C. When connected to a load resistance of 5 k , the output voltage was measured to change by 10 mV, corresponding to a change in temperature of 10°C. What is the source resistance of the sensor? 1.26 Refer to the The´venin and Norton representations of the signal source (Fig. 1.1). If the current supplied by the source is denoted io and the voltage appearing between the source output terminals is denoted vo, sketch and clearly label vo versus io for 0 ≤ io ≤ is. 1.27 The connection of a signal source to an associated signal processor or amplifier generally involves some degree of signal loss as measured at the processor or amplifier input. Considering the two signal-source representations shown in Fig. 1.1, provide two sketches showing each signal-source representation connected to the input terminals (and corresponding input resistance) of a signal processor. What signal-processor input resistance will result in 95% of the open-circuit voltage being delivered to the processor? What input resistance will result in 95% of the short-circuit signal current entering the processor? Section 1.2: Frequency Spectrum of Signals 1.22 Find the complex impedance at 10 kHz of the following networks: (a) 1 k in series with 10 nF (b) 10 k in parallel with 0.01 μF (c) 100 k in parallel with 100 pF (d) 100 in series with 10 mH Section 1.1: Signals 1.23 Any given signal source provides an open-circuit voltage, voc, and a short-circuit current, isc. For the following 1.28 To familiarize yourself with typical values of angular frequency ω, conventional frequency f , and period T , complete the entries in the following table: Case a b c d e f ω (rad/s) 2 × 109 f (Hz) 5 × 109 60 6.28 × 104 T (s) 1 × 10−10 1 × 10−5 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 1 PROBLEMS Problems 49 1.29 For the following peak or rms values of some important amplitude Vˆ and the same frequency. Does this result depend sine waves, calculate the corresponding other value: on equality of the frequencies of the two waveforms? (a) 117 V rms, a household-power voltage in North America (b) 33.9 V peak, a somewhat common peak voltage in rectifier circuits (c) 220 V rms, a household-power voltage in parts of Europe (d) 220 kV rms, a high-voltage transmission-line voltage in North America 1.30 Give expressions for the sine-wave voltage signals having: (a) 10-V peak amplitude and 1-kHz frequency (b) 120-V rms and 60-Hz frequency (c) 0.2-V peak-to-peak and 2000-rad/s frequency (d) 100-mV peak and 1-ms period 1.31 Using the information provided by Eq. (1.2) in association with Fig. 1.5, characterize the signal represented by v(t) = 1/2 + 2/π(sin 2000π t + 1 sin 6000π t + 3 1 sin 10, 000π t + · · · ). Sketch the waveform. What is its 5 average value? Its peak-to-peak value? Its lowest value? Its highest value? Its frequency? Its period? 1.32 Measurements taken of a square-wave signal using a frequency-selective voltmeter (called a spectrum analyzer) show its spectrum to contain adjacent components (spectral lines) at 98 kHz and 126 kHz of amplitudes 63 mV and 49 mV, respectively. For this signal, what would direct measurement of the fundamental show its frequency and amplitude to be? What is the rms value of the fundamental? What are the peak-to-peak amplitude and period of the originating square wave? 1.33 Find the amplitude of a symmetrical square wave of period T that provides the same power as a sine wave of peak Section 1.3: Analog and Digital Signals 1.34 Give the binary representation of the following decimal numbers: 0, 6, 11, 28, and 59. 1.35 Consider a 4-bit digital word b3b2b1b0 in a format called signed-magnitude, in which the most significant bit, b3, is interpreted as a sign bit—0 for positive and 1 for negative values. List the values that can be represented by this scheme. What is peculiar about the representation of zero? For a particular analog-to-digital converter (ADC), each change in b0 corresponds to a 0.5-V change in the analog input. What is the full range of the analog signal that can be represented? What signed-magnitude digital code results for an input of +2.5 V? For −3.0 V? For +2.7 V? For −2.8 V? 1.36 Consider an N-bit ADC whose analog input varies between 0 and VFS (where the subscript FS denotes “full scale”). (a) Show that the least significant bit (LSB) corresponds to a change in the analog signal of VFS/(2N − 1). This is the resolution of the converter. (b) Convince yourself that the maximum error in the conversion (called the quantization error) is half the resolution; that is, the quantization error = VFS/2(2N − 1). (c) For VFS = 5 V, how many bits are required to obtain a resolution of 2 mV or better? What is the actual resolution obtained? What is the resulting quantization error? 1.37 Figure P1.37 shows the circuit of an N-bit digital-to-analog converter (DAC). Each of the N bits of the digital word to be converted controls one of the switches. Vref 2R b1 01 4R b2 01 8R b3 0 1 2NR bN 01 iO Figure P1.37 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 1 PROBLEMS 50 Chapter 1 Signals and Amplifiers When the bit is 0, the switch is in the position labeled 0; when the bit is 1, the switch is in the position labeled 1. The analog output is the current iO.Vref is a constant reference voltage. (a) Show that iO = Vref R b1 + b2 + · · · + bN 21 22 2N 1.42 Symmetrically saturating amplifiers, operating in the so-called clipping mode, can be used to convert sine waves to pseudo-square waves. For an amplifier with a small-signal gain of 1000 and clipping levels of ±10 V, what peak value of input sinusoid is needed to produce an output whose extremes are just at the edge of clipping? Clipped 90% of the time? Clipped 99% of the time? Section 1.5: Circuit Models for Amplifiers (b) Which bit is the LSB? Which is the MSB? (c) For Vref = 10 V, R = 10 k , and N = 8, find the maximum value of iO obtained. What is the change in iO resulting from the LSB changing from 0 to 1? 1.38 In compact-disc (CD) audio technology, the audio signal is sampled at 44.1 kHz. Each sample is represented by 16 bits. What is the speed of this system in bits per second? 1.43 Consider the voltage-amplifier circuit model shown in Fig. 1.16(b), in which Av o = 100 V/V under the following conditions: (a) Ri = 10Rs, RL = 10Ro (b) Ri = Rs, RL = Ro (c) Ri = Rs/10, RL = Ro/10 Calculate the overall voltage gain vo/vs in each case, expressed both directly and in decibels. Section 1.4: Amplifiers 1.39 Various amplifier and load combinations are measured as listed below using rms values. For each, find the voltage, current, and power gains (Av , Ai, and Ap, respectively) both as ratios and in dB: (a) vI = 100 mV, iI = 100 μA, vO = 10 V, RL = 100 (b) vI = 10 μV, iI = 100 nA, vO = 1 V, RL = 10 k (c) vI = 1 V, iI = 1 mA, vO = 5 V, RL = 10 1.40 An amplifier operating from ±3-V supplies provides a 2.2-V peak sine wave across a 100- load when provided with a 0.2-V peak input from which 1.0 mA peak is drawn. The average current in each supply is measured to be 20 mA.Find the voltage gain, current gain, and power gain expressed as ratios and in decibels as well as the supply power, amplifier dissipation, and amplifier efficiency. 1.41 An amplifier using balanced power supplies is known to saturate for signals extending within 1.0 V of either supply. For linear operation, its gain is 200 V/V. What is the rms value of the largest undistorted sine-wave output available, and input needed, with ±5-V supplies? With ±10-V supplies? With ±15-V supplies? 1.44 An amplifier with 40 dB of small-signal, open-circuit voltage gain, an input resistance of 1 M , and an output resistance of 100 , drives a load of 500 . What voltage and power gains (expressed in dB) would you expect with the load connected? If the amplifier has a peak output-current limitation of 20 mA, what is the rms value of the largest sine-wave input for which an undistorted output is possible? What is the corresponding output power available? 1.45 A 10-mV signal source having an internal resistance of 100 k is connected to an amplifier for which the input resistance is 10 k , the open-circuit voltage gain is 1000 V/V, and the output resistance is 1 k . The amplifier is connected in turn to a 100- load. What overall voltage gain results as measured from the source internal voltage to the load? Where did all the gain go? What would the gain be if the source was connected directly to the load? What is the ratio of these two gains? This ratio is a useful measure of the benefit the amplifier brings. 1.46 A buffer amplifier with a gain of 1 V/V has an input resistance of 1 M and an output resistance of 20 . It is connected between a 1-V, 200-k source and a 100- = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 1 PROBLEMS Problems 51 load. What load voltage results? What are the corresponding voltage, current, and power gains (in dB)? 1.47 Consider the cascade amplifier of Example 1.3. Find the overall voltage gain vo/vs obtained when the first and second stages are interchanged. Compare this value with the result in Example 1.3, and comment. 1.48 You are given two amplifiers, A and B, to connect in cascade between a 10-mV, 100-k source and a 100load. The amplifiers have voltage gain, input resistance, and output resistance as follows: for A, 100 V/V, 100 k , 10 k , respectively; for B, 10 V/V, 10 k , 1 k , respectively. Your problem is to decide how the amplifiers should be connected. To proceed, evaluate the two possible connections between source S and load L, namely, SABL and SBAL. Find the voltage gain for each both as a ratio and in decibels. Which amplifier arrangement is best? D *1.49 A designer has available voltage amplifiers with an input resistance of 10 k , an output resistance of 1 k , and an open-circuit voltage gain of 10. The signal source has a 10-k resistance and provides a 5-mV rms signal, and it is required to provide a signal of at least 3 V rms to a 200load. How many amplifier stages are required? What is the output voltage actually obtained? D *1.51 It is required to design a voltage amplifier to be driven from a signal source having a 5-mV peak amplitude and a source resistance of 10 k to supply a peak output of 2 V across a 1-k load. (a) What is the required voltage gain from the source to the load? (b) If the peak current available from the source is 0.1 μA, what is the smallest input resistance allowed? For the design with this value of Ri, find the overall current gain and power gain. (c) If the amplifier power supply limits the peak value of the output open-circuit voltage to 3 V, what is the largest output resistance allowed? (d) For the design with Ri as in (b) and Ro as in (c), what is the required value of open-circuit voltage gain, i.e., vo , vi RL =∞ of the amplifier? (e) If, as a possible design option, you are able to increase Ri to the nearest value of the form 1 × 10n and to decrease Ro to the nearest value of the form 1 × 10m , find (i) the input resistance achievable; (ii) the output resistance achievable; and (iii) the open-circuit voltage gain now required to meet the specifications. D *1.50 Design an amplifier that provides 0.5 W of signal power to a 100- load resistance. The signal source provides a 30-mV rms signal and has a resistance of 0.5 M . Three types of voltage-amplifier stages are available: (a) A high-input-resistance type with Ri = 1 M , Av o = 10, and Ro = 10 k (b) A high-gain type with Ri = 10 k , Av o = 100, and Ro = 1 k (c) A low-output-resistance type with Ri = 10 k , Av o = 1, and Ro = 20 Design a suitable amplifier using a combination of these stages. Your design should utilize the minimum number of stages and should ensure that the signal level is not reduced below 10 mV at any point in the amplifier chain. Find the load voltage and power output realized. D 1.52 A voltage amplifier with an input resistance of 20 k , an output resistance of 100 , and a gain of 1000 V/V is connected between a 100-k source with an open-circuit voltage of 10 mV and a 100- load. For this situation: (a) What output voltage results? (b) What is the voltage gain from source to load? (c) What is the voltage gain from the amplifier input to the load? (d) If the output voltage across the load is twice that needed and there are signs of internal amplifier overload, suggest the location and value of a single resistor that would produce the desired output. Choose an arrangement that would cause minimum disruption to an operating circuit. (Hint: Use parallel rather than series connections.) = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 1 PROBLEMS 52 Chapter 1 Signals and Amplifiers 1.53 A voltage amplifier delivers 200 mV across a load resistance of 1 k . It was found that the output voltage decreases by 5 mV when RL is decreased to 780 . What are the values of the open-circuit output voltage and the output resistance of the amplifier? 1.54 A current amplifier supplies 1 mA to a load resistance of 1 k . When the load resistance is increased to 12 k , the output current decreases to 0.5 mA. What are the values of the short-circuit output current and the output resistance of the amplifier? Rin 1.55 A current amplifier for which Ri = 100 , Ro = 10 k , and Ais = 100 A/A is to be connected between a 100-mV source with a resistance of 10 k and a load of 1 k . What are the values of current gain io/ii, of voltage gain vo/vs, and of power gain expressed directly and in decibels? 1.56 A transconductance amplifier with Ri = 2 k , Gm = 60 mA/V, and Ro = 20 k is fed with a voltage source having a source resistance of 1 k and is loaded with a 1-k resistance. Find the voltage gain realized. D **1.57 A designer is required to provide, across a 10-k load, the weighted sum, vO = 10v1 + 20v2, of input signals v1 and v2, each having a source resistance of 10 k . She has a number of transconductance amplifiers for which the input and output resistances are both 10 k and Gm = 20 mA/V, together with a selection of suitable resistors. Sketch an appropriate amplifier topology with additional resistors selected to provide the desired result. Your design should utilize the minimum number of amplifiers and resistors. (Hint: In your design, arrange to add currents.) 1.58 Figure P1.58 shows a transconductance amplifier whose output is fed back to its input. Find the input resistance Rin of the resulting one-port network. (Hint: Apply a test voltage vx between the two input terminals, and find the current ix drawn from the source. Then, Rin ≡ vx/ix.) Figure P1.58 D 1.59 It is required to design an amplifier to sense the open-circuit output voltage of a transducer and to provide a proportional voltage across a load resistor. The equivalent source resistance of the transducer is specified to vary in the range of 1 k to 10 k . Also, the load resistance varies in the range of 1 k to 10 k . The change in load voltage corresponding to the specified change in Rs should be 10% at most. Similarly, the change in load voltage corresponding to the specified change in RL should be limited to 10%. Also, corresponding to a 10-mV transducer open-circuit output voltage, the amplifier should provide a minimum of 1 V across the load. What type of amplifier is required? Sketch its circuit model, and specify the values of its parameters. Specify appropriate values for Ri and Ro of the form 1 × 10m . D 1.60 It is required to design an amplifier to sense the short-circuit output current of a transducer and to provide a proportional current through a load resistor. The equivalent source resistance of the transducer is specified to vary in the range of 1 k to 10 k . Similarly, the load resistance is known to vary over the range of 1 k to 10 k . The change in load current corresponding to the specified change in Rs is required to be limited to 10%. Similarly, the change in load current corresponding to the specified change in RL = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 1 PROBLEMS Problems 53 should be 10% at most. Also, for a nominal short-circuit output current of the transducer of 10 μA, the amplifier is required to provide a minimum of 1 mA through the load. What type of amplifier is required? Sketch the circuit model of the amplifier, and specify values for its param- eters. Select appropriate values for Ri and Ro in the form 1 × 10m . 1.63 For the circuit in Fig. P1.63, show that vc = −β RL vb rπ + (β + 1)RE and ve = RE vb RE + [rπ /(β + 1)] D 1.61 It is required to design an amplifier to sense the open-circuit output voltage of a transducer and to provide a proportional current through a load resistor. The equivalent source resistance of the transducer is specified to vary in the range of 1 k to 10 k . Also, the load resistance is known to vary in the range of 1 k to 10 k . The change in the B C ϩ ib r␲ ␤ib current supplied to the load corresponding to the specified change in Rs is to be 10% at most. Similarly, the change in load current corresponding to the specified change in RL is to be 10% at most. Also, for a nominal transducer open-circuit vb ϩ Ϫ RL vc E ϩ output voltage of 10 mV, the amplifier is required to provide RE ve a minimum of 1 mA current through the load. What type of amplifier is required? Sketch the amplifier circuit model, and Ϫ Ϫ specify values for its parameters. For Ri and Ro, specify values in the form 1 × 10m . Figure P1.63 D 1.62 It is required to design an amplifier to sense the short-circuit output current of a transducer and to provide a proportional voltage across a load resistor. The equivalent source resistance of the transducer is specified to vary in the range of 1 k to 10 k . Similarly, the load resistance is known to vary in the range of 1 k to 10 k . The change in load voltage corresponding to the specified change in Rs should be 10% at most. Similarly, the change in load voltage corresponding to the specified change in RL is to be limited to 10%. Also, for a nominal transducer short-circuit output current of 10 μA, the amplifier is required to provide a minimum voltage across the load of 1 V. What type of amplifier is required? Sketch its circuit model, and specify the values of the model parameters. For Ri and Ro, specify appropriate values in the form 1 × 10m . 1.64 An amplifier with an input resistance of 5 k , when driven by a current source of 1 μA and a source resistance of 200 k , has a short-circuit output current of 5 mA and an open-circuit output voltage of 10 V. If the amplifier is used to drive a 2-k load, give the values of the voltage gain, current gain, and power gain expressed as ratios and in decibels. 1.65 Figure P1.65(a) shows two transconductance amplifiers connected in a special configuration. Find vo in terms of v1 and v2. Let gm = 100 mA/V and R = 5 k . If v1 = v2 = 1 V, find the value of vo. Also, find vo for the case v1 = 1.01 V and v2 = 0.99 V. (Note: This circuit is called a differential amplifier and is given the symbol shown = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 1 PROBLEMS 54 Chapter 1 Signals and Amplifiers in Fig. P1.65(b). A particular type of differential amplifier known as an operational amplifier will be studied in Chapter 2.) to that of the voltage amplifier in Fig. 1.16(a), identify corresponding currents and voltages as well as the correspondence between the parameters of the amplifier equivalent circuit and the g parameters. Hence give the g parameter that corresponds to each of Ri, Av o, and Ro. Notice that there is an additional g parameter with no correspondence in the amplifier equivalent circuit. Which one? What does it signify? What assumption did we make about the amplifier that resulted in the absence of this particular g parameter from the equivalent circuit in Fig. 1.16(a)? I1 ϩ 1 V1 g11 Ϫ g12I2 ϩ g21V1 Ϫ g22 I2 ϩ V2 Ϫ Figure P1.66 (a) + vo – (b) Section 1.6: Frequency Response of Amplifiers 1.67 Use the voltage-divider rule to derive the transfer functions T (s) ≡ Vo(s)/Vi(s) of the circuits shown in Fig. 1.22, and show that the transfer functions are of the form given at the top of Table 1.2. Figure P1.65 1.66 Any linear two-port network including linear amplifiers can be represented by one of four possible parameter sets, given in Appendix C. For the voltage amplifier, the most convenient representation is in terms of the g parameters. If the amplifier input port is labeled as port 1 and the output port as port 2, its g-parameter representation is described by the two equations: 1.68 Figure P1.68 shows a signal source connected to the input of an amplifier. Here Rs is the source resistance, and Ri and Ci are the input resistance and input capacitance, respectively, of the amplifier. Derive an expression for Vi(s)/Vs(s), and show that it is of the low-pass STC type. Find the 3-dB frequency for the case Rs = 10 k , Ri = 40 k , and Ci = 5 pF. Rs I1 = g11V1 + g12I2 V2 = g21V1 + g22I2 Vs ϩ Ϫ ϩ Ri Ci Vi Ϫ Figure P1.66 shows an equivalent circuit representation of these two equations. By comparing this equivalent circuit Figure P1.68 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 1 PROBLEMS Problems 55 1.69 For the circuit shown in Fig. P1.69, find the transfer function T (s) = Vo(s)/Vi(s), and arrange it in the appropriate standard form from Table 1.2. Is this a high-pass or a low-pass network? What is its transmission at very high frequencies? [Estimate this directly, as well as by letting s → ∞ in your expression for T (s).] What is the corner frequency ω0? For R1 = 10 k , R2 = 40 k , and C = 1 μF, find f0. What is the value of |T ( jωo)|? R1 Vi ϩ Ϫ C ϩ R2 Vo 1.71 Measurement of the frequency response of an amplifier yields the data in the following table: f (Hz) 0 100 1000 104 105 | T | (dB) 40 40 37 20 0 ∠T (◦) 0 0 −45 Provide plausible approximate values for the missing entries. Also, sketch and clearly label the magnitude frequency response (i.e., provide a Bode plot) for this amplifier. Ϫ 1.72 Measurement of the frequency response of an amplifier yields the data in the following table: Figure P1.69 f (Hz) 10 102 103 104 105 106 107 | T | (dB) 0 20 37 40 37 20 0 D 1.70 It is required to couple a voltage source Vs with a resistance Rs to a load RL via a capacitor C. Derive an expression for the transfer function from source to load (i.e., Vl/Vs), and show that it is of the high-pass STC type. For Rs = 5 k and RL = 20 k , find the smallest coupling capacitor that will result in a 3-dB frequency no greater than 100 Hz. Provide approximate plausible values for the missing table entries. Also, sketch and clearly label the magnitude frequency response (Bode plot) of this amplifier. 1.73 The unity-gain voltage amplifiers in the circuit of Fig. P1.73 have infinite input resistances and zero output Vi Vo Figure P1.73 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 1 PROBLEMS 56 Chapter 1 Signals and Amplifiers resistances and thus function as perfect buffers. Furthermore, assume that their gain is frequency independent. Convince yourself that the overall gain Vo/Vi will drop by 3 dB below the value at dc at the frequency for which the gain of each RC circuit is 1.0 dB down. What is that frequency in terms of CR? 1.74 A manufacturing error causes an internal node of a high-frequency amplifier whose The´venin-equivalent node resistance is 100 k to be accidentally shunted to ground by a capacitor (i.e., the node is connected to ground through a capacitor). If the measured 3-dB bandwidth of the amplifier is reduced from the expected 5 MHz to 100 kHz, estimate the value of the shunting capacitor. If the original cutoff frequency can be attributed to a small parasitic capacitor at the same internal node (i.e., between the node and ground), what would you estimate it to be? D *1.75 A designer wishing to lower the overall upper 3-dB frequency of a three-stage amplifier to 10 kHz considers shunting one of two nodes: Node A, between the output of the first stage and the input of the second stage, and Node B, between the output of the second stage and the input of the third stage, to ground with a small capacitor. While measuring the overall frequency response of the amplifier, she connects a capacitor of 1 nF, first to node A and then to node B, lowering the 3-dB frequency from 3 MHz to 200 kHz and 20 kHz, respectively. If she knows that each amplifier stage has an input resistance of 100 k , what output resistance must the driving stage have at node A? At node B? What capacitor value should she connect to which node to solve her design problem most economically? D 1.76 An amplifier with an input resistance of 100 k and an output resistance of 1 k is to be capacitor-coupled to a 10-k source and a 1-k load. Available capacitors have values only of the form 1 × 10−n F. What are the values of the smallest capacitors needed to ensure that the corner frequency associated with each is less than 100 Hz? What actual corner frequencies result? For the situation in which the basic amplifier has an open-circuit voltage gain (Av o) of 100 V/V, find an expression for T (s) = Vo(s)/Vs(s). *1.77 A voltage amplifier has the transfer function Av = 1000 f 102 1+j 1+ 105 jf Using the Bode plots for low-pass and high-pass STC networks (Figs. 1.23 and 1.24), sketch a Bode plot for |Av|. Give approximate values for the gain magnitude at f = 10 Hz, 102 Hz, 103 Hz, 104 Hz, 105 Hz, 106 Hz, 107 Hz, and 108 Hz. Find the bandwidth of the amplifier (defined as the frequency range over which the gain remains within 3 dB of the maximum value). *1.78 For the circuit shown in Fig. P1.78, first evaluate Ti(s) = Vi(s)/Vs(s) and the corresponding cutoff (corner) frequency. Second, evaluate To(s) = Vo(s)/Vi(s) and the corresponding cutoff frequency. Put each of the transfer functions in the standard form (see Table 1.2), and combine them to form the overall transfer function, T (s) = Ti(s) × To(s). Provide a Bode magnitude plot for |T ( jω)|. What is the bandwidth between 3-dB cutoff points? R1 100 k⍀ ϩ Vs ϩ Ϫ C1 Vi 10 pF Ϫ Figure P1.78 C2 100 nF GmVi R2 100 k⍀ Gm ϭ 100 mAրV ϩ R3 Vo 10 k⍀ Ϫ = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 1 PROBLEMS Problems 57 D **1.79 A transconductance amplifier having the equiva- lent circuit shown in Table 1.1 is fed with a voltage source Vs having a source resistance Rs, and its output is connected to a load consisting of a resistance RL in parallel with a capacitance CL. For given values of Rs, RL, and CL, it is required to specify the values of the amplifier parameters Ri, Gm, and Ro to meet the following design constraints: (a) At most, x% of the input signal is lost in coupling the signal source to the amplifier (i.e., Vi ≥ [1 − (x/100)]Vs). (b) The 3-dB frequency of the amplifier is equal to or greater than a specified value f3 dB. (c) The dc gain Vo/Vs is equal to or greater than a specified value A0. Show that these constraints can be met by selecting 100 Ri ≥ x − 1 Rs Ro ≤ 1 2π f3dBCL − (1/RL) Gm ≥ A0/[1 − (x/100)] (RL Ro) Find Ri, Ro, and Gm for Rs = 10 k , x = 10%, A0 = 100 V/V, RL = 10 k , CL = 20 pF, and f3 dB = 2 MHz. *1.80 Use the voltage-divider rule to find the transfer function Vo(s)/Vi(s) of the circuit in Fig. P1.80. Show that the transfer function can be made independent of frequency if the condition C1R1 = C2R2 applies. Under this condition the circuit is called a compensated attenuator and is frequently employed in the design of oscilloscope probes. Find the transmission of the compensated attenuator in terms of R1 and R2. R1 Vi ϩ Ϫ R2 C1 ϩ C2 Vo Ϫ Figure P1.80 *1.81 An amplifier with a frequency response of the type shown in Fig. 1.21 is specified to have a phase shift of magnitude no greater than 5.7° over the amplifier bandwidth, which extends from 100 Hz to 1 kHz. It has been found that the gain falloff at the low-frequency end is determined by the response of a high-pass STC circuit and that at the high-frequency end it is determined by a low-pass STC circuit. What do you expect the corner frequencies of these two circuits to be? What is the drop in gain in decibels (relative to the maximum gain) at the two frequencies that define the amplifier bandwidth? What are the frequencies at which the drop in gain is 3 dB? (Hint: Refer to Figs. 1.23 and 1.24.) = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 2 Operational Amplifiers Introduction 59 2.1 The Ideal Op Amp 60 2.2 The Inverting Configuration 64 2.3 The Noninverting Configuration 73 2.4 Difference Amplifiers 77 2.5 Integrators and Differentiators 87 2.6 DC Imperfections 96 2.7 Effect of Finite Open-Loop Gain and Bandwidth on Circuit Performance 105 2.8 Large-Signal Operation of Op Amps 110 Summary 115 Problems 116 IN THIS CHAPTER YOU WILL LEARN 1. The terminal characteristics of the ideal op amp. 2. How to analyze circuits containing op amps, resistors, and capacitors. 3. How to use op amps to design amplifiers having precise characteristics. 4. How to design more sophisticated op-amp circuits, including summing amplifiers, instrumentation amplifiers, integrators, and differentiators. 5. Important nonideal characteristics of op amps and how these limit the performance of basic op-amp circuits. Introduction Having learned basic amplifier concepts and terminology, we are now ready to undertake the study of a circuit building block of universal importance: the operational amplifier (op amp). Op amps have been in use for a long time, their initial applications being primarily in the areas of analog computation and sophisticated instrumentation. Early op amps were constructed from discrete components (vacuum tubes and then transistors, and resistors), and their cost was prohibitively high (tens of dollars). In the mid-1960s the first integrated-circuit (IC) op amp was produced. This unit (the μA 709) was made up of a relatively large number of transistors and resistors all on the same silicon chip. Although its characteristics were poor (by today’s standards) and its price was still quite high, its appearance signaled a new era in electronic circuit design. Electronics engineers started using op amps in large quantities, which caused their price to drop dramatically. They also demanded better-quality op amps. Semiconductor manufacturers responded quickly, and within the span of a few years, high-quality op amps became available at extremely low prices (tens of cents) from a large number of suppliers. One of the reasons for the popularity of the op amp is its versatility. As we will shortly see, one can do almost anything with op amps! Equally important is the fact that the IC op amp has characteristics that closely approach the assumed ideal. This implies that it is quite easy to design circuits using the IC op amp. Also, op-amp circuits work at performance levels that are quite close to those predicted theoretically. It is for this reason that we are studying op amps at this early stage. It is expected that by the end of this chapter the reader should be able to successfully design nontrivial circuits using op amps. As already implied, an IC op amp is made up of a large number (about 20) of transistors together with resistors, and (usually) one capacitor connected in a rather complex circuit. Since 59 60 Chapter 2 Operational Amplifiers we have not yet studied transistor circuits, the circuit inside the op amp will not be discussed in this chapter. Rather, we will treat the op amp as a circuit building block and study its terminal characteristics and its applications. This approach is quite satisfactory in many op-amp applications. Nevertheless, for the more difficult and demanding applications it is quite useful to know what is inside the op-amp package. This topic will be studied in Chapter 13. More advanced applications of op amps will appear in later chapters. 2.1 The Ideal Op Amp 2.1.1 The Op-Amp Terminals From a signal point of view the op amp has three terminals: two input terminals and one output terminal. Figure 2.1 shows the symbol we shall use to represent the op amp. Terminals 1 and 2 are input terminals, and terminal 3 is the output terminal. As explained in Section 1.4, amplifiers require dc power to operate. Most IC op amps require two dc power supplies, as shown in Fig. 2.2. Two terminals, 4 and 5, are brought out of the op-amp package and connected to a positive voltage VCC and a negative voltage −VEE, respectively. In Fig. 2.2(b) we explicitly show the two dc power supplies as batteries with a common ground. It is interesting to note that the reference grounding point in op-amp circuits is just the common terminal of the two power supplies; that is, no terminal of the op-amp package is physically connected to ground. In what follows we will not, for simplicity, explicitly show the op-amp power supplies. Figure 2.1 Circuit symbol for the op amp. VCC VCC VEE ϪVEE Figure 2.2 The op amp shown connected to dc power supplies. 2.1 The Ideal Op Amp 61 In addition to the three signal terminals and the two power-supply terminals, an op amp may have other terminals for specific purposes. These other terminals can include terminals for frequency compensation and terminals for offset nulling; both functions will be explained in later sections. EXERCISE 2.1 What is the minimum number of terminals required by a single op amp? What is the minimum number of terminals required on an integrated-circuit package containing four op amps (called a quad op amp)? Ans. 5; 14 2.1.2 Function and Characteristics of the Ideal Op Amp We now consider the circuit function of the op amp. The op amp is designed to sense the difference between the voltage signals applied at its two input terminals (i.e., the quantity v2 − v1), multiply this by a number A, and cause the resulting voltage A(v2 − v1) to appear at output terminal 3. Thus v3 = A(v2 − v1). Here it should be emphasized that when we talk about the voltage at a terminal we mean the voltage between that terminal and ground; thus v1 means the voltage applied between terminal 1 and ground. The ideal op amp is not supposed to draw any input current; that is, the signal current into terminal 1 and the signal current into terminal 2 are both zero. In other words, the input impedance of an ideal op amp is supposed to be infinite. How about the output terminal 3? This terminal is supposed to act as the output terminal of an ideal voltage source. That is, the voltage between terminal 3 and ground will always be equal to A(v2 − v1), independent of the current that may be drawn from terminal 3 into a load impedance. In other words, the output impedance of an ideal op amp is supposed to be zero. Putting together all of the above, we arrive at the equivalent circuit model shown in Fig. 2.3. Note that the output is in phase with (has the same sign as) v2 and is out of phase with (has the opposite sign of) v1. For this reason, input terminal 1 is called the inverting input terminal and is distinguished by a “−” sign, while input terminal 2 is called the noninverting input terminal and is distinguished by a “+” sign. As can be seen from the above description, the op amp responds only to the difference signal v2 − v1 and hence ignores any signal common to both inputs. That is, if v1 = v2 = 1 V, then the output will (ideally) be zero. We call this property common-mode rejection, and we conclude that an ideal op amp has zero common-mode gain or, equivalently, infinite common-mode rejection. We will have more to say about this point later. For the time being note that the op amp is a differential-input, single-ended-output amplifier, with the latter term referring to the fact that the output appears between terminal 3 and ground.1 1Some op amps are designed to have differential outputs. This topic will not be discussed in this book. Rather, we confine ourselves here to single-ended-output op amps, which constitute the vast majority of commercially available op amps. 62 Chapter 2 Operational Amplifiers Inverting input Output Noninverting input Figure 2.3 Equivalent circuit of the ideal op amp. Furthermore, gain A is called the differential gain, for obvious reasons. Perhaps not so obvious is another name that we will attach to A: the open-loop gain. The reason for this name will become obvious later on when we “close the loop” around the op amp and define another gain, the closed-loop gain. An important characteristic of op amps is that they are direct-coupled or dc amplifiers, where dc stands for direct-coupled (it could equally well stand for direct current, since a direct-coupled amplifier is one that amplifies signals whose frequency is as low as zero). The fact that op amps are direct-coupled devices will allow us to use them in many important applications. Unfortunately, though, the direct-coupling property can cause some serious practical problems, as will be discussed in a later section. How about bandwidth? The ideal op amp has a gain A that remains constant down to zero frequency and up to infinite frequency. That is, ideal op amps will amplify signals of any frequency with equal gain, and are thus said to have infinite bandwidth. We have discussed all of the properties of the ideal op amp except for one, which in fact is the most important. This has to do with the value of A. The ideal op amp should have a gain A whose value is very large and ideally infinite. One may justifiably ask: If the gain A is infinite, how are we going to use the op amp? The answer is very simple: In almost all applications the op amp will not be used alone in a so-called open-loop configuration. Rather, we will use other components to apply feedback to close the loop around the op amp, as will be illustrated in detail in Section 2.2. For future reference, Table 2.1 lists the characteristics of the ideal op amp. Table 2.1 Characteristics of the Ideal Op Amp 1. Infinite input impedance 2. Zero output impedance 3. Zero common-mode gain or, equivalently, infinite common-mode rejection 4. Infinite open-loop gain A 5. Infinite bandwidth 2.1 The Ideal Op Amp 63 2.1.3 Differential and Common-Mode Signals The differential input signal vId is simply the difference between the two input signals v1 and v2; that is, vId = v2 − v1 (2.1) The common-mode input signal vIcm is the average of the two input signals v1 and v2; namely, v Icm = 1 2 (v 1 + v2) (2.2) Equations (2.1) and (2.2) can be used to express the input signals v1 and v2 in terms of their differential and common-mode components as follows: v1 = vIcm − vId /2 (2.3) and v2 = vIcm + vId /2 These equations can in turn lead to the pictorial representation in Fig. 2.4. (2.4) 1 1 v1 ϩ Ϫ Ϫ ϩ vIdր2 2 v2 ϩ Ϫ vIcm ϩ Ϫ Ϫ ϩ vIdր2 2 Figure 2.4 Representation of the signal sources v1 and v2 in terms of their differential and common-mode components. EXERCISES 2.2 Consider an op amp that is ideal except that its open-loop gain A = 103. The op amp is used in a feedback circuit, and the voltages appearing at two of its three signal terminals are measured. In each of the following cases, use the measured values to find the expected value of the voltage at the third terminal. Also give the differential and common-mode input signals in each case. (a) v2 = 0 V and v3 = 2 V; (b) v2 = +5 V and v3 = −10 V; (c) v1 = 1.002 V and v2 = 0.998 V; (d) v1 = −3.6 V and v3 = −3.6 V. Ans. (a) v1 = −0.002 V, vId = 2 mV, vIcm = −1 mV; (b) v1 = +5.01 V, vId = −10 mV, vIcm = 5.005 5 V; (c) v3 = −4 V, vId = −4 mV, vIcm = 1 V; (d) v2 = −3.6036 V, vId = −3.6 mV, vIcm −3.6 V 64 Chapter 2 Operational Amplifiers 2.3 The internal circuit of a particular op amp can be modeled by the circuit shown in Fig. E2.3. Express v3 as a function of v1 and v2. For the case Gm = 10 mA/V, R = 10 k , and μ = 100, find the value of the open-loop gain A. Ans. v3 = μGmR(v2 − v1); A = 10, 000 V/V or 80 dB Figure E2.3 2.2 The Inverting Configuration As mentioned above, op amps are not used alone; rather, the op amp is connected to passive components in a feedback circuit. There are two such basic circuit configurations employing an op amp and two resistors: the inverting configuration, which is studied in this section, and the noninverting configuration, which we shall study in the next section. Figure 2.5 shows the inverting configuration. It consists of one op amp and two resistors R1 and R2. Resistor R2 is connected from the output terminal of the op amp, terminal 3, back to the inverting or negative input terminal, terminal 1. We speak of R2 as applying negative feedback; if R2 were connected between terminals 3 and 2 we would have called this positive feedback. Note also that R2 closes the loop around the op amp. In addition to adding R2, we have grounded terminal 2 and connected a resistor R1 between terminal 1 and an input signal source 2.2 The Inverting Configuration 65 Figure 2.5 The inverting closed-loop configuration. with a voltage vI. The output of the overall circuit is taken at terminal 3 (i.e., between terminal 3 and ground). Terminal 3 is, of course, a convenient point from which to take the output, since the impedance level there is ideally zero. Thus the voltage vO will not depend on the value of the current that might be supplied to a load impedance connected between terminal 3 and ground. 2.2.1 The Closed-Loop Gain We now wish to analyze the circuit in Fig. 2.5 to determine the closed-loop gain G, defined as G ≡ vO vI We will do so assuming the op amp to be ideal. Figure 2.6(a) shows the equivalent circuit, and the analysis proceeds as follows: The gain A is very large (ideally infinite). If we assume that the circuit is “working” and producing a finite output voltage at terminal 3, then the voltage between the op-amp input terminals should be negligibly small and ideally zero. Specifically, if we call the output voltage vO, then, by definition, v2 − v1 = vO A =0 It follows that the voltage at the inverting input terminal (v1) is given by v1 = v2. That is, because the gain A approaches infinity, the voltage v1 approaches and ideally equals v2. We speak of this as the two input terminals “tracking each other in potential.” We also speak of a “virtual short circuit” that exists between the two input terminals. Here the word virtual should be emphasized, and one should not make the mistake of physically shorting terminals 1 and 2 together while analyzing a circuit. A virtual short circuit means that whatever voltage is at 2 will automatically appear at 1 because of the infinite gain A. But terminal 2 happens to be connected to ground; thus v2 = 0 and v1 = 0. We speak of terminal 1 as being a virtual ground—that is, having zero voltage but not physically connected to ground. Now that we have determined v1 we are in a position to apply Ohm’s law and find the current i1 through R1 (see Fig. 2.6) as follows: i1 = vI − v1 R1 = vI − 0 R1 = vI R1 Where will this current go? It cannot go into the op amp, since the ideal op amp has an infinite input impedance and hence draws zero current. It follows that i1 will have to flow through R2 to the low-impedance terminal 3. We can then apply Ohm’s law to R2 and determine vO; that is, vO = v1 − i1R2 = 0 − vI R1 R2 Thus, vO = − R2 vI R1 66 Chapter 2 Operational Amplifiers 5 3 4 Ϫ 1 ϩ 6 2 Figure 2.6 Analysis of the inverting configuration. The circled numbers indicate the order of the analysis steps. which is the required closed-loop gain. Figure 2.6(b) illustrates these steps and indicates by the circled numbers the order in which the analysis is performed. We thus see that the closed-loop gain is simply the ratio of the two resistances R2 and R1. The minus sign means that the closed-loop amplifier provides signal inversion. Thus if R2/R1 = 10 and we apply at the input (vI ) a sine-wave signal of 1 V peak-to-peak, then the output vO will be a sine wave of 10 V peak-to-peak and phase-shifted 180° with respect to the input sine wave. Because of the minus sign associated with the closed-loop gain, this configuration is called the inverting configuration. 2.2 The Inverting Configuration 67 The fact that the closed-loop gain depends entirely on external passive components (resistors R1 and R2) is very significant. It means that we can make the closed-loop gain as accurate as we want by selecting passive components of appropriate accuracy. It also means that the closed-loop gain is (ideally) independent of the op-amp gain. This is a dramatic illustration of negative feedback: We started out with an amplifier having very large gain A, and through applying negative feedback we have obtained a closed-loop gain R2/R1 that is much smaller than A but is stable and predictable. That is, we are trading gain for accuracy. 2.2.2 Effect of Finite Open-Loop Gain The points just made are more clearly illustrated by deriving an expression for the closed-loop gain under the assumption that the op-amp open-loop gain A is finite. Figure 2.7 shows the analysis. If we denote the output voltage vO, then the voltage between the two input terminals of the op amp will be vO/A. Since the positive input terminal is grounded, the voltage at the negative input terminal must be −vO/A. The current i1 through R1 can now be found from i1 = vI − (−vO/A) R1 = vI + vO/A R1 Figure 2.7 Analysis of the inverting configuration taking into account the finite open-loop gain of the op amp. The infinite input impedance of the op amp forces the current i1 to flow entirely through R2. The output voltage vO can thus be determined from vO = −vO A − i1R2 = −vO − A vI + vO/A R1 R2 Collecting terms, the closed-loop gain G is found as G ≡ vO = −R2/R1 vI 1 + (1 + R2/R1)/A (2.5) We note that as A approaches ∞, G approaches the ideal value of −R2/R1. Also, from Fig. 2.7 we see that as A approaches ∞, the voltage at the inverting input terminal approaches zero. This is the virtual-ground assumption we used in our earlier analysis when the op amp was 68 Chapter 2 Operational Amplifiers assumed to be ideal. Finally, note that Eq. (2.5) in fact indicates that to minimize the dependence of the closed-loop gain G on the value of the open-loop gain A, we should make 1 + R2 A R1 Example 2.1 Consider the inverting configuration with R1 = 1 k and R2 = 100 k , that is, having an ideal closed-loop gain of −100. (a) Find the closed-loop gain for the cases A = 103, 104, and 105. In each case determine the percentage error in the magnitude of G relative to the ideal value of R2/R1 (obtained with A = ∞). Also determine the voltage v1 that appears at the inverting input terminal when vI = 0.1 V. (b) If the open-loop gain A changes from 100,000 to 50,000 (i.e., drops by 50%), what is the corresponding percentage change in the magnitude of the closed-loop gain G? Solution (a) Substituting the given values in Eq. (2.5), we obtain the values given in the following table, where the percentage error e is defined as e ≡ |G| − (R2/R1) × 100 (R2/R1) The values of v1 are obtained from v1 = −vO/A = GvI /A with vI = −0.1 V. A |G| e v1 103 90.83 −9.17% −9.08 mV 104 99.00 −1.00% −0.99 mV 105 99.90 −0.10% −0.10 mV (b) Using Eq. (2.5), we find that for A = 50, 000, |G| = 99.80. Thus a −50% change in the open-loop gain results in a change in |G| from 99.90 to 99.80, which is only −0.1%! 2.2.3 Input and Output Resistances Assuming an ideal op amp with infinite open-loop gain, the input resistance of the closed-loop inverting amplifier of Fig. 2.5 is simply equal to R1. This can be seen from Fig. 2.6(b), where Ri ≡ vI i1 = vI vI /R1 = R1 Now recall that in Section 1.5 we learned that the amplifier input resistance forms a voltage divider with the resistance of the source that feeds the amplifier. Thus, to avoid the loss of signal strength, voltage amplifiers are required to have high input resistance. In the case of the inverting op-amp configuration we are studying, to make Ri high we should select a high value for R1. However, if the required gain R2/R1 is also high, then R2 could become impractically large 2.2 The Inverting Configuration 69 (e.g., greater than a few megohms). We may conclude that the inverting configuration suffers from a low input resistance. A solution to this problem is discussed in Example 2.2 below. Since the output of the inverting configuration is taken at the terminals of the ideal voltage source A(v2 − v1) (see Fig. 2.6a), it follows that the output resistance of the closed-loop amplifier is zero. Example 2.2 Assuming the op amp to be ideal, derive an expression for the closed-loop gain vO/vI of the circuit shown in Fig. 2.8. Use this circuit to design an inverting amplifier with a gain of 100 and an input resistance of 1 M . Assume that for practical reasons it is required not to use resistors greater than 1 M . Compare your design with that based on the inverting configuration of Fig. 2.5. 5 vx 4 7 x 3 2 Ϫ 1 ϩ 6 8 Figure 2.8 Circuit for Example 2.2. The circled numbers indicate the sequence of the steps in the analysis. Solution The analysis begins at the inverting input terminal of the op amp, where the voltage is v1 = −v O A = −v O ∞ =0 Here we have assumed that the circuit is “working” and producing a finite output voltage vO. Knowing v1, we can determine the current i1 as follows: i1 = vI − v1 R1 = vI − 0 R1 = vI R1 Since zero current flows into the inverting input terminal, all of i1 will flow through R2, and thus i2 = i1 = vI R1 Now we can determine the voltage at node x: vx = v1 − i2 R2 = 0 − vI R1 R2 = − R2 R1 vI 70 Chapter 2 Operational Amplifiers Example 2.2 continued This in turn enables us to find the current i3: i3 = 0 − vx R3 = R2 R1R3 v I Next, a node equation at x yields i4: i4 = i2 + i3 = vI R1 + R2 R1R3 v I Finally, we can determine vO from vO = vx − i4R4 Thus the voltage gain is given by = − R2 R1 vI − vI R1 + R2 R1R3 v I R4 which can be written in the form vO = − R2 + R4 1 + R2 vI R1 R1 R3 vO = − R2 1 + R4 + R4 vI R1 R2 R3 Now, since an input resistance of 1 M is required, we select R1 = 1 M . Then, with the limitation of using resistors no greater than 1 M , the maximum value possible for the first factor in the gain expression is 1 and is obtained by selecting R2 = 1 M . To obtain a gain of −100, R3 and R4 must be selected so that the second factor in the gain expression is 100. If we select the maximum allowed (in this example) value of 1 M for R4, then the required value of R3 can be calculated to be 10.2 k . Thus this circuit utilizes three 1-M resistors and a 10.2-k resistor. In comparison, if the inverting configuration were used with R1 = 1 M we would have required a feedback resistor of 100 M , an impractically large value! Before leaving this example it is insightful to inquire into the mechanism by which the circuit is able to realize a large voltage gain without using large resistances in the feedback path. Toward that end, observe that because of the virtual ground at the inverting input terminal of the op amp, R2 and R3 are in effect in parallel. Thus, by making R3 lower than R2 by, say, a factor k (i.e., where k > 1), R3 is forced to carry a current k-times that in R2. Thus, while i2 = i1, i3 = ki1 and i4 = (k + 1)i1. It is the current multiplication by a factor of (k + 1) that enables a large voltage drop to develop across R4 and hence a large vO without using a large value for R4. Notice also that the current through R4 is independent of the value of R4. It follows that the circuit can be used as a current amplifier as shown in Fig. 2.9. i2 ϭ iI R2 i4 R4 v1 ϭ 0 iI R3 i3 ϭ R2 R3 iI Ϫ ϩ ΂ ΃ i4 ϭ 1ϩ R2 R3 iI Figure 2.9 A current amplifier based on the circuit of Fig. 2.8. The amplifier delivers its output current to R4. It has a current gain of (1 + R2/R3), a zero input resistance, and an infinite output resistance. The load (R4), however, must be floating (i.e., neither of its two terminals can be connected to ground). 2.2 The Inverting Configuration 71 EXERCISES D2.4 Use the circuit of Fig. 2.5 to design an inverting amplifier having a gain of −10 and an input resistance of 100 k . Give the values of R1 and R2. Ans. R1 = 100 k ; R2 = 1 M 2.5 The circuit shown in Fig. E2.5(a) can be used to implement a transresistance amplifier (see Table 1.1 in Section 1.5). Find the value of the input resistance Ri, the transresistance Rm, and the output resistance Ro of the transresistance amplifier. If the signal source shown in Fig. E2.5(b) is connected to the input of the transresistance amplifier, find the amplifier output voltage. Ans. Ri = 0; Rm = −10 k ; Ro = 0; vO = −5 V Figure E2.5 2.6 For the circuit in Fig. E2.6 determine the values of v1, i1, i2, vO, iL, and iO. Also determine the voltage gain vO/vI , current gain iL/iI , and power gain PO/PI . Ans. 0 V; 1 mA; 1 mA; −10 V; −10 mA; −11 mA; −10 V/V (20 dB), −10 A/A (20 dB); 100 W/W (20 dB) i1 1V ϩ Ϫ i2 10 k⍀ 1 k⍀ Ϫ iO v1 ϩ iL vO 1 k⍀ Figure E2.6 2.2.4 An Important Application—The Weighted Summer A very important application of the inverting configuration is the weighted-summer circuit shown in Fig. 2.10. Here we have a resistance Rf in the negative-feedback path (as before), but we have a number of input signals v1, v2, . . . , vn each applied to a corresponding resistor 72 Chapter 2 Operational Amplifiers R1, R2, . . . , Rn, which are connected to the inverting terminal of the op amp. From our previous discussion, the ideal op amp will have a virtual ground appearing at its negative input terminal. Ohm’s law then tells us that the currents i1, i2, . . . , in are given by i1 = v1 R1 , i2 = v2 R2 , ..., in = vn Rn 0 Figure 2.10 A weighted summer. All these currents sum together to produce the current i, i = i1 + i2 + · · · + in (2.6) which will be forced to flow through Rf (since no current flows into the input terminals of an ideal op amp). The output voltage vO may now be determined by another application of Ohm’s law, vO = 0 − iRf = −iRf Thus, vO = − Rf R1 v1 + Rf R2 v2 + · · · + Rf Rn vn (2.7) That is, the output voltage is a weighted sum of the input signals v1, v2, . . . , vn. This circuit is therefore called a weighted summer. Note that each summing coefficient may be independently adjusted by adjusting the corresponding “feed-in” resistor (R1 to Rn). This nice property, which greatly simplifies circuit adjustment, is a direct consequence of the virtual ground that exists at the inverting op-amp terminal. As the reader will soon come to appreciate, virtual grounds are extremely “handy.” In the weighted summer of Fig. 2.10 all the summing coefficients must be of the same sign. The need occasionally arises for summing signals with opposite signs. Such a function can be implemented, however, using two op amps as shown in Fig. 2.11. Assuming ideal op amps, it can be easily shown that the output voltage is given by vO = v1 Ra R1 Rc Rb + v2 Ra R2 Rc Rb − v3 Rc R3 − v4 Rc R4 (2.8) Weighted summers are utilized in a variety of applications including in the design of audio systems, where they can be used in mixing signals originating from different musical instruments. 2.3 The Noninverting Configuration 73 Ra Rc R1 v1 Ϫ Rb R2 v2 ϩ R3 v3 Ϫ ϩ vO R4 v4 Figure 2.11 A weighted summer capable of implementing summing coefficients of both signs. EXERCISES D2.7 Design an inverting op-amp circuit to form the weighted sum vO of two inputs v1 and v2. It is required that vO = −(v1 + 5v2). Choose values for R1, R2, and Rf so that for a maximum output voltage of 10 V the current in the feedback resistor will not exceed 1 mA. Ans. A possible choice: R1 = 10 k , R2 = 2 k , and Rf = 10 k D2.8 Use the idea presented in Fig. 2.11 to design a weighted summer that provides vO = 2v1 + v2 − 4v3 Ans. A possible choice: R1 = 5 k , R2 = 10 k , Ra = 10 k , Rb = 10 k , R3 = 2.5 k , Rc = 10 k 2.3 The Noninverting Configuration The second closed-loop configuration we shall study is shown in Fig. 2.12. Here the input signal vI is applied directly to the positive input terminal of the op amp while one terminal of R1 is connected to ground. 2.3.1 The Closed-Loop Gain Analysis of the noninverting circuit to determine its closed-loop gain (vO/vI) is illustrated in Fig. 2.13. Again the order of the steps in the analysis is indicated by circled numbers. Assuming that the op amp is ideal with infinite gain, a virtual short circuit exists between its two input terminals. Hence the difference input signal is v Id = vO A = 0 for A = ∞ Thus the voltage at the inverting input terminal will be equal to that at the noninverting input terminal, which is the applied voltage vI. The current through R1 can then be determined as vI/R1. Because of the infinite input impedance of the op amp, this current will flow through R2, as shown in Fig. 2.13. Now the output voltage can be determined from vO = vI + vI R1 R2 74 Chapter 2 Operational Amplifiers 5 vI R1 R2 3 vI R1 R1 2 vI Ϫ 1 vId ϭ Ϫ 0V ϩ 0 4 ϩ ϩ vI Ϫ Figure 2.12 The noninverting configuration. vO ϭ vI ϩ Rv1I R2 ϭ vI 1 ϩ R2 R1 6 ϩ vO Ϫ Figure 2.13 Analysis of the noninverting circuit. The sequence of the steps in the analysis is indicated by the circled numbers. which yields vO = 1 + RI vI R2 (2.9) Further insight into the operation of the noninverting configuration can be obtained by considering the following: Since the current into the op-amp inverting input is zero, the circuit composed of R1 and R2 acts in effect as a voltage divider feeding a fraction of the output voltage back to the inverting input terminal of the op amp; that is, v1 = vO R1 R1 + R2 (2.10) Then the infinite op-amp gain and the resulting virtual short circuit between the two input terminals of the op amp forces this voltage to be equal to that applied at the positive input terminal; thus, vO R1 R1 + R2 = vI which yields the gain expression given in Eq. (2.9). This is an appropriate point to reflect further on the action of the negative feedback present in the noninverting circuit of Fig. 2.12. Let vI increase. Such a change in vI will cause vId to increase, and vO will correspondingly increase as a result of the high (ideally infinite) gain of the op amp. However, a fraction of the increase in vO will be fed back to the inverting input terminal of the op amp through the (R1, R2) voltage divider. The result of this feedback will be to counteract the increase in vId, driving vId back to zero, albeit at a higher value of vO that corresponds to the increased value of vI. This degenerative action of negative feedback gives it the alternative name degenerative feedback. Finally, note that the argument above applies equally well if vI decreases. A formal and detailed study of feedback is presented in Chapter 11. 2.3 The Noninverting Configuration 75 2.3.2 Effect of Finite Open-Loop Gain As we have done for the inverting configuration, we now consider the effect of the finite op-amp open-loop gain A on the gain of the noninverting configuration. Assuming the op amp to be ideal except for having a finite open-loop gain A, it can be shown that the closed-loop gain of the noninverting amplifier circuit of Fig. 2.12 is given by G ≡ vO vI = 1 + (R2/R1) 1 + 1 + (R2/R1) A (2.11) Observe that the denominator is identical to that for the case of the inverting configuration (Eq. 2.5). This is no coincidence; it is a result of the fact that both the inverting and the noninverting configurations have the same feedback loop, which can be readily seen if the input signal source is eliminated (i.e., short-circuited). The numerators, however, are different, for the numerator gives the ideal or nominal closed-loop gain (−R2/R1 for the inverting configuration, and 1 + R2/R1 for the noninverting configuration). Finally, we note (with reassurance) that the gain expression in Eq. (2.11) reduces to the ideal value for A = ∞. In fact, it approximates the ideal value for A 1 + R2 R1 This is the same condition as in the inverting configuration, except that here the quantity on the right-hand side is the nominal closed-loop gain. The expressions for the actual and ideal values of the closed-loop gain G in Eqs. (2.11) and (2.9), respectively, can be used to determine the percentage error in G resulting from the finite op-amp gain A as Percent gain error = − 1 + (R2/R1) × 100 A + 1 + (R2/R1) (2.12) Thus, as an example, if an op amp with an open-loop gain of 1000 is used to design a noninverting amplifier with a nominal closed-loop gain of 10, we would expect the closed-loop gain to be about 1% below the nominal value. 2.3.3 Input and Output Resistance The gain of the noninverting configuration is positive—hence the name noninverting. The input impedance of this closed-loop amplifier is ideally infinite, since no current flows into the positive input terminal of the op amp. The output of the noninverting amplifier is taken at the terminals of the ideal voltage source A(v2 − v1) (see the op-amp equivalent circuit in Fig. 2.3), and thus the output resistance of the noninverting configuration is zero. 2.3.4 The Voltage Follower The property of high input impedance is a very desirable feature of the noninverting configuration. It enables using this circuit as a buffer amplifier to connect a source with a high impedance to a low-impedance load. We discussed the need for buffer amplifiers in Section 1.5. In many applications the buffer amplifier is not required to provide any voltage gain; rather, it is used mainly as an impedance transformer or a power amplifier. In such cases we may make R2 = 0 and R1 = ∞ to obtain the unity-gain amplifier shown in Fig. 2.14(a). This circuit is commonly referred to as a voltage follower, since the output “follows” the input. In the ideal case, vO = vI , Rin = ∞, Rout = 0, and the follower has the equivalent circuit shown in Fig. 2.14(b). 76 Chapter 2 Operational Amplifiers Ϫ ϩ vI ϩ Ϫ ϩ vO ϭ vI Ϫ ϩ ϩ vI ϩ Ϫ 1 ϫ vI vO Ϫ Ϫ (a) (b ) Figure 2.14 (a) The unity-gain buffer or follower amplifier. (b) Its equivalent circuit model. Since in the voltage-follower circuit the entire output is fed back to the inverting input, the circuit is said to have 100% negative feedback. The infinite gain of the op amp then acts to make vId = 0 and hence vO = vI . Observe that the circuit is elegant in its simplicity! Since the noninverting configuration has a gain greater than or equal to unity, depending on the choice of R2/R1, some prefer to call it “a follower with gain.” EXERCISES 2.9 Use the superposition principle to find the output voltage of the circuit shown in Fig. E2.9. Ans. vO = 6v1 + 4v2 Figure E2.9 2.10 If in the circuit of Fig. E2.9 the 1-k resistor is disconnected from ground and connected to a third signal source v3, use superposition to determine vO in terms of v1, v2, and v3. Ans. vO = 6v1 + 4v2 − 9v3 D2.11 Design a noninverting amplifier with a gain of 2. At the maximum output voltage of 10 V the current in the voltage divider is to be 10 μA. Ans. R1 = R2 = 0.5 M 2.12 (a) Show that if the op amp in the circuit of Fig. 2.12 has a finite open-loop gain A, then the closed-loop gain is given by Eq. (2.11). (b) For R1 = 1 k and R2 = 9 k find the percentage deviation e of the closed-loop gain from the ideal value of (1 + R2/R1) for the cases A = 103, 104, and 105. For vI = 1 V, find in each case the voltage between the two input terminals of the op amp. Ans. e = −1%, −0.1%, −0.01%; v2 − v1 = 9.9 mV, 1 mV, 0.1 mV 2.4 Difference Amplifiers 77 2.13 For the circuit in Fig. E2.13 find the values of iI , v1, i1, i2, vO, iL, and iO. Also find the voltage gain vO/vI , the current gain iL/iI , and the power gain PL/PI . Ans. 0; 1 V; 1 mA; 1 mA; 10 V; 10 mA; 11 mA; 10 V/V (20 dB); ∞; ∞ i2 9 k⍀ i1 1 k⍀ Ϫ v1 ϩ iI vI ϭ 1 V ϩ Ϫ iO iL vO 1 k⍀ Figure E2.13 2.14 It is required to connect a transducer having an open-circuit voltage of 1 V and a source resistance of 1 M to a load of 1-k resistance. Find the load voltage if the connection is done (a) directly, and (b) through a unity-gain voltage follower. Ans. (a) 1 mV; (b) 1 V 2.4 Difference Amplifiers Having studied the two basic configurations of op-amp circuits together with some of their direct applications, we are now ready to consider a somewhat more involved but very important application. Specifically, we shall study the use of op amps to design difference or differential amplifiers.2 A difference amplifier is one that responds to the difference between the two signals applied at its input and ideally rejects signals that are common to the two inputs. The representation of signals in terms of their differential and common-mode components was given in Fig. 2.4. It is repeated here in Fig. 2.15 with slightly different symbols to serve as the input signals for the difference amplifiers we are about to design. Although ideally the difference amplifier will amplify only the differential input signal vId and reject completely the common-mode input signal vIcm, practical circuits will have an output voltage vO given by vO = Ad vId + AcmvIcm (2.13) where Ad denotes the amplifier differential gain and Acm denotes its common-mode gain (ideally zero). The efficacy of a differential amplifier is measured by the degree of its rejection of common-mode signals in preference to differential signals. This is usually quantified by a measure known as the common-mode rejection ratio (CMRR), defined as CMRR = 20 log |Ad | |Acm| (2.14) 2The terms difference and differential are usually used to describe somewhat different amplifier types. For our purposes at this point, the distinction is not sufficiently significant. We will be more precise near the end of this section. 78 Chapter 2 Operational Amplifiers vIcm ϩ Ϫ vI1 ϭ vIcm Ϫ vIdր2 Ϫ ϩ vIdր2 Ϫ ϩ vIdր2 vId ϭ vI2 Ϫ vI1 vIcm ϭ 1 2 (vI1 ϩ vI2) vI2 ϭ vIcm ϩ vIdր2 Figure 2.15 Representing the input signals to a differential amplifier in terms of their differential and common-mode components. The need for difference amplifiers arises frequently in the design of electronic systems, especially those employed in instrumentation. As a common example, consider a transducer providing a small (e.g., 1 mV) signal between its two output terminals while each of the two wires leading from the transducer terminals to the measuring instrument may have a large interference signal (e.g., 1 V) relative to the circuit ground. The instrument front end obviously needs a difference amplifier. Before we proceed any further we should address a question that the reader might have: The op amp is itself a difference amplifier; why not just use an op amp? The answer is that the very high (ideally infinite) gain of the op amp makes it impossible to use by itself. Rather, as we did before, we have to devise an appropriate feedback network to connect to the op amp to create a circuit whose closed-loop gain is finite, predictable, and stable. 2.4.1 A Single-Op-Amp Difference Amplifier Our first attempt at designing a difference amplifier is motivated by the observation that the gain of the noninverting amplifier configuration is positive, (1 + R2/R1), while that of the inverting configuration is negative, (−R2/R1). Combining the two configurations together is then a step in the right direction—namely, getting the difference between two input signals. Of course, we have to make the two gain magnitudes equal in order to reject common-mode signals. This, however, can be easily achieved by attenuating the positive input signal to reduce the gain of the positive path from (1 + R2/R1) to (R2/R1). The resulting circuit would then look like that shown in Fig. 2.16, where the attenuation in the positive input path is achieved by the voltage divider (R3, R4). The proper ratio of this voltage divider can be determined from which can be put in the form R4 1 + R2 = R2 R4 + R3 R1 R1 R4 = R2 R4 + R3 R2 + R1 This condition is satisfied by selecting R4 = R2 R3 R1 (2.15) 2.4 Difference Amplifiers 79 vI 1 vI 2 Figure 2.16 A difference amplifier. This completes our work. However, we have perhaps proceeded a little too fast! Let’s step back and verify that the circuit in Fig. 2.16 with R3 and R4 selected according to Eq. (2.15) does in fact function as a difference amplifier. Specifically, we wish to determine the output voltage vO in terms of vI1 and vI2. Toward that end, we observe that the circuit is linear, and thus we can use superposition. To apply superposition, we first reduce vI2 to zero—that is, ground the terminal to which vI2 is applied—and then find the corresponding output voltage, which will be due entirely to vI1. We denote this output voltage vO1. Its value may be found from the circuit in Fig. 2.17(a), which we recognize as that of the inverting configuration. The existence of R3 and R4 does not affect the gain expression, since no current flows through either of them. Thus, v O1 = − R2 R1 v I1 Next, we reduce vI1 to zero and evaluate the corresponding output voltage vO2. The circuit will now take the form shown in Fig. 2.17(b), which we recognize as the noninverting configuration with an additional voltage divider, made up of R3 and R4, connected to the input vI2. The output voltage vO2 is therefore given by v O2 = vI2 R3 R4 + R4 1 + R2 R1 = R2 R1 vI2 where we have utilized Eq. (2.15). The superposition principle tells us that the output voltage vO is equal to the sum of vO1 and vO2. Thus we have vO = R2 R1 (v I 2 − vI1) = R2 R1 v Id (2.16) Thus, as expected, the circuit acts as a difference amplifier with a differential gain Ad of Ad = R2 R1 (2.17) Of course this is predicated on the op amp being ideal and furthermore on the selection of R3 and R4 so that their ratio matches that of R1 and R2 (Eq. 2.15). To make this matching requirement a little easier to satisfy, we usually select R3 = R1 and R4 = R2 80 Chapter 2 Operational Amplifiers vI 1 vI 2 Figure 2.17 Application of superposition to the analysis of the circuit of Fig. 2.16. Let’s next consider the circuit with only a common-mode signal applied at the input, as shown in Fig. 2.18. The figure also shows some of the analysis steps. Thus, i1 = 1 R1 v Icm − R4 R4 + R3 v Icm = v Icm R4 R3 + R3 1 R1 The output voltage can now be found from (2.18) vO = R4 R4 + R3 v Icm − i2R2 Substituting i2 = i1 and for i1 from Eq. (2.18), vO = R4 R4 + R3 v Icm − R2 R1 R3 R4 + R3 v Icm = R4 R4 + R3 1 − R2 R3 R1 R4 v Icm Thus, Acm ≡ vO v Icm = R4 R4 + R3 1 − R2 R3 R1 R4 For the design with the resistor ratios selected according to Eq. (2.15), we obtain (2.19) Acm = 0 as expected. Note, however, that any mismatch in the resistance ratios can make Acm nonzero, and hence CMRR finite. In addition to rejecting common-mode signals, a difference amplifier is usually required to have a high input resistance. To find the input resistance between the two input terminals 2.4 Difference Amplifiers 81 i2 R2 i1 R1 Ϫ vO vIcm ϩ Ϫ R3 ΂ ΃ R4 R4 ϩ R3 vIcm ϩ R4 Figure 2.18 Analysis of the difference amplifier to determine its common-mode gain Acm ≡ vO/vIcm. (i.e., the resistance seen by vId), called the differential input resistance Rid, consider Fig. 2.19. Here we have assumed that the resistors are selected so that R3 = R1 and R4 = R2 Now Rid ≡ v Id iI Since the two input terminals of the op amp track each other in potential, we may write a loop equation and obtain Thus, vId = R1iI + 0 + R1iI Rid = 2R1 (2.20) Note that if the amplifier is required to have a large differential gain (R2/R1), then R1 of necessity will be relatively small and the input resistance will be correspondingly low, a drawback of this circuit. Another drawback of the circuit is that it is not easy to vary the differential gain of the amplifier. Both of these drawbacks are overcome in the instrumentation amplifier discussed next. I vId Rid I Figure 2.19 Finding the input resistance of the difference amplifier for the case R3 = R1 and R4 = R2. 82 Chapter 2 Operational Amplifiers EXERCISES 2.15 Consider the difference-amplifier circuit of Fig. 2.16 for the case R1 = R3 = 2 k and R2 = R4 = 200 k . (a) Find the value of the differential gain Ad. (b) Find the value of the differential input resistance Rid and the output resistance Ro. (c) If the resistors have 1% tolerance (i.e., each can be within ±1% of its nominal value), use Eq. (2.19) to find the worst-case common-mode gain Acm and hence the corresponding value of CMRR. Ans. (a) 100 V/V (40 dB); (b) 4 k , 0 ; (c) 0.04 V/V, 68 dB D2.16 Find values for the resistances in the circuit of Fig. 2.16 so that the circuit behaves as a difference amplifier with an input resistance of 20 k and a gain of 10. Ans. R1 = R3 = 10 k ; R2 = R4 = 100 k 2.4.2 A Superior Circuit—The Instrumentation Amplifier The low-input-resistance problem of the difference amplifier of Fig. 2.16 can be solved by using voltage followers to buffer the two input terminals; that is, a voltage follower of the type in Fig. 2.14 is connected between each input terminal and the corresponding input terminal of the difference amplifier. However, if we are going to use two additional op amps, we should ask the question: Can we get more from them than just impedance buffering? An obvious answer would be that we should try to get some voltage gain. It is especially interesting that vI1 ϩ ΂ ΃ 1 ϩ R2 R1 vI1 A1 Ϫ R2 R4 R1 X R1 R2 R3 Ϫ A3 ϩ R3 ϩ R4 vO Ϫ Ϫ A2 vI2 ϩ ΂ ΃ 1 ϩ R2 R1 vI2 (a) Figure 2.20 A popular circuit for an instrumentation amplifier. (a) Initial approach to the circuit. (b) The circuit in (a) with the connection between node X and ground removed and the two resistors R1 and R1 lumped together. This simple wiring change dramatically improves performance. (c) Analysis of the circuit in (b) assuming ideal op amps. 2.4 Difference Amplifiers 83 vI1 ϩ A1 Ϫ R2 2R1 Ϫ R2 A2 vI2 ϩ R4 R3 Ϫ R3 A3 ϩ ϩ vO R4 Ϫ vI1 ϩϩ 0V A1 ϪϪ vI1 0 R2 Ϫ vId/2R1 (vI2 Ϫ vI1) ϭ vId ϩ 2R1 vId 2R1 vI2 0 ϪϪ R2 vId/2R1 0V A2 vI2 ϩϩ (b) vO1 R3 Ϫ vId ϩ 1ϩ 2R2 2R1 R3 vO2 R4 Ϫ A3 ϩ R4 ϩ vO ϭ R4 R3 1 ϩ R2 R1 vId Ϫ (c ) Figure 2.20 continued we can achieve this without compromising the high input resistance simply by using followers with gain rather than unity-gain followers. Achieving some or indeed the bulk of the required gain in this new first stage of the differential amplifier eases the burden on the difference amplifier in the second stage, leaving it to its main task of implementing the differencing function and thus rejecting common-mode signals. The resulting circuit is shown in Fig. 2.20(a). It consists of two stages in cascade. The first stage is formed by op amps A1 and A2 and their associated resistors, and the second stage is the by-now-familiar difference amplifier formed by op amp A3 and its four associated resistors. Observe that as we set out to do, each of A1 and A2 is connected in the noninverting configuration and thus realizes a gain of (1 + R2/R1). It follows that each of vI1 and vI2 is amplified by this factor, and the resulting amplified signals appear at the outputs of A1 and A2, respectively. The difference amplifier in the second stage operates on the difference signal (1 + R2/R1)(vI2 − vI1) = (1 + R2/R1)vId and provides at its output vO = R4 R3 1 + R2 R1 v Id 84 Chapter 2 Operational Amplifiers Thus the differential gain realized is Ad = R4 R3 1 + R2 R1 (2.21) The common-mode gain will be zero because of the differencing action of the second-stage amplifier. The circuit in Fig. 2.20(a) has the advantage of very high (ideally infinite) input resistance and high differential gain. Also, provided A1 and A2 and their corresponding resistors are matched, the two signal paths are symmetric—a definite advantage in the design of a differential amplifier. The circuit, however, has three major disadvantages: 1. The input common-mode signal vIcm is amplified in the first stage by a gain equal to that experienced by the differential signal vId. This is a very serious issue, for it could result in the signals at the outputs of A1 and A3 being of such large magnitudes that the op amps saturate (more on op-amp saturation in Section 2.8). But even if the op amps do not saturate, the difference amplifier of the second stage will now have to deal with much larger common-mode signals, with the result that the CMRR of the overall amplifier will inevitably be reduced. 2. The two amplifier channels in the first stage have to be perfectly matched, otherwise a spurious signal may appear between their two outputs. Such a signal would get amplified by the difference amplifier in the second stage. 3. To vary the differential gain Ad, two resistors have to be varied simultaneously, say the two resistors labeled R1. At each gain setting the two resistors have to be perfectly matched: a difficult task. All three problems can be solved with a very simple wiring change: Simply disconnect the node between the two resistors labeled R1, node X, from ground. The circuit with this small but functionally profound change is redrawn in Fig. 2.20(b), where we have lumped the two resistors (R1 and R1) together into a single resistor (2R1). Analysis of the circuit in Fig. 2.20(b), assuming ideal op amps, is straightforward, as is illustrated in Fig. 2.20(c). The key point is that the virtual short circuits at the inputs of op amps A1 and A2 cause the input voltages vI1 and vI2 to appear at the two terminals of resistor (2R1). Thus the differential input voltage vI2 − vI1 ≡ vId appears across 2R1 and causes a current i = vId/2R1 to flow through 2R1 and the two resistors labeled R2. This current in turn produces a voltage difference between the output terminals of A1 and A2 given by vO2 − vO1 = 1 + 2R2 2R1 v Id The difference amplifier formed by op amp A3 and its associated resistors senses the voltage difference (vO2 − vO1) and provides a proportional output voltage vO: vO = R4 R3 (v O2 − v O1 ) = R4 R3 1 + R2 R1 v Id 2.4 Difference Amplifiers 85 Thus the overall differential voltage-gain is given by Ad ≡ vO v Id = R4 R3 1 + R2 R1 (2.22) Observe that proper differential operation does not depend on the matching of the two resistors labeled R2. Indeed, if one of the two is of different value, say R2, the expression for Ad becomes Ad = R4 R3 1 + R2 + R2 2R1 (2.23) Consider next what happens when the two input terminals are connected together to a common-mode input voltage vIcm. It is easy to see that an equal voltage appears at the negative input terminals of A1 and A2, causing the current through 2R1 to be zero. Thus there will be no current flowing in the R2 resistors, and the voltages at the output terminals of A1 and A2 will be equal to the input (i.e., vIcm). Thus the first stage no longer amplifies vIcm; it simply propagates vIcm to its two output terminals, where they are subtracted to produce a zero common-mode output by A3. The difference amplifier in the second stage, however, now has a much improved situation at its input: The difference signal has been amplified by (1 + R2/R1) while the common-mode voltage remained unchanged. Finally, we observe from the expression in Eq. (2.22) that the gain can be varied by changing only one resistor, 2R1. We conclude that this is an excellent differential amplifier circuit and is widely employed as an instrumentation amplifier, that is, as the input amplifier used in a variety of electronic instruments. INTEGRATED INSTRUMENTATION AMPLIFIERS: The conventional combination of three op amps and a number of precision resistors to form an instrumentation amplifier is an extremely powerful tool for the design of instruments for many applications. While the earliest applications used separate op amps and discrete resistors, fully integrated versions incorporating most required components in a single integrated-circuit package are increasingly available from many manufacturers. Low-power versions of these units are extremely important in the design of portable, wearable, and implantable medical monitoring devices, such as wristband activity monitors. Example 2.3 Design the instrumentation amplifier circuit in Fig. 2.20(b) to provide a gain that can be varied over the range of 2 to 1000 utilizing a 100-k variable resistance (a potentiometer, or “pot” for short). Solution It is usually preferable to obtain all the required gain in the first stage, leaving the second stage to perform the task of taking the difference between the outputs of the first stage and thereby rejecting the common-mode signal. In other words, the second stage is usually designed for a gain of 1. Adopting this approach, we select 86 Chapter 2 Operational Amplifiers Example 2.3 continued all the second-stage resistors to be equal to a practically convenient value, say 10 k . The problem then reduces to designing the first stage to realize a gain adjustable over the range of 2 to 1000. Implementing 2R1 as the series combination of a fixed resistor R1f and the variable resistor R1v obtained using the 100-k pot (Fig. 2.21), we can write 1 + 2R2 = 2 to 1000 R1f + R1v Thus, 1 + 2R2 = 1000 R1f and 1+ 2R2 =2 R1f + 100 k These two equations yield R1f = 100.2 and R2 = 50.050 k . Other practical values may be selected; for instance, R1f = 100 and R2 = 49.9 k (both values are available as standard 1%-tolerance metal-film resistors; see Appendix J) results in a gain covering approximately the required range. R1f 2R1 100 k⍀ pot R1v Figure 2.21 To make the gain of the circuit in Fig. 2.20(b) variable, 2R1 is implemented as the series combination of a fixed resistor R1f and a variable resistor R1v . Resistor R1f ensures that the maximum available gain is limited. EXERCISE 2.17 Consider the instrumentation amplifier of Fig. 2.20(b) with a common-mode input voltage of +5 V (dc) and a differential input signal of 10-mV-peak sine wave. Let (2R1) = 1 k , R2 = 0.5 M , and R3 = R4 = 10 k . Find the voltage at every node in the circuit. Ans. vI1 = 5 − 0.005 sin ωt; vI2 = 5 + 0.005 sin ωt; v–(op amp A1) = 5 − 0.005 sin ωt; v–(op amp A2) = 5+ 0.005 sin ωt; vO1 = 5 − 5.005 sin ωt; vO2 = 5 + 5.005 sin ωt; v– (A3) = v+(A3) = 2.5 + 2.5025 sin ωt; vO = 10.01 sin ωt (all in volts) 2.5 Integrators and Differentiators 87 2.5 Integrators and Differentiators The op-amp circuit applications we have studied thus far utilized resistors in the op-amp feedback path and in connecting the signal source to the circuit, that is, in the feed-in path. As a result, circuit operation has been (ideally) independent of frequency. By allowing the use of capacitors together with resistors in the feedback and feed-in paths of op-amp circuits, we open the door to a very wide range of useful and exciting applications of the op amp. We begin our study of op-amp–RC circuits by considering two basic applications, namely, signal integrators and differentiators.3 2.5.1 The Inverting Configuration with General Impedances To begin with, consider the inverting closed-loop configuration with impedances Z1(s) and Z2(s) replacing resistors R1 and R2, respectively. The resulting circuit is shown in Fig. 2.22 and, for an ideal op amp, has the closed-loop gain or, more appropriately, the closed-loop transfer function Vo(s) = − Z2(s) Vi(s) Z1(s) (2.24) As explained in Section 1.6, replacing s by jω provides the transfer function for physical frequencies ω, that is, the transmission magnitude and phase for a sinusoidal input signal of frequency ω. Figure 2.22 The inverting configuration with general impedances in the feedback and the feed-in paths. 3At this point, a review of Section 1.6 would be helpful. Also, an important fact to remember: Passing a constant current I through a capacitor C for a time t causes a change of It to accumulate on the capacitor. Thus the capacitor voltage changes by V = Q/C = It/C; that is, the capacitor voltage increases linearly with time. 88 Chapter 2 Operational Amplifiers EARLY OP AMPS AND ANALOG COMPUTATION: In 1941, Karl D. Swartzel Jr. of Bell Labs patented “the summing amplifier,” a high-gain dc inverting amplifier, intended to be used with negative feedback. This precursor of the op amp used three vacuum tubes (the predecessor of the transistor) and ±350-V power supplies to achieve a gain of 90 dB. Though lacking a differential input, it provided the usual applications of summation, integration, and general filtering using convenient passive resistive and capacitive components. Soon after (1942), Loebe Julie, working with Professor John R. Regazzini at Columbia University, created a differential version, still using vacuum tubes. During World War II, these units were used extensively to provide analog computational functions in association with radar-directed antiaircraft firing control involving aircraft speed projection. In the early 1950s, driven by the demonstrated wartime success of op-amp-based computation, general-purpose commercial systems called “analog computers” began to appear. They consisted of a few dozen op amps and associated passive components, including potentiometers; the interconnections required for programming were achieved with plug boards. These computers were used to solve differential equations. Example 2.4 For the circuit in Fig. 2.23, derive an expression for the transfer function Vo(s)/Vi(s). Show that the transfer function is that of a low-pass STC circuit. By expressing the transfer function in the standard form shown in Table 1.2 on page 36, find the dc gain and the 3-dB frequency. Design the circuit to obtain a dc gain of 40 dB, a 3-dB frequency of 1 kHz, and an input resistance of 1 k . At what frequency does the magnitude of transmission become unity? What is the phase angle at this frequency? Figure 2.23 Circuit for Example 2.4. Solution To obtain the transfer function of the circuit in Fig. 2.23, we substitute in Eq. (2.24), Z1 = R1 and Z2 = R2 (1/sC2). Since Z2 is the parallel connection of two components, it is more convenient to work in terms of Y2; that is, we use the following alternative form of the transfer function: Vo(s) = − 1 Vi(s) Z1(s)Y2(s) 2.5 Integrators and Differentiators 89 and substitute Z1 = R1 and Y2(s) = (1/R2) + sC2 to obtain Vo(s) Vi(s) = − R1 R2 1 + sC2R1 This transfer function is of first order, has a finite dc gain (at s = 0, Vo/Vi = −R2/R1), and has zero gain at infinite frequency. Thus it is the transfer function of a low-pass STC network and can be expressed in the standard form of Table 1.2 as follows: Vo(s) = −R2/R1 Vi(s) 1 + sC2R2 from which we find the dc gain K to be K = − R2 R1 and the 3-dB frequency ω0 as ω0 = 1 C2 R2 We could have found all this from the circuit in Fig. 2.23 by inspection. Specifically, note that the capacitor behaves as an open circuit at dc; thus at dc the gain is simply (−R2/R1). Furthermore, because there is a virtual ground at the inverting input terminal, the resistance seen by the capacitor is R2, and thus the time constant of the STC network is C2R2. Now to obtain a dc gain of 40 dB, that is, 100 V/V, we select R2/R1 = 100. For an input resistance of 1 k , we select R1 = 1 k , and thus R2 = 100 k . Finally, for a 3-dB frequency f0 = 1 kHz, we select C2 from 2π × 1 × 103 = C2 × 1 100 × 103 which yields C2 = 1.59 nF. The circuit has gain and phase Bode plots of the standard form in Fig. 1.23. As the gain falls off at the rate of –20 dB/decade, it will reach 0 dB in two decades, that is, at f = 100f0 = 100 kHz. As Fig. 1.23(b) indicates, at such a frequency, which is much greater than f0, the phase is approximately −90°. To this, however, we must add the 180° arising from the inverting nature of the amplifier (i.e., the negative sign in the transfer function expression). Thus at 100 kHz, the total phase shift will be −270° or, equivalently, +90°. 2.5.2 The Inverting Integrator By placing a capacitor in the feedback path (i.e., in place of Z2 in Fig. 2.22) and a resistor at the input (in place of Z1), we obtain the circuit of Fig. 2.24(a). We shall now show that this circuit realizes the mathematical operation of integration. Let the input be a time-varying function vI(t). The virtual ground at the inverting op-amp input causes vI(t) to appear in effect 90 Chapter 2 Operational Amplifiers across R, and thus the current i1(t) will be vI(t)/R. This current flows through the capacitor C, causing charge to accumulate on C. If we assume that the circuit begins operation at time t = 0, then at an arbitrary time t the current i1(t) will have deposited on C a charge equal to t 0 i1(t)dt. Thus the capacitor voltage v C (t ) will change by 1 C t 0 i1 (t)dt. If the initial voltage on C (at t = 0) is denoted VC, then 1t vC(t) = VC + C i1(t)dt 0 Now the output voltage vO(t) = −vC(t); thus, v O (t) = − 1 CR t vI (t)dt − VC 0 (2.25) Thus the circuit provides an output voltage that is proportional to the time integral of the input, with VC being the initial condition of integration and CR the integrator time constant. Note that, as expected, there is a negative sign attached to the output voltage, and thus this integrator circuit is said to be an inverting integrator. It is also known as a Miller integrator after an early worker in this field. The operation of the integrator circuit can be described alternatively in the frequency domain by substituting Z1(s) = R and Z2(s) = 1/sC in Eq. (2.24) to obtain the transfer function Vo(s) = − 1 Vi(s) sCR For physical frequencies, s = jω and (2.26) Vo( jω) = − 1 Vi( jω) jωCR Thus the integrator transfer function has magnitude (2.27) and phase Vo = 1 Vi ωCR (2.28) φ = +90° (2.29) The Bode plot for the integrator magnitude response can be obtained by noting from Eq. (2.28) that as ω doubles (increases by an octave) the magnitude is halved (decreased by 6 dB). Thus the Bode plot is a straight line of slope –6 dB/octave (or, equivalently, –20 dB/ decade). This line (shown in Fig. 2.24b) intercepts the 0-dB line at the frequency that makes |Vo/Vi| = 1, which from Eq. (2.28) is ωint = 1 CR (2.30) The frequency ωint is known as the integrator frequency and is simply the inverse of the integrator time constant. Comparison of the frequency response of the integrator to that of an STC low-pass network indicates that the integrator behaves as a low-pass filter with a corner frequency of zero. Observe also that at ω = 0, the magnitude of the integrator transfer function is infinite. This 2.5 Integrators and Differentiators 91 i1 ϩ vI (t) Ϫ ϩ vC Ϫ i1 R0 C Ϫ 0V ϩ Ύ vO(t) ϭ Ϫ1 CR t 0 vI (t) dtϪVC ϩ vO(t) Ϫ Vo ϭ Ϫ 1 Vi sCR (a) (b) Figure 2.24 (a) The Miller or inverting integrator. (b) Frequency response of the integrator. indicates that at dc the op amp is operating with an open loop. This should also be obvious from the integrator circuit itself. Reference to Fig. 2.24(a) shows that the feedback element is a capacitor, and thus at dc, where the capacitor behaves as an open circuit, there is no negative feedback! This is a very significant observation and one that indicates a source of problems with the integrator circuit: Any tiny dc component in the input signal will theoretically produce an infinite output. Of course, no infinite output voltage results in practice; rather, the output of the amplifier saturates at a voltage close to the op-amp positive or negative power supply (L+ or L−), depending on the polarity of the input dc signal. The dc problem of the integrator circuit can be alleviated by connecting a resistor RF across the integrator capacitor C, as shown in Fig. 2.25, and thus the gain at dc will be –RF/R rather than infinite. Such a resistor provides a dc feedback path. Unfortunately, however, the integration is no longer ideal, and the lower the value of RF, the less ideal the integrator circuit becomes. This is because RF causes the frequency of the integrator pole to move from its ideal location at ω = 0 to one determined by the corner frequency of the STC network (RF, C). Specifically, the integrator transfer function becomes Vo(s) = − RF/R Vi(s) 1 + sCRF 92 Chapter 2 Operational Amplifiers R ϩ vI (t) Ϫ RF C Ϫ ϩ ϩ vO (t) Ϫ Figure 2.25 The Miller integrator with a large resistance RF connected in parallel with C in order to provide negative feedback and hence finite gain at dc. as opposed to the ideal function of −1/sCR. The lower the value we select for RF, the higher the corner frequency (1/CRF) will be and the more nonideal the integrator becomes. Thus selecting a value for RF presents the designer with a trade-off between dc performance and signal performance. The effect of RF on integrator performance is investigated further in Example 2.5. Example 2.5 Find the output produced by a Miller integrator in response to an input pulse of 1-V height and 1-ms width [Fig. 2.26(a)]. Let R = 10 k and C = 10 nF. If the integrator capacitor is shunted by a 1-M resistor, how will the response be modified? The op amp is specified to saturate at ±13 V. Solution In response to a 1-V, 1-ms input pulse, the integrator output will be 1t vO(t) = − CR 1 dt, 0 0 ≤ t ≤ 1 ms where we have assumed that the initial voltage on the integrator capacitor is 0. For C = 10 nF and R = 10 k , CR = 0.1 ms, and vO(t) = −10t, 0 ≤ t ≤ 1 ms which is the linear ramp shown in Fig. 2.26(b). It reaches a magnitude of −10 V at t = 1 ms and remains constant thereafter. That the output is a linear ramp should also be obvious from the fact that the 1-V input pulse produces a constant current through the capacitor of 1 V/10 k = 0.1 mA. This constant current I = 0.1 mA supplies the capacitor with a charge It, and thus the capacitor voltage changes linearly as (It/C), resulting in vO = −(I/C)t. It is worth remembering that charging a capacitor with a constant current produces a linear voltage across it. vI (t) 1V 0 0 vO(t) 0 1 ms (a) 1 ms 2.5 Integrators and Differentiators 93 t t Ϫ10 V (b) vO(t) 0 1 ms t to 0 V Ϫ9.5 V Exponentials with time constant of 10 ms to Ϫ100 V (c) Figure 2.26 Waveforms for Example 2.5: (a) Input pulse. (b) Output linear ramp of ideal integrator with time constant of 0.1 ms. (c) Output exponential ramp with resistor RF connected across integrator capacitor. Next consider the situation with resistor RF = 1 M connected across C. As before, the 1-V pulse will provide a constant current I = 0.1 mA. Now, however, this current is supplied to an STC network composed of RF in parallel with C. Thus, the output will be an exponential heading toward −100 V with 94 Chapter 2 Operational Amplifiers Example 2.5 continued a time constant of CRF = 10 × 10−9 × 1 × 106 = 10 ms, vO(t) = −100(1 − e−t/10), 0 ≤ t ≤ 1 ms Of course, the exponential will be interrupted at the end of the pulse, that is, at t = 1 ms, and the output will reach the value vO(1 ms) = −100(1 − e−1/10) = −9.5 V The output waveform is shown in Fig. 2.26(c), from which we see that including RF causes the ramp to be slightly rounded such that the output reaches only −9.5 V, 0.5 V short of the ideal value of −10 V. Furthermore, for t > 1 ms, the capacitor discharges through RF with the relatively long time constant of 10 ms. Finally, we note that op-amp saturation, specified to occur at ±13 V, has no effect on the operation of this circuit. The preceding example hints at an important application of integrators, namely, their use in providing triangular waveforms in response to square-wave inputs. This application is explored in Exercise 2.18. Integrators have many other applications, including their use in the design of filters (Chapter 17). 2.5.3 The Op-Amp Differentiator Interchanging the location of the capacitor and the resistor of the integrator circuit results in the circuit in Fig. 2.27(a), which performs the mathematical function of differentiation. To see how this comes about, let the input be the time-varying function vI(t), and note that the virtual ground at the inverting input terminal of the op amp causes vI(t) to appear in effect across the capacitor C. Thus the current through C will be C(dvI/dt), and this current flows through the feedback resistor R providing at the op-amp output a voltage vO(t), v O (t) = −CR d vI (t) dt (2.31) The frequency-domain transfer function of the differentiator circuit can be found by substituting in Eq. (2.24), Z1(s) = 1/sC and Z2(s) = R to obtain Vo(s) = −sCR Vi(s) (2.32) which for physical frequencies s = jω yields Vo( jω) = −jωCR Vi( jω) Thus the transfer function has magnitude (2.33) Vo = ωCR Vi (2.34) 2.5 Integrators and Differentiators 95 and phase φ = −90° (2.35) The Bode plot of the magnitude response can be found from Eq. (2.34) by noting that for an octave increase in ω, the magnitude doubles (increases by 6 dB). Thus the plot is simply a straight line of slope +6 dB/octave (or, equivalently, +20 dB/decade) intersecting the 0-dB line (where |Vo/Vi| = 1) at ω = 1/CR, where CR is the differentiator time constant [see Fig. 2.27(b)]. The frequency response of the differentiator can be thought of as the response of an STC high-pass filter with a corner frequency at infinity (refer to Fig. 1.24). Finally, we should note that the very nature of a differentiator circuit causes it to be a “noise magnifier.” This is due to the spike introduced at the output every time there is a sharp change in vI(t); such a change could be interference coupled electromagnetically (“picked up”) from adjacent signal sources. For this reason and because they suffer from stability problems (Chapter 11), differentiator circuits are generally avoided in practice. When the circuit of Fig. 2.27(a) is used, it is usually necessary to connect a small-valued resistor in series with the capacitor. This modification, unfortunately, turns the circuit into a nonideal differentiator. iC ϩ vI (t) Ϫ 0V iR 0 Ϫ ϩ Vo (dB) Vi ϩ vO (t) Ϫ i (t) ϭ C dvI(t) vO (t) ϭ dt ϪCR dvI (t) dt Vo ϭ ϪsCR Vi (a) 0 1 CR ϩ6 dB/octave ␻ (log scale) (b) Figure 2.27 (a) A differentiator. (b) Frequency response of a differentiator with a time constant CR. 96 Chapter 2 Operational Amplifiers EXERCISES 2.18 Consider a symmetrical square wave of 20-V peak-to-peak, 0 average, and 2-ms period applied to a Miller integrator. Find the value of the time constant CR such that the triangular waveform at the output has a 20-V peak-to-peak amplitude. Ans. 0.5 ms D2.19 Use an ideal op amp to design an inverting integrator with an input resistance of 10 k and an integration time constant of 10−3 s. What is the gain magnitude and phase angle of this circuit at 10 rad/s and at 1 rad/s? What is the frequency at which the gain magnitude is unity? Ans. R = 10 k , C = 0.1 μF; at ω = 10 rad/s: |Vo/Vi| = 100 V/V and φ = +90°; at ω = 1 rad/s: |Vo/Vi| = 1000 V/V and φ = +90°; 1000 rad/s D2.20 Design a differentiator to have a time constant of 10−2 s and an input capacitance of 0.01 μF. What is the gain magnitude and phase of this circuit at 10 rad/s, and at 103 rad/s? In order to limit the high-frequency gain of the differentiator circuit to 100, a resistor is added in series with the capacitor. Find the required resistor value. Ans. C = 0.01 μF; R = 1 M ; at ω = 10 rad/s: |Vo/Vi| = 0.1 V/V and φ = −90°; at ω = 1000 rad/s: |Vo/Vi| = 10 V/V and φ = −90°; 10 k 2.6 DC Imperfections Thus far we have considered the op amp to be ideal. The only exception has been a brief discussion of the effect of the op-amp finite gain A on the closed-loop gain of the inverting and noninverting configurations. Although in many applications the assumption of an ideal op amp is not a bad one, a circuit designer has to be thoroughly familiar with the characteristics of practical op amps and the effects of such characteristics on the performance of op-amp circuits. Only then will the designer be able to use the op amp intelligently, especially if the application at hand is not a straightforward one. The nonideal properties of op amps will, of course, limit the range of operation of the circuits analyzed in the previous examples. In this and the two sections that follow, we consider some of the important nonideal properties of the op amp.4 We do this by treating one nonideality at a time, beginning in this section with the dc problems to which op amps are susceptible. 2.6.1 Offset Voltage Because op amps are direct-coupled devices with large gains at dc, they are prone to dc problems. The first such problem is the dc offset voltage. To understand this problem consider 4We should note that real op amps have nonideal effects additional to those discussed in this chapter. These include finite (nonzero) common-mode gain or, equivalently, noninfinite CMRR, noninfinite input resistance, and nonzero output resistance. The effect of these, however, on the performance of most of the closed-loop circuits studied here is not very significant, and their study will be postponed to later chapters (in particular, Chapters 9, 10, and 13). 2.6 DC Imperfections 97 the following conceptual experiment: If the two input terminals of the op amp are tied together and connected to ground, it will be found that despite the fact that vId = 0, a finite dc voltage exists at the output. In fact, if the op amp has a high dc gain, the output will be at either the positive or negative saturation level. The op-amp output can be brought back to its ideal value of 0 V by connecting a dc voltage source of appropriate polarity and magnitude between the two input terminals of the op amp. This external source balances out the input offset voltage of the op amp. It follows that the input offset voltage (VOS) must be of equal magnitude and of opposite polarity to the voltage we applied externally. The input offset voltage arises as a result of the unavoidable mismatches present in the input differential stage inside the op amp. In later chapters (in particular Chapters 9 and 13) we shall study this topic in detail. Here, however, our concern is to investigate the effect of VOS on the operation of closed-loop op-amp circuits. Toward that end, we note that general-purpose op amps exhibit VOS in the range of 1 mV to 5 mV. Also, the value of VOS depends on temperature. The op-amp data sheets usually specify typical and maximum values for VOS at room temperature as well as the temperature coefficient of VOS (usually in μV/°C). They do not, however, specify the polarity of VOS because the component mismatches that give rise to VOS are obviously not known a priori; different units of the same op-amp type may exhibit either a positive or a negative VOS. To analyze the effect of VOS on the operation of op-amp circuits, we need a circuit model for the op amp with input offset voltage. Such a model is shown in Fig. 2.28. It consists of a dc source of value VOS placed in series with the positive input lead of an offset-free op amp. The justification for this model follows from the description above. Figure 2.28 Circuit model for an op amp with input offset voltage VOS. EXERCISE 2.21 Use the model of Fig. 2.28 to sketch the transfer characteristic vO versus vId(vO ≡ v3 and vId ≡ v2 − v1) of an op amp having an open-loop dc gain A0 = 104 V/V, output saturation levels of ±10 V, and VOS of +5 mV. Ans. See Fig. E2.21. Observe that true to its name, the input offset voltage causes an offset in the voltage-transfer characteristic; rather than passing through the origin it is now shifted to the left by VOS. 98 Chapter 2 Operational Amplifiers Figure E2.21 Transfer characteristic of an op amp with VOS = 5 mV. Analysis of op-amp circuits to determine the effect of the op-amp VOS on their performance is straightforward: The input voltage signal source is short-circuited and the op amp is replaced with the model of Fig. 2.28. (Eliminating the input signal, done to simplify matters, is based on the principle of superposition.) Following this procedure, we find that both the inverting and the noninverting amplifier configurations result in the same circuit, that shown in Fig. 2.29, from which the output dc voltage due to VOS is found to be VO = VOS 1 + R2 R1 (2.36) This output dc voltage can have a large magnitude. For instance, a noninverting amplifier with a closed-loop gain of 1000, when constructed from an op amp with a 5-mV input offset voltage, will have a dc output voltage of +5 V or −5 V (depending on the polarity of VOS) rather than the ideal value of 0 V. Now, when an input signal is applied to the amplifier, the corresponding signal output will be superimposed on the 5-V dc. Obviously then, the R2 R1 Ϫ Ϫϩ ϩ VOS Offset-free op amp VO ϭ VOS 1ϩ R2 R1 VO Figure 2.29 Evaluating the output dc off- set voltage due to VOS in a closed-loop amplifier. Vϩ To rest Ϫ of circuit ϩ Offset-nulling terminals VϪ 2.6 DC Imperfections 99 Figure 2.30 The output dc offset voltage of an op amp can be trimmed to zero by connecting a potentiometer to the two offset-nulling terminals. The wiper of the potentiometer is connected to the negative supply of the op amp. R2 Ϫ Ϫϩ ϩ VO ϭ VOS VOS Offset free (a) (b) Figure 2.31 (a) A capacitively coupled inverting amplifier. (b) The equivalent circuit for determining its dc output offset voltage VO. allowable signal swing at the output will be reduced. Even worse, if the signal to be amplified is dc, we would not know whether the output is due to VOS or to the signal! Some op amps are provided with two additional terminals to which a specified circuit can be connected to trim to zero the output dc voltage due to VOS. Figure 2.30 shows such an arrangement that is typically used with general-purpose op amps. A potentiometer is connected between the offset-nulling terminals with the wiper of the potentiometer connected to the op-amp negative supply. Moving the potentiometer wiper introduces an imbalance that counteracts the asymmetry present in the internal op-amp circuitry and that gives rise to VOS. We shall return to this point in the context of our study of the internal circuitry of op amps in Chapter 13. It should be noted, however, that even though the dc output offset can be trimmed to zero, the problem remains of the variation (or drift) of VOS with temperature. One way to overcome the dc offset problem is by capacitively coupling the amplifier. This, however, will be possible only in applications where the closed-loop amplifier is not required to amplify dc or very-low-frequency signals. Figure 2.31(a) shows a capacitively coupled amplifier. Because of its infinite impedance at dc, the coupling capacitor will cause the gain to be zero at dc. As a result, the equivalent circuit for determining the dc output voltage resulting from the op-amp input offset voltage VOS will be that shown in Fig. 2.31(b). Thus VOS sees in effect a unity-gain voltage follower, and the dc output voltage VO will be equal to VOS rather than VOS(1 + R2/R1), which is the case without the coupling capacitor. As far as input signals are concerned, the coupling capacitor C forms together with R1 an STC high-pass circuit with a corner frequency of ω0 = 1/CR1. Thus the gain of the capacitively 100 Chapter 2 Operational Amplifiers coupled amplifier will fall off at the low-frequency end [from a magnitude of (1 + R2/R1) at high frequencies] and will be 3 dB down at ω0. EXERCISES 2.22 Consider an inverting amplifier with a nominal gain of 1000 constructed from an op amp with an input offset voltage of 3 mV and with output saturation levels of ±10 V. (a) What is (approximately) the peak sine-wave input signal that can be applied without output clipping? (b) If the effect of VOS is nulled at room temperature (25°C), how large an input can one now apply if: (i) the circuit is to operate at a constant temperature? (ii) the circuit is to operate at a temperature in the range 0°C to 75°C and the temperature coefficient of VOS is 10 μV/°C? Ans. (a) 7 mV; (b) 10 mV, 9.5 mV 2.23 Consider the same amplifier as in Exercise 2.22—that is, an inverting amplifier with a nominal gain of 1000 constructed from an op amp with an input offset voltage of 3 mV and with output saturation levels of ±10 V—except here let the amplifier be capacitively coupled as in Fig. 2.31(a). (a) What is the dc offset voltage at the output, and what (approximately) is the peak sine-wave signal that can be applied at the input without output clipping? Is there a need for offset trimming? (b) If R1 = 1 k and R2 = 1 M , find the value of the coupling capacitor C1 that will ensure that the gain will be greater than 57 dB down to 100 Hz. Ans. (a) 3 mV, 10 mV, no need for offset trimming; (b) 1.6 μF 2.6.2 Input Bias and Offset Currents The second dc problem encountered in op amps is illustrated in Fig. 2.32. In order for the op amp to operate, its two input terminals have to be supplied with dc currents, termed the input bias currents.5 In Fig. 2.32 these two currents are represented by two current sources, IB1 and IB2, connected to the two input terminals. It should be emphasized that the input bias currents are independent of the fact that a real op amp has finite (though large) input resistance (not shown in Fig. 2.32). The op-amp manufacturer usually specifies the average value of IB1 and IB2 as well as their expected difference. The average value IB is called the input bias current, IB = IB1 + IB2 2 and the difference is called the input offset current and is given by IOS = |IB1 − IB2| Typical values for general-purpose op amps that use bipolar transistors are IB = 100 nA and IOS = 10 nA. 5This is the case for op amps constructed using bipolar junction transistors (BJTs). Those using MOSFETs in the first (input) stage do not draw an appreciable input bias current; nevertheless, the input terminals should have continuous dc paths to ground. More on this in later chapters. 2.6 DC Imperfections 101 Figure 2.32 The op-amp input bias currents represented by two current sources IB1 and IB2. Figure 2.33 Analysis of the closed-loop amplifier, taking into account the input bias currents. We now wish to find the dc output voltage of the closed-loop amplifier due to the input bias currents. To do this we ground the signal source and obtain the circuit shown in Fig. 2.33 for both the inverting and noninverting configurations. As shown in Fig. 2.33, the output dc voltage is given by VO = IB1R2 IBR2 (2.37) 102 Chapter 2 Operational Amplifiers Figure 2.34 Reducing the effect of the input bias currents by introducing a resistor R3. This obviously places an upper limit on the value of R2. Fortunately, however, a technique exists for reducing the value of the output dc voltage due to the input bias currents. The method consists of introducing a resistance R3 in series with the noninverting input lead, as shown in Fig. 2.34. From a signal point of view, R3 has a negligible effect (ideally no effect). The appropriate value for R3 can be determined by analyzing the circuit in Fig. 2.34, where analysis details are shown, and the output voltage is given by VO = −IB2R3 + R2(IB1 − IB2R3/R1) Consider first the case IB1 = IB2 = IB, which results in VO = IB[R2 − R3(1 + R2/R1)] (2.38) Thus we can reduce VO to zero by selecting R3 such that R3 = R2 1 + R2/R1 = R1R2 R1 + R2 (2.39) That is, R3 should be made equal to the parallel equivalent of R1 and R2. Having selected R3 as above, let us evaluate the effect of a finite offset current IOS. Let IB1 = IB + IOS/2 and IB2 = IB − IOS/2, and substitute in Eq. (2.38). The result is VO = IOSR2 (2.40) which is usually about an order of magnitude smaller than the value obtained without R3 (Eq. 2.37). We conclude that to minimize the effect of the input bias currents, one should place in the positive lead a resistance equal to the equivalant dc resistance seen by the inverting terminal. We emphasize the word dc in the last statement; note that if the amplifier is ac-coupled, we should select R3 = R2, as shown in Fig. 2.35. 2.6 DC Imperfections 103 Figure 2.35 In an ac-coupled amplifier the dc resis- tance seen by the inverting terminal is R2; hence R3 is chosen equal to R2. ϭ R2 Figure 2.36 Illustrating the need for a continuous dc path for each of the op-amp input terminals. Specifically, note that the amplifier will not work without resistor R3. While we are on the subject of ac-coupled amplifiers, we should note that one must always provide a continuous dc path between each of the input terminals of the op amp and ground. This is the case no matter how small IB is. For this reason the ac-coupled noninverting amplifier of Fig. 2.36 will not work without the resistance R3 to ground. Unfortunately, including R3 lowers considerably the input resistance of the closed-loop amplifier. EXERCISE 2.24 Consider an inverting amplifier circuit designed using an op amp and two resistors, R1 = 10 k and R2 = 1 M . If the op amp is specified to have an input bias current of 100 nA and an input offset current of 10 nA, find the output dc offset voltage resulting and the value of a resistor R3 to be placed in series with the positive input lead in order to minimize the output offset voltage. What is the new value of VO? Ans. 0.1 V; 9.9 k ( 10 k ); 0.01 V 2.6.3 Effect of VOS and IOS on the Operation of the Inverting Integrator Our discussion of the inverting integrator circuit in Section 2.5.2 mentioned the susceptibility of this circuit to saturation in the presence of small dc voltages or currents. It behooves us therefore to consider the effect of the op-amp dc offsets on its operation. As will be seen, these effects can be quite dramatic. 104 Chapter 2 Operational Amplifiers t Figure 2.37 Determining the effect of the op-amp input offset voltage VOS on the Miller integrator circuit. Note that since the output rises with time, the op amp eventually saturates. To see the effect of the input dc offset voltage VOS, consider the integrator circuit in Fig. 2.37, where for simplicity we have short-circuited the input signal source. Analysis of the circuit is straightforward and is shown in Fig. 2.37. Assuming for simplicity that at time t = 0 the voltage across the capacitor is zero, the output voltage as a function of time is given by vO = VOS + VOS CR t (2.41) Thus vO increases linearly with time until the op amp saturates—clearly an unacceptable situation! As should be expected, the dc input offset current IOS produces a similar problem. Figure 2.38 illustrates the situation. Observe that we have added a resistance R in the op-amp positive-input lead in order to keep the input bias current IB from flowing through C. Nevertheless, the offset current IOS will flow through C and cause vO to ramp linearly with time until the op amp saturates. As mentioned in Section 2.5.2 the dc problem of the integrator circuit can be alleviated by connecting a resistor RF across the integrator capacitor C, as shown in Fig. 2.25. Such a resistor provides a dc path through which the dc currents (VOS/R) and IOS can flow (assuming a resistance equal to R RF is connected in the positive op-amp lead), with the result that vO will now have a dc component [VOS(1 + RF/R) + IOSRF] instead of rising linearly. To keep the dc offset at the output small, one would select a low value for RF. Unfortunately, however, the lower the value of RF, the less ideal the integrator circuit becomes. C (IB1 IB2) IOS IB2R R IB2 R IB1 vO R IB2 vO IB2R IOS C t IB2R Figure 2.38 Effect of the op-amp input bias and offset currents on the performance of the Miller integrator circuit. 2.7 Effect of Finite Open-Loop Gain and Bandwidth on Circuit Performance 105 EXERCISE 2.25 Consider a Miller integrator with a time constant of 1 ms and an input resistance of 10 k . Let the op amp have VOS = 2 mV and output saturation voltages of ±12 V. (a) Assuming that when the power supply is turned on the capacitor voltage is zero, how long does it take for the amplifier to saturate? (b) Select the largest possible value for a feedback resistor RF so that at least ±10 V of output signal swing remains available. What is the corner frequency of the resulting STC network? Ans. (a) 6 s; (b) 10 M , 0.16 Hz 2.7 Effect of Finite Open-Loop Gain and Bandwidth on Circuit Performance 2.7.1 Frequency Dependence of the Open-Loop Gain The differential open-loop gain A of an op amp is not infinite; rather, it is finite and decreases with frequency. Figure 2.39 shows a plot for |A|, with the numbers typical of some commercially available general-purpose op amps (such as the popular 741-type op amp, available from many semiconductor manufacturers; its internal circuit is studied in Chapter 13). Figure 2.39 Open-loop gain of a typical general-purpose, internally compensated op amp. 106 Chapter 2 Operational Amplifiers Note that although the gain is quite high at dc and low frequencies, it starts to fall off at a rather low frequency (10 Hz in our example). The uniform –20-dB/decade gain rolloff shown is typical of internally compensated op amps. These are units that have a network (usually a single capacitor) included within the same IC chip whose function is to cause the op-amp gain to have the single-time-constant (STC) low-pass response shown. This process of modifying the open-loop gain is termed frequency compensation, and its purpose is to ensure that op-amp circuits will be stable (as opposed to oscillatory). The subject of stability of op-amp circuits—or, more generally, of feedback amplifiers—will be studied in Chapter 11. By analogy to the response of low-pass STC circuits (see Section 1.6 and, for more detail, Appendix E), the gain A(s) of an internally compensated op amp may be expressed as A(s) = 1 A0 + s/ωb which for physical frequencies, s = jω, becomes (2.42) A( jω) = A0 1 + jω/ωb (2.43) where A0 denotes the dc gain and ωb is the 3-dB frequency (corner frequency or “break” frequency). For the example shown in Fig. 2.39, A0 = 105 and ωb = 2π × 10 rad/s. For frequencies ω ωb (about 10 times and higher) Eq. (2.43) may be approximated by A( jω) A0ωb jω (2.44) Thus, |A( jω)| = A0ωb ω (2.45) from which it can be seen that the gain |A| reaches unity (0 dB) at a frequency denoted by ωt and given by ωt = A0ωb (2.46) Substituting in Eq. (2.44) gives A(jω) ωt jω (2.47) The frequency ft = ωt/2π is usually specified on the data sheets of commercially available op amps and is known as the unity-gain bandwidth.6 Also note that for ω ωb the open-loop gain in Eq. (2.42) becomes A(s) ωt s (2.48) 6Since ft is the product of the dc gain A0 and the 3-dB bandwidth fb (where fb = ωb/2π ), it is also known as the gain–bandwidth product (GB). The reader is cautioned, however, that in some amplifiers (those that do not have an STC response), the unity-gain frequency and the gain–bandwidth product are not equal. 2.7 Effect of Finite Open-Loop Gain and Bandwidth on Circuit Performance 107 The gain magnitude can be obtained from Eq. (2.47) as |A(jω)| ωt = ft ωf (2.49) Thus if ft is known (106 Hz in our example), one can easily determine the magnitude of the op-amp gain at a given frequency f . Furthermore, observe that this relationship correlates with the Bode plot in Fig. 2.39. Specifically, for f fb, doubling f (an octave increase) results in halving the gain (a 6-dB reduction). Similarly, increasing f by a factor of 10 (a decade increase) results in reducing |A| by a factor of 10 (20 dB). As a matter of practical importance, we note that the production spread in the value of ft between op-amp units of the same type is usually much smaller than that observed for A0 and fb. For this reason ft is preferred as a specification parameter. Finally, it should be mentioned that an op amp having this uniform –6-dB/octave (or equivalently –20-dB/decade) gain rolloff is said to have a single-pole model. Also, since this single pole dominates the amplifier frequency response, it is called a dominant pole. For more on poles (and zeros), the reader may wish to consult Appendix F. EXERCISE 2.26 An internally compensated op amp is specified to have an open-loop dc gain of 106 dB and a unity-gain bandwidth of 3 MHz. Find fb and the open-loop gain (in dB) at fb, 300 Hz, 3 kHz, 12 kHz, and 60 kHz. Ans. 15 Hz; 103 dB; 80 dB; 60 dB; 48 dB; 34 dB 2.7.2 Frequency Response of Closed-Loop Amplifiers We next consider the effect of limited op-amp gain and bandwidth on the closed-loop transfer functions of the two basic configurations: the inverting circuit of Fig. 2.5 and the noninverting circuit of Fig. 2.12. The closed-loop gain of the inverting amplifier, assuming a finite op-amp open-loop gain A, was derived in Section 2.2 and given in Eq. (2.5), which we repeat here as Vo = −R2/R1 Vi 1 + (1 + R2/R1)/A (2.50) Substituting for A from Eq. (2.42) and using Eq. (2.46) gives For A0 Vo(s) = Vi(s) 1 + 1 A0 −R2/R1 1 + R2 R1 + s ωt/(1 + R2/R1) 1 + R2/R1, which is usually the case, Vo(s) Vi(s) −R2/R1 1 + ωt /(1 s + R2/R1) (2.51) (2.52) 108 Chapter 2 Operational Amplifiers which is of the same form as that for a low-pass STC network (see Table 1.2, page 36). Thus the inverting amplifier has an STC low-pass response with a dc gain of magnitude equal to R2/R1. The closed-loop gain rolls off at a uniform –20-dB/decade slope with a corner frequency (3-dB frequency) given by ω3dB = ωt 1 + R2/R1 (2.53) Similarly, analysis of the noninverting amplifier of Fig. 2.12, assuming a finite open-loop gain A, yields the closed-loop transfer function Vo = 1 + R2/R1 Vi 1 + (1 + R2/R1)/A Substituting for A from Eq. (2.42) and making the approximation A0 (2.54) 1 + R2/R1 results in Vo(s) Vi(s) 1 + R2/R1 1+ s ωt/(1 + R2/R1) (2.55) Thus the noninverting amplifier has an STC low-pass response with a dc gain of (1 + R2/R1) and a 3-dB frequency given also by Eq. (2.53). Example 2.6 Consider an op amp with ft = 1 MHz. Find the 3-dB frequency of closed-loop amplifiers with nominal gains of +1000, +100, +10, +1, −1, −10, −100, and −1000. Sketch the magnitude frequency response for the amplifiers with closed-loop gains of +10 and −10. Solution We use Eq. (2.53) to obtain the results given in the following table. Closed-Loop Gain +1000 +100 +10 +1 −1 −10 −100 −1000 R2/R1 999 99 9 0 1 10 100 1000 f3 dB = ft/(1 + R2/R1) 1 kHz 10 kHz 100 kHz 1 MHz 0.5 MHz 90.9 kHz 9.9 kHz 1 kHz Figure 2.40 shows the frequency response for the amplifier whose nominal dc gain is +10 (20 dB), and Fig. 2.41 shows the frequency response for the –10 (also 20 dB) case. An interesting observation follows 2.7 Effect of Finite Open-Loop Gain and Bandwidth on Circuit Performance 109 from the table above: The unity-gain inverting amplifier has a 3-dB frequency of ft/2 as compared to ft for the unity-gain noninverting amplifier (the unity-gain voltage follower). Figure 2.40 Frequency response of an amplifier with a nominal gain of +10 V/V. Figure 2.41 Frequency response of an amplifier with a nominal gain of −10 V/V. The table in Example 2.6 above clearly illustrates the trade-off between gain and bandwidth: For a given op amp, the lower the closed-loop gain required, the wider the bandwidth achieved. Indeed, the noninverting configuration exhibits a constant gain–bandwidth product equal to ft of the op amp. An interpretation of these results in terms of feedback theory will be given in Chapter 11. 110 Chapter 2 Operational Amplifiers EXERCISES 2.27 An internally compensated op amp has a dc open-loop gain of 106 V/V and an open-loop gain of 40 dB at 10 kHz. Estimate its 3-dB frequency, its unity-gain frequency, its gain–bandwidth product, and its expected gain at 1 kHz. Ans. 1 Hz; 1 MHz; 1 MHz; 60 dB 2.28 An op amp having a 106-dB gain at dc and a single-pole frequency response with ft = 2 MHz is used to design a noninverting amplifier with nominal dc gain of 100. Find the 3-dB frequency of the closed-loop gain. Ans. 20 kHz 2.8 Large-Signal Operation of Op Amps In this section, we study the limitations on the performance of op-amp circuits when large output signals are present. 2.8.1 Output Voltage Saturation Similar to all other amplifiers, op amps operate linearly over a limited range of output voltages. Specifically, the op-amp output saturates in the manner shown in Fig. 1.14 with L+ and L− within 1 V or so of the positive and negative power supplies, respectively. Thus, an op amp that is operating from ±15-V supplies will saturate when the output voltage reaches about +13 V in the positive direction and –13 V in the negative direction. For this particular op amp the rated output voltage is said to be ±13 V. To avoid clipping off the peaks of the output waveform, and the resulting waveform distortion, the input signal must be kept correspondingly small. 2.8.2 Output Current Limits Another limitation on the operation of op amps is that their output current is limited to a specified maximum. For instance, the popular 741 op amp is specified to have a maximum output current of ±20 mA. Thus, in designing closed-loop circuits utilizing the 741, the designer has to ensure that under no condition will the op amp be required to supply an output current, in either direction, exceeding 20 mA. This, of course, has to include both the current in the feedback circuit as well as the current supplied to a load resistor. If the circuit requires a larger current, the op-amp output voltage will saturate at the level corresponding to the maximum allowed output current. 2.8 Large-Signal Operation of Op Amps 111 Example 2.7 Consider the noninverting amplifier circuit shown in Fig. 2.42. As shown, the circuit is designed for a nominal gain (1 + R2/R1) = 10 V/V. It is fed with a low-frequency sine-wave signal of peak voltage Vp and is connected to a load resistor RL. The op amp is specified to have output saturation voltages of ±13 V and output current limits of ±20 mA. (a) For Vp = 1 V and RL = 1 k , specify the signal resulting at the output of the amplifier. (b) For Vp = 1.5 V and RL = 1 k , specify the signal resulting at the output of the amplifier. (c) For RL = 1 k , what is the maximum value of Vp for which an undistorted sine-wave output is obtained? (d) For Vp = 1 V, what is the lowest value of RL for which an undistorted sine-wave output is obtained? R2 ϭ 9 k⍀ vO 15 V 13 V 1 k⍀ Ϫ iO iF R1 ϩ vO 0 iL t Vp 0 t vI ϩ Ϫ RL Ϫ13 V Ϫ15 V (a) (b) Figure 2.42 (a) A noninverting amplifier with a nominal gain of 10 V/V designed using an op amp that saturates at ±13-V output voltage and has ±20-mA output current limits. (b) When the input sine wave has a peak of 1.5 V, the output is clipped off at ±13 V. Solution (a) For Vp = 1 V and RL = 1 k , the output will be a sine wave with peak value of 10 V. This is lower than output saturation levels of ±13 V, and thus the amplifier is not limited that way. Also, when the output is at its peak (10 V), the current in the load will be 10 V/1 k = 10 mA, and the current in the feedback network will be 10 V/(9 + 1) k = 1 mA, for a total op-amp output current of 11 mA, well under its limit of 20 mA. (b) Now if Vp is increased to 1.5 V, ideally the output would be a sine wave of 15-V peak. The op amp, however, will saturate at ±13 V, thus clipping the sine-wave output at these levels. Let’s next check on the op-amp output current: At 13-V output and RL = 1 k , iL = 13 mA and iF = 1.3 mA; thus iO = 14.3 mA, again under the 20-mA limit. Thus the output will be a sine wave with its peaks clipped off at ±13 V, as shown in Fig. 2.42(b). (c) For RL = 1 k , the maximum value of Vp for undistorted sine-wave output is 1.3 V. The output will be a 13-V peak sine wave, and the op-amp output current at the peaks will be 14.3 mA. 112 Chapter 2 Operational Amplifiers Example 2.7 continued (d) For Vp = 1 V and RL reduced, the lowest value possible for RL while the output is remaining an undistorted sine wave of 10-V peak can be found from which results in 10 V 10 V iOmax = 20 mA = RLmin + 9 k + 1 k RLmin = 526 2.8.3 Slew Rate Another phenomenon that can cause nonlinear distortion when large output signals are present is slew-rate limiting. The name refers to the fact that there is a specific maximum rate of change possible at the output of a real op amp. This maximum is known as the slew rate (SR) of the op amp and is defined as SR = dvO dt max (2.56) and is usually specified on the op-amp data sheet in units of V/μs. It follows that if the input signal applied to an op-amp circuit is such that it demands an output response that is faster than the specified value of SR, the op amp will not comply. Rather, its output will change at the maximum possible rate, which is equal to its SR. As an example, consider an op amp connected in the unity-gain voltage-follower configuration shown in Fig. 2.43(a), and let the input signal be the step voltage shown in Fig. 2.43(b). The output of the op amp will not be able to rise instantaneously to the ideal value V ; rather, the output will be the linear ramp of slope equal to SR, shown in Fig. 2.43(c). The amplifier is then said to be slewing, and its output is slew-rate limited. In order to understand the origin of the slew-rate phenomenon, we need to know about the internal circuit of the op amp, and we will study it in Chapter 13. For the time being, however, it is sufficient to know about the phenomenon and to note that it is distinct from the finite op-amp bandwidth that limits the frequency response of the closed-loop amplifiers, studied in the previous section. The limited bandwidth is a linear phenomenon and does not result in a change in the shape of an input sinusoid; that is, it does not lead to nonlinear distortion. The slew-rate limitation, on the other hand, can cause nonlinear distortion to an input sinusoidal signal when its frequency and amplitude are such that the corresponding ideal output would require vO to change at a rate greater than SR. This is the origin of another related op-amp specification, its full-power bandwidth, to be explained later. Before leaving the example in Fig. 2.43, however, we should point out that if the step input voltage V is sufficiently small, the output can be the exponentially rising ramp shown 2.8 Large-Signal Operation of Op Amps 113 v1 V 0 (b) vO Slope ϭ SR V 0 (c) vO t t Slope ϭ ␻tV Յ SR V 0 t (d) Figure 2.43 (a) Unity-gain follower. (b) Input step waveform. (c) Linearly rising output waveform obtained when the amplifier is slew-rate limited. (d) Exponentially rising output waveform obtained when V is sufficiently small so that the initial slope (ωtV ) is smaller than or equal to SR. in Fig. 2.43(d). Such an output would be expected from the follower if the only limitation on its dynamic performance were the finite op-amp bandwidth. Specifically, the transfer function of the follower can be found by substituting R1 = ∞ and R2 = 0 in Eq. (2.55) to obtain Vo = 1 Vi 1 + s/ωt (2.57) which is a low-pass STC response with a time constant 1/ωt. Its step response would therefore be (see Appendix E) vO(t) = V (1 − e−ωtt) (2.58) The initial slope of this exponentially rising function is (ωtV ). Thus, as long as V is sufficiently small so that ωtV ≤ SR, the output will be as in Fig. 2.43(d). 114 Chapter 2 Operational Amplifiers EXERCISE 2.29 An op amp that has a slew rate of 1 V/μs and a unity-gain bandwidth ft of 1 MHz is connected in the unity-gain follower configuration. Find the largest possible input voltage step for which the output waveform will still be given by the exponential ramp of Eq. (2.58). For this input voltage, what is the 10% to 90% rise time of the output waveform? If an input step 10 times as large is applied, find the 10% to 90% rise time of the output waveform. Ans. 0.16 V; 0.35 μs; 1.28 μs 2.8.4 Full-Power Bandwidth Op-amp slew-rate limiting can cause nonlinear distortion in sinusoidal waveforms. Consider once more the unity-gain follower with a sine-wave input given by vI = Vˆi sin ωt The rate of change of this waveform is given by dvI dt = ωVˆi cos ωt with a maximum value of ωVˆi. This maximum occurs at the zero crossings of the input sinusoid. Now if ωVˆi exceeds the slew rate of the op amp, the output waveform will be distorted in the manner shown in Fig. 2.44. Observe that the output cannot keep up with the large rate of change of the sinusoid at its zero crossings, and the op amp slews. The op-amp data sheets usually specify a frequency fM called the full-power bandwidth. It is the frequency at which an output sinusoid with amplitude equal to the rated output voltage of the op amp begins to show distortion due to slew-rate limiting. If we denote the rated output Figure 2.44 Effect of slew-rate limiting on output sinusoidal waveforms. Summary 115 voltage Vomax, then fM is related to SR as follows: ωM Vomax = SR Thus, SR fM = 2π Vomax (2.59) It should be obvious that output sinusoids of amplitudes smaller than Vomax will show slew-rate distortion at frequencies higher than ωM. In fact, at a frequency ω higher than ωM, the maximum amplitude of the undistorted output sinusoid is given by Vo = Vomax ωM ω (2.60) EXERCISE 2.30 An op amp has a rated output voltage of ±10 V and a slew rate of 1 V/μs. What is its full-power bandwidth? If an input sinusoid with frequency f = 5fM is applied to a unity-gain follower constructed using this op amp, what is the maximum possible amplitude that can be accommodated at the output without incurring SR distortion? Ans. 15.9 kHz; 2 V (peak) Summary The IC op amp is a versatile circuit building block. It is easy to apply, and the performance of op-amp circuits closely matches theoretical predictions. The op-amp terminals are the inverting input terminal (1), the noninverting input terminal (2), the output terminal (3), the positive-supply terminal (4) to be connected to the positive power supply (VCC), and the negative-supply terminal (5) to be connected to the negative supply (−VEE). The common terminal of the two supplies is the circuit ground. The ideal op amp responds only to the difference input signal, that is, (v2 − v1); it provides at the output, between terminal 3 and ground, a signal A(v2 − v1), where A, the open-loop gain, is very large (104 to 106) and ideally infinite; and it has an infinite input resistance and a zero output resistance. (See Table 2.1.) Negative feedback is applied to an op amp by connecting a passive component between its output terminal and its inverting (negative) input terminal. Negative feedback causes the voltage between the two input terminals to become very small and ideally zero. Correspondingly, a virtual short circuit is said to exist between the two input terminals. If the positive input terminal is connected to ground, a virtual ground appears on the negative input terminal. The two most important assumptions in the analysis of op-amp circuits, presuming negative feedback exists and the op amps are ideal, are as follows: the two input terminals of the op amp are at the same voltage, and zero current flows into the op-amp input terminals. With negative feedback applied and the loop closed, the closed-loop gain is almost entirely determined by external components: For the inverting configuration, Vo/Vi = −R2/R1; and for the noninverting configuration, Vo/Vi = 1 + R2/R1. The noninverting closed-loop configuration features a very high input resistance. A special case is the unity-gain 116 Chapter 2 Operational Amplifiers follower, frequently employed as a buffer amplifier to connect a high-resistance source to a low-resistance load. The difference amplifier of Fig. 2.16 is designed with R4/R3 = R2/R1, resulting in vO = (R2/R1)(vI2 − vI1). The instrumentation amplifier of Fig. 2.20(b) is a very popular circuit. It provides vO = (1 + R2/R1)(R4/R3) (vI2 − vI1). It is usually designed with R3 = R4, and R1 and R2 selected to provide the required gain. If an adjustable gain is needed, part of R1 can be made variable. The inverting Miller integrator of Fig. 2.24(a) is a popular circuit, frequently employed in analog signal-processing functions such as filters (Chapter 17) and oscillators (Chapter 18). The input offset voltage, VOS, is the magnitude of dc voltage that when applied between the op-amp input terminals, with appropriate polarity, reduces the dc offset voltage at the output to zero. The effect of VOS on performance can be evaluated by including in the analysis a dc source VOS in series with the op-amp positive input lead. For both the inverting and the noninverting configurations, VOS results in a dc offset voltage at the output of VOS(1 + R2/R1). Capacitively coupling an op amp reduces the dc offset voltage at the output considerably. The average of the two dc currents, IB1 and IB2, that flow in the input terminals of the op amp, is called the input bias current, IB. In a closed-loop amplifier, IB gives rise to a dc offset voltage at the output of magnitude IBR2. This voltage can be reduced to IOSR2 by connecting a resistance in series with the positive input terminal equal to the total dc resistance seen by the negative input terminal. IOS is the input offset current; that is, IOS = |IB1 − IB2|. Connecting a large resistance in parallel with the capacitor of an op-amp inverting integrator prevents op-amp saturation (due to the effect of VOS and IB). For most internally compensated op amps, the open-loop gain falls off with frequency at a rate of −20 dB/decade, reaching unity at a frequency ft (the unity-gain bandwidth). Frequency ft is also known as the gain–bandwidth product of the op amp: ft = A0 fb, where A0 is the dc gain, and fb is the 3-dB frequency of the open-loop gain. At any frequency f (f fb), the op-amp gain |A| ft/f . For both the inverting and the noninverting closedloop configurations, the 3-dB frequency is equal to ft/(1 + R2/R1). The maximum rate at which the op-amp output voltage can change is called the slew rate. The slew rate, SR, is usually specified in V/μs. Op-amp slewing can result in nonlinear distortion of output signal waveforms. The full-power bandwidth, fM, is the maximum frequency at which an output sinusoid with an amplitude equal to the op-amp rated output voltage (Vomax) can be produced without distortion: fM = SR/2π Vomax. PROBLEMS Computer Simulation Problems Problems identified by the Multisim/PSpice icon are intended to demonstrate the value of using SPICE simulation to verify hand analysis and design, and to investigate important issues such as allowable signal swing and amplifier nonlinear distortion. Instructions to assist in setting up PSPice and Multisim simulations for all the indicated problems can be found in the corresponding files on the website. Note that if a particular parameter value is not specified in the problem statement, you are to make a reasonable assumption. Section 2.1: The Ideal Op Amp 2.1 What is the minimum number of pins required for a so-called dual-op-amp IC package, one containing two op amps? What is the number of pins required for a so-called quad-op-amp package, one containing four op-amps? = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem Problems 117 CHAPTER 2 PROBLEMS 2.2 The circuit of Fig. P2.2 uses an op amp that is ideal except for having a finite gain A. Measurements indicate vO = 4.0 V when vI = 1.0 V. What is the op-amp gain A? For equal transconductances Gm and a transresistance Rm, find an expression for the open-loop gain A. For Gm = 40 mA/V and Rm = 1 × 106 , what value of A results? 2.6 The two wires leading from the output terminals of a transducer pick up an interference signal that is a 60-Hz, 2-V sinusoid. The output signal of the transducer is sinusoidal of 5-mV amplitude and 1000-Hz frequency. Give expressions for vcm, vd, and the total signal between each wire and the system ground. Figure P2.2 2.7 Nonideal (i.e., real) operational amplifiers respond to both the differential and common-mode components of their input signals (refer to Fig. 2.4 for signal representation). Thus the output voltage of the op amp can be expressed as 2.3 Measurement of a circuit incorporating what is thought to be an ideal op amp shows the voltage at the op-amp output to be −2.000 V and that at the negative input to be −1.000 V. For the amplifier to be ideal, what would you expect the voltage at the positive input to be? If the measured voltage at the positive input is −1.005 V, what is likely to be the actual gain of the amplifier? 2.4 A set of experiments is run on an op amp that is ideal except for having a finite gain A. The results are tabulated below. Are the results consistent? If not, are they reasonable, in view of the possibility of experimental error? What do they show the gain to be? Using this value, predict values of the measurements that were accidentally omitted (the blank entries). vO = Ad vId + AcmvIcm where Ad is the differential gain (referred to simply as A in the text) and Acm is the common-mode gain (assumed to be zero in the text). The op amp’s effectiveness in rejecting common-mode signals is measured by its CMRR, defined as CMRR = 20 log Ad Acm Consider an op amp whose internal structure is of the type shown in Fig. E2.3 except for a mismatch Gm between the transconductances of the two channels; that is, Gm1 = Gm − 1 2 Gm Gm2 = Gm + 1 2 Gm Experiment # v1 v2 vO 1 0.00 0.00 0.00 2 1.00 1.00 0.00 3 1.00 1.00 4 1.00 1.10 10.1 5 2.01 2.00 −0.99 6 1.99 2.00 1.00 7 5.10 −5.10 2.5 Refer to Exercise 2.3. This problem explores an alternative internal structure for the op amp. In particular, we wish to model the internal structure of a particular op amp using two transconductance amplifiers and one transresistance amplifier. Suggest an appropriate topology. Find expressions for Ad, Acm, and CMRR. What is the maximum permitted percentage mismatch between the two Gm values if a minimum CMRR of 60 dB is required? Section 2.2: The Inverting Configuration 2.8 Assuming ideal op amps, find the voltage gain vo/vi and input resistance Rin of each of the circuits in Fig. P2.8. 2.9 A particular inverting circuit uses an ideal op amp and two 10-k resistors. What closed-loop gain would you expect? If a dc voltage of +1.00 V is applied at the input, what outputs result? If the 10-k resistors are said to be “1% resistors,” having values somewhere in the range (1 ± 0.01) times the nominal value, what range of outputs would you expect to actually measure for an input of precisely 1.00 V? = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem 118 Chapter 2 Operational Amplifiers CHAPTER 2 PROBLEMS 20 k⍀ Ϫ ϩ (a) 20 k⍀ Ϫ ϩ (b) 20 k⍀ 20 k⍀ Ϫ 20 k⍀ ϩ (c) Figure P2.8 20 k⍀ Ϫ ϩ 20 k⍀ (d) 2.10 You are provided with an ideal op amp and three 10-k resistors. Using series and parallel resistor combinations, how many different inverting-amplifier circuit topologies are possible? What is the largest (noninfinite) available voltage gain magnitude? What is the smallest (nonzero) available gain magnitude? What are the input resistances in these two cases? 2.11 For ideal op amps operating with the following feedback networks in the inverting configuration, what closed-loop gain results? (a) R1 = 10 k , R2 = 10 k (b) R1 = 10 k , R2 = 100 k (c) R1 = 10 k , R2 = 1 k (d) R1 = 100 k , R2 = 10 M (e) R1 = 100 k , R2 = 1 M D 2.12 Given an ideal op amp, what are the values of the resistors R1 and R2 to be used to design amplifiers with the closed-loop gains listed below? In your designs, use at least one 10-k resistor and another equal or larger resistor. (a) −1 V/V (b) −2 V/V (c) −5 V/V (d) −100 V/V D 2.13 Design an inverting op-amp circuit for which the gain is −10 V/V and the total resistance used is 110 k . D 2.14 Using the circuit of Fig. 2.5 and assuming an ideal op amp, design an inverting amplifier with a gain of 46 dB having the largest possible input resistance under the constraint of having to use resistors no larger than 1 M . What is the input resistance of your design? 2.15 An ideal op amp is connected as shown in Fig. 2.5 with R1 = 10 k and R2 = 100 k . A symmetrical square-wave signal with levels of 0 V and −1 V is applied at the input. Sketch and clearly label the waveform of the resulting output voltage. What is its average value? What is its highest value? What is its lowest value? 2.16 For the circuit in Fig. P2.16, assuming an ideal op amp, find the currents through all branches and the voltages at all nodes. Since the current supplied by the op amp is greater than the current drawn from the input signal source, where does the additional current come from? = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 2 PROBLEMS Ϫ1 V ϩ Ϫ 1 k⍀ 10 k⍀ Ϫ ϩ 2 k⍀ Problems 119 voltage ranges from −10 V to +10 V, what is the maximum voltage by which the “virtual ground node” departs from its ideal value? 2.22 The circuit in Fig. P2.22 is frequently used to provide an output voltage vo proportional to an input signal current ii. Figure P2.16 2.17 An inverting op-amp circuit is fabricated with the resistors R1 and R2 having x% tolerance (i.e., the value of each resistance can deviate from the nominal value by as much as ±x%). What is the tolerance on the realized closed-loop gain? Assume the op amp to be ideal. If the nominal closed-loop gain is −100 V/V and x = 1, what is the range of gain values expected from such a circuit? 2.18 An ideal op amp with 5-k and 15-k resistors is used to create a +5-V supply from a −15-V reference. Sketch the circuit. What are the voltages at the ends of the 5-k resistor? If these resistors are so-called 1% resistors, whose actual values are the range bounded by the nominal value ±1%, what are the limits of the output voltage produced? If the −15-V supply can also vary by ±1%, what is the range of the output voltages that might be found? 2.19 An inverting op-amp circuit for which the required gain is −50 V/V uses an op amp whose open-loop gain is only 500 V/V. If the larger resistor used is 100 k , to what must the smaller be adjusted? With what resistor must a 2-k resistor connected to the input be shunted to achieve this goal? (Note that a resistor Ra is said to be shunted by resistor Rb when Rb is placed in parallel with Ra.) D 2.20 (a) Design an inverting amplifier with a closed-loop gain of −200 V/V and an input resistance of 1 k . (b) If the op amp is known to have an open-loop gain of 5000 V/V, what do you expect the closed-loop gain of your circuit to be (assuming the resistors have precise values)? (c) Give the value of a resistor you can place in parallel (shunt) with R1 to restore the closed-loop gain to its nominal value. Use the closest standard 1% resistor value (see Appendix J). ϩ vi vo Ϫ Figure P2.22 Derive expressions for the transresistance Rm ≡ vo/ii and the input resistance Ri ≡ vi/ii for the following cases: (a) A is infinite. (b) A is finite. 2.23 Show that for the inverting amplifier if the op-amp gain is A, the input resistance is given by Rin = R1 + R2 A+ 1 2.24 For an inverting amplifier with nominal closed-loop gain R2/R1, find the minimum value that the op-amp open-loop gain A must have (in terms of R2/R1) so that the gain error (due to the finite A) is limited to 0.1%, 1%, and 10%. In each case find the value of a resistor RIa such that when it is placed in shunt with R1, the gain is restored to its nominal value. *2.25 Figure P2.25 shows an op amp that is ideal except for having a finite open-loop gain and is used to realize an inverting amplifier whose gain has a nominal magnitude G = R2/R1. To compensate for the gain reduction due to Rc R2 Vi Ϫ R1 Vo ϩ 2.21 An op amp with an open-loop gain of 5000 V/V is used in the inverting configuration. If in this application the output Figure P2.25 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem 120 Chapter 2 Operational Amplifiers CHAPTER 2 PROBLEMS the finite A, a resistor Rc is shunted across R1. Show that perfect compensation is achieved when Rc is selected according to D 2.29 An inverting op-amp circuit using an ideal op amp must be designed to have a gain of −500 V/V using resistors no larger than 100 k . Rc = A − G R1 1 + G D *2.26 (a) Use Eq. (2.5) to obtain the amplifier open-loop gain A required to realize a specified closed-loop gain (Gnominal = −R2/R1) within a specified gain error e, e ≡ G − Gnominal Gnominal (b) Design an inverting amplifer for a nominal closed-loop gain of −100, an input resistance of 1 k , and a gain error of ≤10%. Specify R1, R2, and the minimum A required. *2.27 (a) Use Eq. (2.5) to show that a reduction A in the op-amp gain A gives rise to a reduction |G| in the magnitude of the closed-loop gain G with |G| and A related by (a) For the simple two-resistor circuit, what input resistance would result? (b) If the circuit in Fig. 2.8 is used with three resistors of maximum value, what input resistance results? What is the value of the smallest resistor needed? 2.30 The inverting circuit with the T network in the feedback is redrawn in Fig. P2.30 in a way that emphasizes the observation that R2 and R3 in effect are in parallel (because the ideal op amp forces a virtual ground at the inverting input terminal). Use this observation to derive an expression for the gain (vO/vI ) by first finding (vX /vI ) and (vO/vX ). For the latter use the voltage-divider rule applied to R4 and (R2 R3). R2 vX R4 |G|/|G| 1 + R2/R1 R3 A/A A Assume that 1 + R2 R1 A A and 1. A iI vI (b) If in a closed-loop amplifier with a nominal gain (i.e., R1 Ϫ 0V vO R2/R1) of 100, A decreases by 10%, what is the minimum ϩ nominal A required to limit the percentage change in | G | to 0.1%? 2.28 Consider the circuit in Fig. 2.8 with R1 = R2 = R4 = 1 M , and assume the op amp to be ideal. Find values for R3 to obtain the following gains: (a) −100 V/V (b) −10 V/V (c) −2 V/V Figure P2.30 *2.31 The circuit in Fig. P2.31 can be considered to be an extension of the circuit in Fig. 2.8. (a) Find the resistances looking into node 1, R1; node 2, R2; node 3, R3; and node 4, R4. R1 R/2 2 R/2 3 R/2 4 I R1 R I1 2 0V 1 Figure P2.31 Ideal R2 R I2 R3 R I3 R4 I4 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem Problems 121 CHAPTER 2 PROBLEMS (b) Find the currents I1, I2, I3, and I4, in terms of the input current I. 10 kV iL RL (c) Find the voltages at nodes 1, 2, 3, and 4, that is, V1, V2, V3, and V4 in terms of (IR). R iI 2.32 The circuit in Fig. P2.32 utilizes an ideal op amp. 2 (a) Find I1, I2, I3, IL, and Vx. (b) If VO is not to be lower than −13 V, find the maximum 1 vO allowed value for RL. (c) If RL is varied in the range 100 to 1 k , what is the corresponding change in IL and in VO? Figure P2.34 I2 10 kV VX RL IL I1 10 kV 1V I3 100 V 2 1 D 2.35 Design the circuit shown in Fig. P2.35 to have an input resistance of 100 k and a gain that can be varied from −1 V/V to −100 V/V using the 100-k potentiometer R4. What voltage gain results when the potentiometer is set exactly at its middle value? VO R3 Figure P2.32 D 2.33 Use the circuit in Fig. P2.32 as an inspiration to design a circuit that supplies a constant current IL of 3.1 mA to a variable resistance RL. Assume the availability of a 1.5-V battery and design so that the current drawn from the battery is 0.1 mA. For the smallest resistance in the circuit, use 500 . If the op amp saturates at ±10 V, what is the maximum value that RL can have while the current source supplying it operates properly? D 2.34 Assuming the op amp to be ideal, it is required to design the circuit shown in Fig. P2.34 to implement a current amplifier with gain iL/iI = 11 A/A. (a) Find the required value for R. (b) What are the input and the output resistance of this current amplifier? (c) If RL = 1 k and the op amp operates in an ideal manner as long as vO is in the range ±12 V, what range of iI is possible? (d) If the amplifier is fed with a current source having a current of 0.2 mA and a source resistance of 10 k , find iL. R1 vI R2 2 1 R4 vO Figure P2.35 2.36 A weighted summer circuit using an ideal op amp has three inputs using 10-k resistors and a feedback resistor of 50 k . A signal v1 is connected to two of the inputs while a signal v2 is connected to the third. Express vO in terms of v1 and v2. If v1 = 1 V and v2 = −1 V, what is vO? D 2.37 Design an op-amp circuit to provide an output vO = −[2v1 + (v2/2)]. Choose relatively low values of resistors but ones for which the input current (from each input signal source) does not exceed 50 μA for 1-V input signals. D 2.38 Use the scheme illustrated in Fig. 2.10 to design an op-amp circuit with inputs v1, v2, and v3, whose output is = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem 122 Chapter 2 Operational Amplifiers CHAPTER 2 PROBLEMS vO = −(2v1+ 4v2+ 8v3) using small resistors but no smaller than 1 k . D 2.39 An ideal op amp is connected in the weighted summer configuration of Fig. 2.10. The feedback resistor Rf = 100 k , and six 100-k resistors are connected to the inverting input terminal of the op amp. Show, by sketching the various circuit configurations, how this basic circuit can be used to implement the following functions: power supply. Show that vO is given by vO = − Rf 16 [20 a0 + 21a1 + 22 a2 + 23a3] where Rf is in kilohms. Find the value of Rf so that vO ranges from 0 to −12 volts. (a) vO = –(v1 + 2v2 + 3v3) (b) vO = –(v1 + v2 + 2v3 + 2v4) (c) vO = –(v1 + 5v2) (d) vO = –6v1 In each case find the input resistance seen by each of the signal sources supplying v1, v2, v3, and v4. Suggest at least two additional summing functions that you can realize with this circuit. How would you realize a summing coefficient that is 0.5? D 2.40 Give a circuit, complete with component values, for a weighted summer that shifts the dc level of a sine-wave signal of 3 sin(ωt) V from zero to −3 V. Assume that in addition to the sine-wave signal you have a dc reference voltage of 1.5 V available. Sketch the output signal waveform. D 2.41 Use two ideal op amps and resistors to implement the summing function vO = v1 + 2v2 – 3v3 – 5v4 D 2.42 In an instrumentation system, there is a need to take the difference between two signals, one of v1 = 2 sin(2π × 60t) + 0.01 sin(2π × 1000t) volts and another of v2 = 2 sin(2π × 60t) − 0.01 sin(2π × 1000t) volts. Draw a circuit that finds the required difference using two op amps and mainly 100-k resistors. Since it is desirable to amplify the 1000-Hz component in the process, arrange to provide an overall gain of 100 as well. The op amps available are ideal except that their output voltage swing is limited to ±10 V. *2.43 Figure P2.43 shows a circuit for a digital-to-analog converter (DAC). The circuit accepts a 4-bit input binary word a3a2a1a0, where a0, a1, a2, and a3 take the values of 0 or 1, and it provides an analog output voltage vO proportional to the value of the digital input. Each of the bits of the input word controls the correspondingly numbered switch. For instance, if a2 is 0 then switch S2 connects the 20-k resistor to ground, while if a2 is 1 then S2 connects the 20-k resistor to the +5-V Figure P2.43 Section 2.3: The Noninverting Configuration D 2.44 Given an ideal op amp to implement designs for the following closed-loop gains, what values of resistors (R1, R2) should be used? Where possible, use at least one 10-k resistor as the smallest resistor in your design. (a) +1 V/V (b) +2 V/V (c) +21 V/V (d) +100 V/V D 2.45 Design a circuit based on the topology of the noninverting amplifier to obtain a gain of +1.5 V/V, using only 10-k resistors. Note that there are two possibilities. Which of these can be easily converted to have a gain of = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 2 PROBLEMS either +1.0 V/V or +2.0 V/V simply by short-circuiting a single resistor in each case? D 2.46 Figure P2.46 shows a circuit for an analog voltmeter of very high input resistance that uses an inexpensive moving-coil meter. The voltmeter measures the voltage V applied between the op amp’s positive-input terminal and ground. Assuming that the moving coil produces full-scale deflection when the current passing through it is 100 μA, find the value of R such that a full-scale reading is obtained when V is +10 V. Does the meter resistance shown affect the voltmeter calibration? Problems 123 RP0 Figure P2.47 V Figure P2.46 D *2.48 Design a circuit, using one ideal op amp, whose output is vO = vI1 + 2vI2 − 9vI3 + 4vI4. (Hint: Use a structure similar to that shown in general form in Fig. P2.47.) 2.49 Derive an expression for the voltage gain, vO/vI , of the circuit in Fig. P2.49. R2 D *2.47 (a) Use superposition to show that the output of the circuit in Fig. P2.47 is given by vO = Rf RN 1 vN1 + Rf RN 2 vN2 + ··· + Rf RNn v Nn + 1 + Rf RN RP RP1 vP1 + RP RP2 vP2 + ··· + RP RPn v Pn where RN = RN1 RN2 · · · RNn, and RP = RP1 RP2 · · · RPn RP0 (b) Design a circuit to obtain vO = –4vN1 + vP1 + 3vP2 R1 Ϫ ϩ ϩ ϩ R3 vO vI R4 Ϫ Ϫ Figure P2.49 2.50 For the circuit in Fig. P2.50, use superposition to find vO in terms of the input voltages v1 and v2. Assume an ideal op amp. For v1 = 10 sin(2π × 60t) − 0.1 sin(2π × 1000t), volts v2 = 10 sin(2π × 60t) + 0.1 sin(2π × 1000t), volts The smallest resistor used should be 10 k . find vO. = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem 124 Chapter 2 Operational Amplifiers CHAPTER 2 PROBLEMS 10R 10R Figure P2.50 In each case find the load current and the current supplied by the source. Where does the load current come from in case (b)? 2.54 Derive an expression for the gain of the voltage follower of Fig. 2.14, assuming the op amp to be ideal except for having a finite gain A. Calculate the value of the closed-loop gain for A = 1000, 100, and 10. In each case find the percentage error in gain magnitude from the nominal value of unity. 2.55 Complete the following table for feedback amplifiers created using one ideal op amp. Note that Rin signifies input resistance and R1 and R2 are feedback-network resistors as labeled in the inverting and noninverting configurations. D 2.51 The circuit shown in Fig. P2.51 utilizes a 10-k potentiometer to realize an adjustable-gain amplifier. Derive an expression for the gain as a function of the potentiometer setting x. Assume the op amp to be ideal. What is the range of gains obtained? Show how to add a fixed resistor so that the gain range can be 1 to 11 V/V. What should the resistor value be? Case Gain Rin a −10 V/V 10 k b −1 V/V c −2 V/V d +1 V/V ∞ e +2 V/V f +11 V/V g −0.5 V/V 20 k R1 100 k 100 k R2 200 k 100 k Figure P2.51 D 2.52 Given the availability of resistors of value 1 k and 10 k only, design a circuit based on the noninverting configuration to realize a gain of +10 V/V. What is the input resistance of your amplifier? 2.53 It is required to connect a 10-V source with a source resistance of 1 M to a 1-k load. Find the voltage that will appear across the load if: (a) The source is connected directly to the load. (b) A unity-gain op-amp buffer is inserted between the source and the load. D 2.56 A noninverting op-amp circuit with nominal gain of 10 V/V uses an op amp with open-loop gain of 100 V/V and a lowest-value resistor of 10 k . What closed-loop gain actually results? With what value resistor can which resistor be shunted to achieve the nominal gain? If in the manufacturing process, an op amp of gain 200 V/V were used, what closed-loop gain would result in each case (the uncompensated one, and the compensated one)? 2.57 Use Eq. (2.11) to show that if the reduction in the closed-loop gain G from the nominal value G0 = 1 + R2/R1 is to be kept less than x% of G0, then the open-loop gain of the op amp must exceed G0 by at least a factor F = (100/x) − 1 100/x. Find the required F for x = 0.01, 0.1, 1, and 10. Utilize these results to find for each value of x the minimum required open-loop gain to obtain closed-loop gains of 1, 10, 102, 103, and 104 V/V. 2.58 For each of the following combinations of op-amp open-loop gain A and nominal closed-loop gain G0, calculate the actual closed-loop gain G that is achieved. Also, calculate the percentage by which |G| falls short of the nominal gain magnitude |G0|. = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 2 PROBLEMS Problems 125 Case a b c d e f g G0 (V/V) −1 +1 −1 +10 −10 −10 +1 A (V/V) 10 10 100 10 100 1000 2 2.59 Figure P2.59 shows a circuit that provides an output voltage vO whose value can be varied by turning the wiper of the 100-k potentiometer. Find the range over which vO can be varied. If the potentiometer is a “20-turn” device, find the change in vO corresponding to each turn of the pot. (a) 1 V/V (b) 5 V/V (c) 100 V/V (d) 0.5 V/V 2.62 For the circuit shown in Fig. P2.62, express vO as a function of v1 and v2. What is the input resistance seen by v1 alone? By v2 alone? By a source connected between the two input terminals? By a source connected to both input terminals simultaneously? 25 Figure P2.62 25 Figure P2.59 Section 2.4: Difference Amplifiers 2.60 Find the voltage gain vO/vId for the difference amplifier of Fig. 2.16 for the case R1 = R3 = 5 k and R2 = R4 = 100 k . What is the differential input resistance Rid? If the two key resistance ratios (R2/R1) and (R4/R3) are different from each other by 1%, what do you expect the common-mode gain Acm to be? Also, find the CMRR in this case. Neglect the effect of the ratio mismatch on the value of Ad. D 2.61 Using the difference amplifier configuration of Fig. 2.16 and assuming an ideal op amp, design the circuit to provide the following differential gains. In each case, the differential input resistance should be 20 k . 2.63 Consider the difference amplifier of Fig. 2.16 with the two input terminals connected together to an input common-mode signal source. For R2/R1 = R4/R3, show that the input common-mode resistance is (R3 + R4) (R1 + R2). 2.64 Consider the circuit of Fig. 2.16, and let each of the vI1 and vI2 signal sources have a series resistance Rs. What condition must apply in addition to the condition in Eq. (2.15) in order for the amplifier to function as an ideal difference amplifier? *2.65 For the difference amplifier shown in Fig. P2.62, let all the resistors be 10 k ± x%. Find an expression for the worst-case common-mode gain that results. Evaluate this for x = 0.1, 1, and 5. Also, evaluate the resulting CMRR in each case. Neglect the effect of resistor tolerances on Ad. 2.66 For the difference amplifier of Fig. 2.16, show that if each resistor has a tolerance of ±100 e% (i.e., for, say, a 5% resistor, e = 0.05) then the worst-case CMRR is given approximately by K +1 CMRR 20 log 4e = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem 126 Chapter 2 Operational Amplifiers CHAPTER 2 PROBLEMS where K is the nominal (ideal) value of the ratios (R2/R1) and (R4/R3). Calculate the value of worst-case CMRR for an amplifier designed to have a differential gain of ideally 100 V/V, assuming that the op amp is ideal and that 1% resistors are used. What resistor tolerance is needed if a CMRR of 80 dB is required? D *2.67 Design the difference amplifier circuit of Fig. 2.16 to realize a differential gain of 1000, a differential input resistance of 2 k , and a minimum CMRR of 88 dB. Assume the op amp to be ideal. Specify both the resistor values and their required tolerance (e.g., better than x%). *2.68 (a) Find Ad and Acm for the difference amplifier circuit shown in Fig. P2.68. (b) If the op amp is specified to operate properly as long as the common-mode voltage at its positive and negative inputs falls in the range ±2.5 V, what is the corresponding limitation on the range of the input common-mode signal vIcm? (This is known as the common-mode range of the differential amplifier.) (c) The circuit is modified by connecting a 10-k resistor between node A and ground, and another 10-k resistor between node B and ground. What will now be the values of Ad, Acm, and the input common-mode range? β R6|(R5 + R6). Show that the differential gain is given by Ad ≡ vO v Id = 1 1−β (Hint: Use superposition.) Design the circuit to obtain a differential gain of 10 V/V and differential input resistance of 2 M . Select values for R, R5, and R6, such that (R5 + R6) ≤ R/100. R v1 Ϫ vId ϩ v2 R R Ϫ ϩ R vO R5 bvO R6 Figure P2.69 100 k⍀ 100 k⍀ vI1 Ϫ A vO vI2 100 k⍀ B ϩ 100 k⍀ *2.70 Figure P2.70 shows a modified version of the difference amplifier. The modified circuit includes a resistor RG, which can be used to vary the gain. Show that the differential voltage gain is given by vO = −2 R2 1 + R2 v Id R1 RG Figure P2.68 (Hint: The virtual short circuit at the op-amp input causes the current through the R1 resistors to be vId/2R1). D *2.69 To obtain a high-gain, high-input-resistance difference amplifier, the circuit in Fig. P2.69 employs positive feedback, in addition to the negative feedback provided by the resistor R connected from the output to the negative input of the op amp. Specifically, a voltage divider (R5, R6) connected across the output feeds a fraction β of the output, that is, a voltage βvO, back to the positive-input terminal of the op amp through a resistor R. Assume that R5 and R6 are much smaller than R so that the current through R is much lower than the current in the voltage divider, with the result that vId Figure P2.70 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 2 PROBLEMS Problems 127 D *2.71 The circuit shown in Fig. P2.71 is a representation of a versatile, commercially available IC, the INA105, manufactured by Burr-Brown and known as a differential amplifier module. It consists of an op amp and precision, laser-trimmed, metal-film resistors. The circuit can be configured for a variety of applications by the appropriate connection of terminals A, B, C, D, and O. (a) Show how the circuit can be used to implement a difference amplifier of unity gain. (b) Show how the circuit can be used to implement single-ended amplifiers with gains: (i) −1 V/V (ii) +1 V/V (iii) +2 V/V (iv) +1/2 V/V Avoid leaving a terminal open-circuited, for such a terminal may act as an “antenna,” picking up interference and noise through capacitive coupling. Rather, find a convenient node to connect such a terminal in a redundant way. When more than one circuit implementation is possible, comment on the relative merits of each, taking into account such considerations as dependence on component matching and input resistance. 2.74 (a) Expressing vI1 and vI2 in terms of differential and common-mode components, find vO1 and vO2 in the circuit in Fig. 2.20(a) and hence find their differential component v O2 − vO1 and their common-mode component 1 2 (v O1 + vO2). Now find the differential gain and the common-mode gain of the first stage of this instrumentation amplifier and hence the CMRR. (b) Repeat for the circuit in Fig. 2.20(b), and comment on the difference between the two circuits. *2.75 For an instrumentation amplifier of the type shown in Fig. 2.20(b), a designer proposes to make R2 = R3 = R4 = 100 k , and 2R1 = 10 k . For ideal components, what difference-mode gain, common-mode gain, and CMRR result? Reevaluate the worst-case values for these for the situation in which all resistors are specified as ±1% units. Repeat the latter analysis for the case in which 2R1 is reduced to 1 k . What do you conclude about the effect of the gain of the first stage on CMRR? (Hint: Eq. (2.19) can be used to evaluate Acm of the second stage.) D 2.76 Design the instrumentation-amplifier circuit of Fig. 2.20(b) to realize a differential gain, variable in the range 2 to 100, utilizing a 100-k pot as variable resistor. 25 k⍀ A B 25 k⍀ Figure P2.71 25 k⍀ C Ϫ O ϩ D 25 k⍀ *2.77 The circuit shown in Fig. P2.77 is intended to supply a voltage to floating loads (those for which both terminals are ungrounded) while making greatest possible use of the available power supply. 20 k⍀ 2.72 Consider the instrumentation amplifier of Fig. 2.20(b) with a common-mode input voltage of +3 V (dc) and a differential input signal of 100-mV peak sine wave. Let 2R1 = 2 k , R2 = 50 k , R3 = R4 = 10 k . Find the voltage at every node in the circuit. 30 k⍀ 2.73 (a) Consider the instrumentation amplifier circuit of Fig. 2.20(a). If the op amps are ideal except that their outputs saturate at ±12 V, in the manner shown in Fig. 1.14, find the maximum allowed input common-mode signal for the case R1 = 1 k and R2 = 100 k . (b) Repeat (a) for the circuit in Fig. 2.20(b), and comment on the difference between the two circuits. Figure P2.77 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 2 PROBLEMS 128 Chapter 2 Operational Amplifiers (a) Assuming ideal op amps, sketch the voltage waveforms at nodes B and C for a 1-V peak-to-peak sine wave applied at A. Also sketch vO. (b) What is the voltage gain vO/vI ? (c) Assuming that the op amps operate from ±15-V power supplies and that their output saturates at ±14 V (in the manner shown in Fig. 1.14), what is the largest sine-wave output that can be accommodated? Specify both its peak-to-peak and rms values. (c) If the frequency is lowered by a factor of 10 from that found in (a), by what factor does the output voltage change, and in what direction (smaller or larger)? (d) What is the phase relation between the input and output in situation (c)? D 2.80 Design a Miller integrator with a time constant of 1 s and an input resistance of 100 k . A dc voltage of −1 volt is applied at the input at time 0, at which moment vO = −10 V. How long does it take the output to reach 0 V? +10 V? *2.78 The two circuits in Fig. P2.78 are intended to function as voltage-to-current converters; that is, they supply the load impedance ZL with a current proportional to vI and independent of the value of ZL. Show that this is indeed the case, and find for each circuit iO as a function of vI . Comment on the differences between the two circuits. Section 2.5: Integrators and Differentiators 2.79 A Miller integrator incorporates an ideal op amp, a resistor R of 10 k , and a capacitor C of 1 nF. A sine-wave signal is applied to its input. 2.81 An op-amp-based inverting integrator is measured at 10 kHz to have a voltage gain of −100 V/V. At what frequency is its gain reduced to −1 V/V? What is the integrator time constant? D 2.82 Design a Miller integrator that has a unity-gain frequency of 10 krad/s and an input resistance of 100 k . Sketch the output you would expect for the situation in which, with output initially at 0 V, a 2-V, 100-μs pulse is applied to the input. Characterize the output that results when a sine wave 2 sin 104t is applied to the input. (a) At what frequency (in Hz) are the input and output signals equal in amplitude? (b) At that frequency, how does the phase of the output sine wave relate to that of the input? D 2.83 Design a Miller integrator whose input resistance is 10 k and unity-gain frequency is 100 kHz. What components are needed? For long-term stability, a feedback resistor is introduced across the capacitor to limit the dc gain ϩ vI R ϩ Ϫ ZL iO R1 ϩ vI Ϫ R1 R1 Ϫ ϩ R1 R Ϫ Ϫ ϩ Ϫ Ϫ ϩ ϩ ZL iO (a) (b) Figure P2.78 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 2 PROBLEMS Problems 129 to 40 dB. What is its value? What is the associated lower 3-dB frequency? Sketch and label the output that results with a 10-μs, 1-V positive-input pulse (initially at 0 V) with (a) no dc stabilization (but with the output initially at 0 V) and (b) the feedback resistor connected. *2.84 A Miller integrator whose input and output voltages are initially zero and whose time constant is 1 ms is driven by the signal shown in Fig. P2.84. Sketch and label the output waveform that results. Indicate what happens if the input levels are ±2 V, with the time constant the same (1 ms) and with the time constant raised to 2 ms. low-pass active filter. Derive the transfer function and show that the dc gain is (−R2/R1) and the 3-dB frequency ω0 = 1/CR2. Design the circuit to obtain an input resistance of 10 k , a dc gain of 40 dB, and a 3-dB frequency of 1 kHz. At what frequency does the magnitude of the transfer function reduce to unity? Vo Figure P2.86 Figure P2.84 2.85 Consider a Miller integrator having a time constant of 1 ms and an output that is initially zero, when fed with a string of pulses of 10-μs duration and 1-V amplitude rising from 0 V (see Fig. P2.85). Sketch and label the output waveform resulting. How many pulses are required for an output voltage change of 1 V? *2.87 Show that a Miller integrator implemented with an op amp with open-loop gain A0 has a low-pass STC transfer function. What is the pole frequency of the STC function? How does this compare with the pole frequency of the ideal integrator? If an ideal Miller integrator is fed with a −1-V pulse signal with a width T = CR, what will the output voltage be at t = T ? Assume that at t = 0, vO = 0. Repeat for an integrator with an op amp having A0 = 1000. 2.88 A differentiator utilizes an ideal op amp, a 10-k resistor, and a 1-nF capacitor. What is the frequency f0 (in Hz) at which its input and output sine-wave signals have equal magnitude? What is the output signal for a 1-V peak-to-peak sine-wave input with frequency equal to 10f0? 2.89 An op-amp differentiator with 1-ms time constant is driven by the rate-controlled step shown in Fig. P2.89. Assuming vO to be zero initially, sketch and label its waveform. Figure P2.85 D 2.86 Figure P2.86 shows a circuit that performs a low-pass STC function. Such a circuit is known as a first-order, 0.2 Figure P2.89 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem 130 Chapter 2 Operational Amplifiers CHAPTER 2 PROBLEMS 2.90 An op-amp differentiator, employing the circuit shown in Fig. 2.27(a), has R = 20 k and C = 0.1 μF. When a triangle wave of ±1-V peak amplitude at 1 kHz is applied to the input, what form of output results? What is its frequency? What is its peak amplitude? What is its average value? What value of R is needed to cause the output to have a 12-V peak amplitude? for the transfer function in the following frequency regions: (a) ω ω1 (b) ω1 ω ω2 (c) ω ω2 2.91 Use an ideal op amp to design a differentiation circuit for which the time constant is 10−3 s using a 10-nF capacitor. What are the gains and phase shifts found for this circuit at one-tenth and 10 times the unity-gain frequency? A series input resistor is added to limit the gain magnitude at high frequencies to 100 V/V. What is the associated 3-dB frequency? What gain and phase shift result at 10 times the unity-gain frequency? D 2.92 Figure P2.92 shows a circuit that performs the high-pass, single-time-constant function. Such a circuit is known as a first-order high-pass active filter. Derive the transfer function and show that the high-frequency gain is (−R2/R1) and the 3-dB frequency ω0 = 1/CR1. Design the circuit to obtain a high-frequency input resistance of 1 k , a high-frequency gain of 40 dB, and a 3-dB frequency of 2 kHz. At what frequency does the magnitude of the transfer function reduce to unity? Vo Figure P2.93 Use these approximations to sketch a Bode plot for the magnitude response. Observe that the circuit performs as an amplifier whose gain rolls off at the low-frequency end in the manner of a high-pass STC network, and at the high-frequency end in the manner of a low-pass STC network. Design the circuit to provide a gain of 40 dB in the “middle-frequency range,” a low-frequency 3-dB point at 200 Hz, a high-frequency 3-dB point at 200 kHz, and an input resistance (at ω ω1) of 2 k . Vo Section 2.6: DC Imperfections Figure P2.92 D **2.93 Derive the transfer function of the circuit in Fig. P2.93 (for an ideal op amp) and show that it can be written in the form 2.94 An op amp wired in the inverting configuration with the input grounded, having R2 = 100 k and R1 = 2 k , has an output dc voltage of −0.2 V. If the input bias current is known to be very small, find the input offset voltage. 2.95 A noninverting amplifier with a gain of 100 uses an op amp having an input offset voltage of ±2 mV. Find the output when the input is 0.01 sin ωt, volts. Vo = −R2/R1 Vi [1 + (ω1/jω)][1 + j(ω/ω2)] where ω1 = 1/C1R1 and ω2 = 1/C2R2. Assuming that the circuit is designed such that ω2 ω1, find approximate expressions 2.96 A noninverting amplifier with a closed-loop gain of 1000 is designed using an op amp having an input offset voltage of 3 mV and output saturation levels of ±12 V. What is the maximum amplitude of the sine wave that can be applied at the input without the output clipping? If the amplifier is = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem Problems 131 CHAPTER 2 PROBLEMS capacitively coupled in the manner indicated in Fig. 2.36, what would the maximum possible amplitude be? 2.97 An op amp connected in a closed-loop inverting configuration having a gain of 1000 V/V and using relatively small-valued resistors is measured with input grounded to have a dc output voltage of −1.8 V. What is its input offset voltage? Prepare an offset-voltage-source sketch resembling that in Fig. 2.28. Be careful of polarities. through a capacitor C? If, instead, a large capacitor is placed in series with a 1-k resistor, what does the output offset become? 2.98 A particular inverting amplifier with nominal gain of −100 V/V uses an imperfect op amp in conjunction with 100-k and 10-M resistors. The output voltage is found to be +5.3 V when measured with the input open and +5 V with the input grounded. (a) What is the bias current of this amplifier? In what direction does it flow? (b) Estimate the value of the input offset voltage. (c) A 10-M resistor is connected between the positive-input terminal and ground. With the input left floating (disconnected), the output dc voltage is measured to be −0.6 V. Estimate the input offset current. D *2.99 A noninverting amplifier with a gain of +10 V/V using 100 k as the feedback resistor operates from a 5-k source. For an amplifier offset voltage of 0 mV, but with a bias current of 2 μA and an offset current of 0.2 μA, what range of outputs would you expect? Indicate where you would add an additional resistor to compensate for the bias currents. What does the range of possible outputs then become? A designer wishes to use this amplifier with a 15-k source. In order to compensate for the bias current in this case, what resistor would you use? And where? D 2.100 The circuit of Fig. 2.36 is used to create an ac-coupled noninverting amplifier with a gain of 100 V/V using resistors no larger than 100 k . What values of R1, R2, and R3 should be used? For a break frequency due to C1 at 100 Hz, and that due to C2 at 10 Hz, what values of C1 and C2 are needed? *2.101 Consider the difference amplifier circuit in Fig. 2.16. Let R1 = R3 = 10 k and R2 = R4 = 1 M . If the op amp has VOS = 5 mV, IB = 1 μA, and IOS = 0.2 μA, find the worst-case (largest) dc offset voltage at the output. *2.102 The circuit shown in Fig. P2.102 uses an op amp having a ±3-mV offset. What is its output offset voltage? What does the output offset become with the input ac coupled Figure P2.102 2.103 Using offset-nulling facilities provided for the op amp, a closed-loop amplifier with gain of +1000 is adjusted at 25°C to produce zero output with the input grounded. If the input offset-voltage drift is specified to be 20 μV/°C, what output would you expect at 0°C and at 100°C? While nothing can be said separately about the polarity of the output offset at either 0 or 75°C, what would you expect their relative polarities to be? 2.104 An op amp is connected in a closed loop with gain of +100 utilizing a feedback resistor of 1 M . (a) If the input bias current is 200 nA, what output voltage results with the input grounded? (b) If the input offset voltage is ±2 mV and the input bias current as in (a), what is the largest possible output that can be observed with the input grounded? (c) If bias-current compensation is used, what is the value of the required resistor? If the offset current is no more than one-tenth the bias current, what is the resulting output offset voltage (due to offset current alone)? (d) With bias-current compensation as in (c) in place, what is the largest dc voltage at the output due to the combined effect of offset voltage and offset current? *2.105 An op amp intended for operation with a closed-loop gain of –100 V/V uses resistors of 10 k and 1 M with a bias-current-compensation resistor R3. What should the value of R3 be? With input grounded, the output offset voltage is found to be +0.30 V. Estimate the input offset current assuming zero input offset voltage. If the input offset voltage = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 2 PROBLEMS 132 Chapter 2 Operational Amplifiers can be as large as 1 mV of unknown polarity, what range of offset current is possible? 2.106 A Miller integrator with R = 10 k and C = 10 nF is implemented by using an op amp with VOS = 2 mV, IB = 0.1 μA, and IOS = 20 nA. To provide a finite dc gain, a 1-M resistor is connected across the capacitor. (a) To compensate for the effect of IB, a resistor is connected in series with the positive-input terminal of the op amp. What should its value be? (b) With the resistor of (a) in place, find the worst-case dc output voltage of the integrator when the input is grounded. Section 2.7: Effect of Finite Open-Loop Gain and Bandwidth on Circuit Performance 2.107 The data in the following table apply to internally compensated op amps. Fill in the blank entries. A0 105 106 2 × 105 fb (Hz) 102 103 10−1 10 ft (Hz) 106 108 106 2.108 A measurement of the open-loop gain of an internally compensated op amp at very low frequencies shows it to be 98 dB; at 100 kHz, this shows it is 40 dB. Estimate values for A0, fb, and ft. 2.109 Measurements of the open-loop gain of a compensated op amp intended for high-frequency operation indicate that the gain is 4 × 103 at 100 kHz and 20 × 103 at 10 kHz. Estimate its 3-dB frequency, its unity-gain frequency, and its dc gain. 2.110 Measurements made on the internally compensated amplifiers listed below provide the dc gain and the frequency at which the gain has dropped by 20 dB. For each, what are the 3 dB and unity-gain frequencies? (a) 2 × 105 V/V and 5 × 102 Hz (b) 20 × 105 V/V and 10 Hz (c) 1800 V/V and 0.1 MHz (d) 100 V/V and 0.1 GHz (e) 25 V/mV and 250 kHz 2.111 An inverting amplifier with nominal gain of −50 V/V employs an op amp having a dc gain of 104 and a unity-gain frequency of 106 Hz. What is the 3-dB frequency f3dB of the closed-loop amplifier? What is its gain at 0.1 f3dB and at 10 f3dB? 2.112 A particular op amp, characterized by a gain–bandwidth product of 20 MHz, is operated with a closed-loop gain of +100 V/V. What 3-dB bandwidth results? At what frequency does the closed-loop amplifier exhibit a −6° phase shift? A −84° phase shift? 2.113 Find the ft required for internally compensated op amps to be used in the implementation of closed-loop amplifiers with the following nominal dc gains and 3-dB bandwidths: (a) −50 V/V; 100 kHz (b) +50 V/V; 100 kHz (c) +2 V/V; 5 MHz (d) −2 V/V; 5 MHz (e) −1000 V/V; 10 kHz (f) +1 V/V; 1 MHz (g) −1 V/V; 1 MHz 2.114 A noninverting op-amp circuit with a gain of 96 V/V is found to have a 3-dB frequency of 8 kHz. For a particular system application, a bandwidth of 32 kHz is required. What is the highest gain available under these conditions? 2.115 Consider a unity-gain follower utilizing an internally compensated op amp with ft = 2 MHz. What is the 3-dB frequency of the follower? At what frequency is the gain of the follower 1% below its low-frequency magnitude? If the input to the follower is a 1-V step, find the 10% to 90% rise time of the output voltage. (Note: The step response of STC low-pass networks is discussed in Appendix E. Specifically, note that the 10%–90% rise time of a low-pass STC circuit with a time constant τ is 2.2τ .) D *2.116 It is required to design a noninverting amplifier with a dc gain of 10. When a step voltage of 100 mV is applied at the input, it is required that the output be within 1% of its final value of 1 V in at most 200 ns. What must the ft of the op amp be? (Note: The step response of STC low-pass networks is discussed in Appendix E.) D *2.117 This problem illustrates the use of cascaded closed-loop amplifiers to obtain an overall bandwidth greater than can be achieved using a single-stage amplifier with the same overall gain. = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem Problems 133 CHAPTER 2 PROBLEMS (a) Show that cascading two identical amplifier stages, each having a low-pass STC frequency response with a 3-dB frequency f1, results in an overall amplifier with a 3-dB frequency given by √ f3dB = 2–1 f1 (b) It is required to design a noninverting amplifier with a dc gain of 40 dB utilizing a single internally compensated op amp with ft = 2 MHz. What is the 3-dB frequency obtained? (c) Redesign the amplifier of (b) by cascading two identical noninverting amplifiers each with a dc gain of 20 dB. What is the 3-dB frequency of the overall amplifier? Compare this to the value obtained in (b) above. D **2.118 A designer, wanting to achieve a stable gain of 100 V/V at 5 MHz, considers her choice of amplifier topologies. Whatunity-gainfrequencywouldasingleoperationalamplifier require to satisfy her need? Unfortunately, the best available amplifier has an ft of 40 MHz. How many such amplifiers connected in a cascade of identical noninverting stages would she need to achieve her goal? What is the 3-dB frequency of each stage she can use? What is the overall 3-dB frequency? 2.119 Consider the use of an op amp with a unity-gain frequency ft in the realization of: (a) An inverting amplifier with dc gain of magnitude K. (b) A noninverting amplifier with a dc gain of K. In each case find the 3-dB frequency and the gain–bandwidth product (GBP ≡ |Gain| × f3dB). Comment on the results. *2.120 Consider an inverting summer with two inputs V1 and V2 and with Vo = −(V1+ 3V2). Find the 3-dB frequency of each of the gain functions Vo/V1 and Vo/V2 in terms of the op amp ft. (Hint: In each case, the other input to the summer can be set to zero—an application of superposition.) Section 2.8: Large-Signal Operation of Op Amps 2.121 A particular op amp using ±15-V supplies operates linearly for outputs in the range −14 V to +14 V. If used in an inverting amplifier configuration of gain −100, what is the rms value of the largest possible sine wave that can be applied at the input without output clipping? 2.122 Consider an op amp connected in the inverting configuration to realize a closed-loop gain of −100 V/V utilizing resistors of 1 k and 100 k . A load resistance RL is connected from the output to ground, and a low-frequency sine-wave signal of peak amplitude Vp is applied to the input. Let the op amp be ideal except that its output voltage saturates at ±10 V and its output current is limited to the range ±20 mA. (a) For RL = 1 k , what is the maximum possible value of Vp while an undistorted output sinusoid is obtained? (b) Repeat (a) for RL = 200 . (c) If it is desired to obtain an output sinusoid of 10-V peak amplitude, what minimum value of RL is allowed? 2.123 An op amp having a slew rate of 10 V/μs is to be used in the unity-gain follower configuration, with input pulses that rise from 0 to 2 V. What is the shortest pulse that can be used while ensuring full-amplitude output? For such a pulse, describe the output resulting. 2.124 For operation with 10-V output pulses with the requirement that the sum of the rise and fall times represent only 20% of the pulse width (at half-amplitude), what is the slew-rate requirement for an op amp to handle pulses 2 μs wide? (Note: The rise and fall times of a pulse signal are usually measured between the 10%- and 90%-height points.) 2.125 What is the highest frequency of a triangle wave of 10-V peak-to-peak amplitude that can be reproduced by an op amp whose slew rate is 20 V/μs? For a sine wave of the same frequency, what is the maximum amplitude of output signal that remains undistorted? 2.126 For an amplifier having a slew rate of 40 V/μs, what is the highest frequency at which a 20-V peak-to-peak sine wave can be produced at the output? D *2.127 In designing with op amps one has to check the limitations on the voltage and frequency ranges of operation of the closed-loop amplifier, imposed by the op-amp finite bandwidth (ft), slew rate (SR), and output saturation (Vomax). This problem illustrates the point by considering the use of an op amp with ft = 20 MHz, SR = 10 V/μs, and Vomax = 10 V in the design of a noninverting amplifier with a nominal gain of 10. Assume a sine-wave input with peak amplitude Vi. (a) If Vi = 0.5 V, what is the maximum frequency before the output distorts? (b) If f = 200 kHz, what is the maximum value of Vi before the output distorts? (c) If Vi = 50 mV, what is the useful frequency range of operation? (d) If f = 50 kHz, what is the useful input voltage range? = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 3 Semiconductors Introduction 135 3.1 Intrinsic Semiconductors 136 3.2 Doped Semiconductors 139 3.3 Current Flow in Semiconductors 142 3.4 The pn Junction 148 3.5 The pn Junction with an Applied Voltage 155 3.6 Capacitive Effects in the pn Junction 164 Summary 168 Problems 171 IN THIS CHAPTER YOU WILL LEARN 1. The basic properties of semiconductors and in particular silicon, which is the material used to make most of today’s electronic circuits. 2. How doping a pure silicon crystal dramatically changes its electrical conductivity, which is the fundamental idea underlying the use of semiconductors in the implementation of electronic devices. 3. The two mechanisms by which current flows in semiconductors: drift and diffusion of charge carriers. 4. The structure and operation of the pn junction; a basic semiconductor structure that implements the diode and plays a dominant role in transistors. Introduction Thus far we have dealt with electronic circuits, and notably amplifiers, as system building blocks. For instance, in Chapter 2 we learned how to use op amps to design interesting and useful circuits, taking advantage of the terminal characteristics of the op amp and without any knowledge of what is inside the op-amp package. Though interesting and motivating, this approach has its limitations. Indeed, to achieve our goal of preparing the reader to become a proficient circuit designer, we have to go beyond this black-box or system-level abstraction and learn about the basic devices from which electronic circuits are assembled, namely, diodes (Chapter 4) and transistors (Chapters 5 and 6). These solid-state devices are made using semiconductor materials, predominantly silicon. In this chapter, we briefly introduce the properties and physics of semiconductors. The objective is to provide a basis for understanding the physical operation of diodes and transistors in order to enable their effective use in the design of circuits. Although many of the concepts studied in this chapter apply to semiconductor materials in general, our treatment is heavily biased toward silicon, simply because it is the material used in the vast majority of microelectronic circuits. To complement the material presented here, Appendix A provides a description of the integrated-circuit fabrication process. As discussed in Appendix A, whether our circuit consists of a single transistor or is an integrated circuit containing more than 2 billion transistors, it is fabricated in a single silicon crystal, which gives rise to the name monolithic circuit. This chapter therefore begins with a study of the crystal structure of semiconductors and introduces the two types of charge carriers available for current conduction: electrons and holes. The most significant property of semiconductors is that their conductivity can be varied over a very wide range through the introduction of 135 136 Chapter 3 Semiconductors controlled amounts of impurity atoms into the semiconductor crystal in a process called doping. Doped semiconductors are discussed in Section 3.2. This is followed by the study in Section 3.3 of the two mechanisms for current flow in semiconductors, namely, carrier drift and carrier diffusion. Armed with these basic semiconductor concepts, we spend the remainder of the chapter on the study of an important semiconductor structure: the pn junction. In addition to being essentially a diode, the pn junction is the basic element of the bipolar junction transistor (BJT, Chapter 6) and plays an important role in the operation of field-effect transistors (FETs, Chapter 5). 3.1 Intrinsic Semiconductors As their name implies, semiconductors are materials whose conductivity lies between that of conductors, such as copper, and insulators, such as glass. There are two kinds of semiconductors: single-element semiconductors, such as germanium and silicon, which are in group IV in the periodic table; and compound semiconductors, such as gallium-arsenide, which are formed by combining elements from groups III and V or groups II and VI. Compound semiconductors are useful in special electronic circuit applications as well as in applications that involve light, such as light-emitting diodes (LEDs). Of the two elemental semiconductors, germanium was used in the fabrication of very early transistors (late 1940s, early 1950s). It was quickly supplanted, however, with silicon, on which today’s integrated-circuit technology is almost entirely based. For this reason, we will deal mostly with silicon devices throughout this book.1 A silicon atom has four valence electrons, and thus it requires another four to complete its outermost shell. This is achieved by sharing one of its valence electrons with each of its four neighboring atoms. Each pair of shared electrons forms a covalent bond. The result is that a crystal of pure or intrinsic silicon has a regular lattice structure, where the atoms are held in their position by the covalent bonds. Figure 3.1 shows a two-dimensional representation of such a structure. At sufficiently low temperatures, approaching absolute zero (0 K), all the covalent bonds are intact and no electrons are available to conduct electric current. Thus, at such low temperatures, the intrinsic silicon crystal behaves as an insulator. At room temperature, sufficient thermal energy exists to break some of the covalent bonds, a process known as thermal generation. As shown in Fig. 3.2, when a covalent bond is broken, an electron is freed. The free electron can wander away from its parent atom, and it becomes available to conduct electric current if an electric field is applied to the crystal. As the electron leaves its parent atom, it leaves behind a net positive charge, equal to the magnitude of the electron charge. Thus, an electron from a neighboring atom may be attracted to this positive charge, and leaves its parent atom. This action fills up the “hole” that existed in the ionized atom but creates a new hole in the other atom. This process may repeat itself, with the result that we effectively have a positively charged carrier, or hole, moving through the silicon crystal structure and being available to conduct electric current. The charge of a hole is equal in magnitude to the charge of an electron. We can thus see that as temperature increases, more covalent bonds are broken and electron–hole pairs are generated. The increase in the numbers of free electrons and holes results in an increase in the conductivity of silicon. 1An exception is the subject of gallium arsenide (GaAs) circuits, which though not covered in this edition of the book, is studied in some detail in material provided on the text website. 3.1 Intrinsic Semiconductors 137 Ϫ Valence electrons Covalent bonds Ϫ ϩ4 Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Silicon atoms Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ Figure 3.1 Two-dimensional representation of the silicon crystal. The circles represent the inner core of silicon atoms, with +4 indicating its positive charge of +4q, which is neutralized by the charge of the four valence electrons. Observe how the covalent bonds are formed by sharing of the valence electrons. At 0 K, all bonds are intact and no free electrons are available for current conduction. Broken covalent bond Covalent bond Ϫ Ϫ Ϫ Valence electrons Free electron Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ 0 ϩ4 Ϫ Ϫ Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ Hole Silicon atoms ϩ4 Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Figure 3.2 At room temperature, some of the covalent bonds are broken by thermal generation. Each broken bond gives rise to a free electron and a hole, both of which become available for current conduction. Thermal generation results in free electrons and holes in equal numbers and hence equal concentrations, where concentration refers to the number of charge carriers per unit volume (cm3). The free electrons and holes move randomly through the silicon crystal structure, and in the process some electrons may fill some of the holes. This process, called recombination, results in the disappearance of free electrons and holes. The recombination rate is 138 Chapter 3 Semiconductors proportional to the number of free electrons and holes, which in turn is determined by the thermal generation rate. The latter is a strong function of temperature. In thermal equilibrium, the recombination rate is equal to the generation rate, and one can conclude that the concentration of free electrons n is equal to the concentration of holes p, n = p = ni (3.1) where ni denotes the number of free electrons and holes in a unit volume (cm3) of intrinsic silicon at a given temperature. Results from semiconductor physics gives ni as ni = BT e 3/2 −Eg /2kT (3.2) where B is a material-dependent parameter that is 7.3 × 1015cm−3K−3/2 for silicon; T is the temperature in K; Eg, a parameter known as the bandgap energy, is 1.12 electron volt (eV) for silicon2; and k is Boltzmann’s constant (8.62 × 10−5 eV/K). It is interesting to know that the bandgap energy Eg is the minimum energy required to break a covalent bond and thus generate an electron-hole pair. Example 3.1 Calculate the value of ni for silicon at room temperature (T 300 K). Solution Substituting the values given above in Eq. (3.2) provides ni = 7.3 × 1015(300)3/2e−1.12/(2×8.62×10−5 ) ×300 = 1.5 × 1010carriers/cm3 Although this number seems large, to place it into context note that silicon has 5 × 1022 atoms/cm3. Thus at room temperature only one in about 5 × 1012 atoms is ionized and contributing a free electron and a hole! Finally, it is useful for future purposes to express the product of the hole and free-electron concentration as pn = ni2 (3.3) where for silicon at room temperature, ni 1.5 × 1010/cm3. As will be seen shortly, this relationship extends to extrinsic or doped silicon as well. 2 Note that 1 eV = 1.6 × 10−19 J. LCDs, THE FACE OF ELECTRONICS: 3.2 Doped Semiconductors 139 The existence of liquid crystals whose color could be changed by means of an external heat source was first reported in 1888 by an Austrian botanical physiologist. The LC idea lay dormant until the late 1940s, however. Subsequent developments in the field of solid-state electronics provided the technology to harness the technique in display media, with the first LCDs being demonstrated by RCA beginning in 1962. Today, LCDs are an essential component in every mobile device as the interface to the world of electronics within. At the other end of the scale, large LCDs are used in flat-panel TVs, and very large LCDs are appearing as “dynamic” wallpaper in museum display settings. EXERCISE 3.1 Calculate the intrinsic carrier density ni for silicon at T = 50 K and 350 K. Ans. 9.6 × 10−39/cm3; 4.15 × 1011/cm3 3.2 Doped Semiconductors The intrinsic silicon crystal described above has equal concentrations of free electrons and holes, generated by thermal generation. These concentrations are far too small for silicon to conduct appreciable current at room temperature. Also, the carrier concentrations and hence the conductivity are strong functions of temperature, not a desirable property in an electronic device. Fortunately, a method was developed to change the carrier concentration in a semiconductor crystal substantially and in a precisely controlled manner. This process is known as doping, and the resulting silicon is referred to as doped silicon. Doping involves introducing impurity atoms into the silicon crystal in sufficient numbers to substantially increase the concentration of either free electrons or holes but with little or no change in the crystal properties of silicon. To increase the concentration of free electrons, n, silicon is doped with an element with a valence of 5, such as phosphorus. The resulting doped silicon is then said to be of n type. To increase the concentration of holes, p, silicon is doped with an element having a valence of 3, such as boron, and the resulting doped silicon is said to be of p type. Figure 3.3 shows a silicon crystal doped with phosphorus impurity. The dopant (phosphorus) atoms replace some of the silicon atoms in the crystal structure. Since the phosphorus atom has five electrons in its outer shell, four of these electrons form covalent bonds with the neighboring atoms, and the fifth electron becomes a free electron. Thus each phosphorus atom donates a free electron to the silicon crystal, and the phosphorus impurity is called a donor. It should be clear, though, that no holes are generated by this process. The net positive charge associated with the phosphorus atom is a bound charge that does not move through the crystal. If the concentration of donor atoms is ND, where ND is usually much greater than ni, the concentration of free electrons in the n-type silicon will be nn ND (3.4) 140 Chapter 3 Semiconductors Ϫ Valence electrons Covalent bonds Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ ϩ5 Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Free electron donated by impurity atom Pentavalent impurity atom (donor) Silicon atoms Ϫ Ϫ Figure 3.3 A silicon crystal doped by a pentavalent element. Each dopant atom donates a free electron and is thus called a donor. The doped semiconductor becomes n type. where the subscript n denotes n-type silicon. Thus nn is determined by the doping concentration and not by temperature. This is not the case, however, for the hole concentration. All the holes in the n-type silicon are those generated by thermal ionization. Their concentration pn can be found by noting that the relationship in Eq. (3.3) applies equally well for doped silicon, provided thermal equilibrium is achieved. Thus for n-type silicon pnnn = ni2 Substituting for nn from Eq. (3.4), we obtain for pn pn ni2 ND (3.5) Thus pn will have the same dependence on temperature as that of ni2. Finally, we note that in n-type silicon the concentration of free electrons nn will be much larger than that of holes. Hence electrons are said to be the majority charge carriers and holes the minority charge carriers in n-type silicon. To obtain p-type silicon in which holes are the majority charge carriers, a trivalent impurity such as boron is used. Figure 3.4 shows a silicon crystal doped with boron. Note that the boron atoms replace some of the silicon atoms in the silicon crystal structure. Since each boron atom has three electrons in its outer shell, it accepts an electron from a neighboring atom, thus forming covalent bonds. The result is a hole in the neighboring atom and a bound negative charge at the acceptor (boron) atom. It follows that each acceptor atom provides a hole. If the acceptor doping concentration is NA, where NA ni, the hole concentration becomes pp NA (3.6) where the subscript p denotes p-type silicon. Thus, here the majority carriers are holes and their concentration is determined by NA. The concentration of minority electrons can be found 3.2 Doped Semiconductors 141 Ϫ Valence electrons Covalent bonds Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ Ϫ ϩ3 Ϫ Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ Ϫ ϩ4 Ϫ Ϫ Ϫ 0 ϩ4 Ϫ Ϫ Silicon atom Trivalent impurity atom (acceptor) Electron accepted from this atom, thus creating a hole Ϫ Ϫ Figure 3.4 A silicon crystal doped with boron, a trivalent impurity. Each dopant atom gives rise to a hole, and the semiconductor becomes p type. by using the relationship ppnp = ni2 and substituting for pp from Eq. (3.6), np ni2 NA (3.7) Thus, the concentration of the minority electrons will have the same temperature dependence as that of ni2. It should be emphasized that a piece of n-type or p-type silicon is electrically neutral; the charge of the majority free carriers (electrons in the n-type and holes in the p-type silicon) are neutralized by the bound charges associated with the impurity atoms. Example 3.2 Consider an n-type silicon for which the dopant concentration ND = 1017/cm3. Find the electron and hole concentrations at T = 300 K. Solution The concentration of the majority electrons is nn ND = 1017/cm3 142 Chapter 3 Semiconductors Example 3.2 continued The concentration of the minority holes is pn ni2 ND In Example 3.1 we found that at T = 300 K, ni = 1.5 × 1010/cm3. Thus, 2 1.5 × 1010 pn = 1017 = 2.25 × 103/cm3 Observe that nn ni and that nn is vastly higher than pn. EXERCISES 3.2 For the situation in Example 3.2, find the electron and hole concentrations at 350 K. You may use the value of ni at T = 350 K found in Exercise 3.1. Ans. nn = 1017/cm3, pn = 1.72 × 106/cm3 3.3 For a silicon crystal doped with boron, what must NA be if at T = 300 K the electron concentration drops below the intrinsic level by a factor of 106? Ans. NA = 1.5 × 1016/cm3 3.3 Current Flow in Semiconductors There are two distinctly different mechanisms for the movement of charge carriers and hence for current flow in semiconductors: drift and diffusion. 3.3.1 Drift Current When an electrical field E is established in a semiconductor crystal, holes are accelerated in the direction of E, and free electrons are accelerated in the direction opposite to that of E. This situation is illustrated in Fig. 3.5. The holes acquire a velocity νp-drift given by νp-drift = μpE (3.8) where μp is a constant called the hole mobility: It represents the degree of ease by which holes move through the silicon crystal in response to the electrical field E. Since velocity has the units of centimeters per second and E has the units of volts per centimeter, we see from Eq. (3.8) that the mobility μp must have the units of centimeters squared per volt-second (cm2/V · s). For intrinsic silicon μp = 480 cm2/ V · s. 3.3 Current Flow in Semiconductors 143 E ϩ Holes Ϫ Electrons x V Figure 3.5 An electric field E established in a bar of silicon causes the holes to drift in the direction of E and the free electrons to drift in the opposite direction. Both the hole and electron drift currents are in the direction of E. The free electrons acquire a drift velocity νn-drift given by νn-drift = −μnE (3.9) where the result is negative because the electrons move in the direction opposite to E. Here μn is the electron mobility, which for intrinsic silicon is about 1350 cm2/V · s. Note that μn is about 2.5 times μp, signifying that electrons move with much greater ease through the silicon crystal than do holes. Let’s now return to the single-crystal silicon bar shown in Fig. 3.5. Let the concentration of holes be p and that of free electrons n. We wish to calculate the current component due to the flow of holes. Consider a plane perpendicular to the x direction. In one second, the hole charge that crosses that plane will be (Aqpνp-drift) coulombs, where A is the cross-sectional area of the silicon bar and q is the magnitude of electron charge. This then must be the hole component of the drift current flowing through the bar, Ip = Aqpνp-drift (3.10) Substituting for νp-drift from Eq. (3.8), we obtain Ip = AqpμpE We are usually interested in the current density Jp, which is the current per unit crosssectional area, Jp = Ip A = qpμpE (3.11) The current component due to the drift of free electrons can be found in a similar manner. Note, however, that electrons drifting from right to left result in a current component from left to right. This is because of the convention of taking the direction of current flow as the direction of flow of positive charge and opposite to the direction of flow of negative charge. Thus, In = −Aqnνn-drift Substituting for νn-drift from Eq. (3.9), we obtain the current density Jn = In/A as Jn = qnμnE (3.12) The total drift current density can now be found by summing Jp and Jn from Eqs. (3.11) and (3.12), J = Jp + Jn = q pμp + nμn E (3.13) This relationship can be written as J =σE (3.14) 144 Chapter 3 Semiconductors or J = E/ρ where the conductivity σ is given by and the resistivity ρ is given by σ = q pμp + nμn 1 1 ρ≡ = σ q pμp + nμn Observe that Eq. (3.15) is a form of Ohm’s law and can be written alternately as ρ=E J V/cm Thus the units of ρ are obtained from: = · cm. A/cm2 Example 3.3 (3.15) (3.16) (3.17) (3.18) Find the resistivity of (a) intrinsic silicon and (b) p-type silicon with NA = 1016/cm3. Use ni = 1.5 × 1010/cm3, and assume that for intrinsic silicon μn = 1350 cm2/V · s and μp = 480 cm2/V · s, and for the doped silicon μn = 1110 cm2/V · s and μp = 400 cm2/V · s. (Note that doping results in reduced carrier mobilities.) Solution (a) For intrinsic silicon, p = n = ni = 1.5 × 1010/cm3 Thus, 1 ρ= q pμp + nμn ρ= 1.6 × 10−19 1 1.5 × 1010 × 480 + 1.5 × 1010 × 1350 = 2.28 × 105 · cm (b) For the p-type silicon pp NA = 1016/cm3 np ni2 = NA 1.5 × 1010 1016 2 = 2.25 × 104/cm3 3.3 Current Flow in Semiconductors 145 Thus, ρ= 1 q pμp + nμn = 1.6 × 10−19 1 1016 × 400 + 2.25 × 104 × 1110 1.6 × 10−19 1 × 1016 × 400 = 1.56 · cm Observe that the resistivity of the p-type silicon is determined almost entirely by the doping concentration. Also observe that doping the silicon reduces its resistivity by a factor of about 104, a truly remarkable change. EXERCISE 3.4 A uniform bar of n-type silicon of 2-μm length has a voltage of 1 V applied across it. If ND = 1016/cm3 and μn = 1350 cm2/V · s, find (a) the electron drift velocity, (b) the time it takes an electron to cross the 2-μm length, (c) the drift-current density, and (d) the drift current in the case that the silicon bar has a cross-sectional area of 0.25 μm2. Ans. 6.75 × 106 cm/s; 30 ps; 1.08 × 104 A/cm2; 27 μA 3.3.2 Diffusion Current Carrier diffusion occurs when the density of charge carriers in a piece of semiconductor is not uniform. For instance, if by some mechanism the concentration of, say, holes, is made higher in one part of a piece of silicon than in another, then holes will diffuse from the region of high concentration to the region of low concentration. Such a diffusion process is like that observed if one drops a few ink drops in a water-filled tank. The diffusion of charge carriers gives rise to a net flow of charge, or diffusion current. As an example, consider the bar of silicon shown in Fig. 3.6(a): By some unspecified process, we have arranged to inject holes into its left side. This continuous hole injection gives rise to and maintains a hole concentration profile such as that shown in Fig. 3.6(b). This profile in turn causes holes to diffuse from left to right along the silicon bar, resulting in a hole current in the x direction. The magnitude of the current at any point is proportional to the slope of the concentration profile, or the concentration gradient, at that point, Jp = −qDp dp(x) dx (3.19) 146 Chapter 3 Semiconductors Hole injection ϩϩϩϩϩϩϩϩϩϩϩϩ ϩϩ ϩ ϩ ϩ ϩ ϩ ϩ ϩ ϩ ϩ ϩ ϩ ϩ ϩ ϩ ϩ ϩ ϩ ϩ ϩ (a) x Hole diffusion Hole current Hole concentration, p 0 x (b) Figure 3.6 A bar of silicon (a) into which holes are injected, thus creating the hole concentration profile along the x axis, shown in (b). The holes diffuse in the positive direction of x and give rise to a hole diffusion current in the same direction. Note that we are not showing the circuit to which the silicon bar is connected. Electron diffusion Electron current Electron concentration, n Figure 3.7 If the electron concentration pro- file shown is established in a bar of silicon, electrons diffuse in the x direction, giving rise to an electron diffusion current in the negative-x 0 x direction. where Jp is the hole-current density (A/cm2), q is the magnitude of electron charge, Dp is a constant called the diffusion constant or diffusivity of holes; and p(x) is the hole concentration at point x. Note that the gradient (dp/dx) is negative, resulting in a positive current in the x direction, as should be expected. In the case of electron diffusion resulting from an electron concentration gradient (see Fig. 3.7), a similar relationship applies, giving the electron-current density, dn(x) Jn = qDn dx (3.20) where Dn is the diffusion constant or diffusivity of electrons. Observe that a negative (dn/dx) gives rise to a negative current, a result of the convention that the positive direction of current is taken to be that of the flow of positive charge (and opposite to that of the flow of negative 3.3 Current Flow in Semiconductors 147 charge). For holes and electrons diffusing in intrinsic silicon, typical values for the diffusion constants are Dp = 12 cm2/s and Dn = 35 cm2/s. At this point the reader is probably wondering where the diffusion current in the silicon bar in Fig. 3.6(a) goes. A good question, as we are not showing how the right-side end of the bar is connected to the rest of the circuit. We will address this and related questions in detail in our discussion of the pn junction in later sections. Example 3.4 Consider a bar of silicon in which a hole concentration profile described by p(x) = p0 e−x/Lp is established. Find the hole-current density at x = 0. Let p0 = 1016/cm3, Lp = 1 μm, and Dp = 12 cm2/s. If the cross-sectional area of the bar is 100 μm2, find the current Ip. Solution Thus, The current Ip can be found from dp(x) Jp = −qDp dx = −qDp d dx p0 e−x/Lp = q D p L p0 e−x/Lp p Jp (0) = q Dp Lp p0 = 1.6 × 10−19 × 12 1 × 10−4 × 1016 = 192 A/cm2 Ip = Jp × A = 192 × 100 × 10−8 = 192 μA EXERCISE 3.5 The linear electron-concentration profile shown in Fig. E3.5 has been established in a piece of silicon. If n0 = 1017/cm3 and W = 1 μm, find the electron-current density in microamperes per micron squared (μA/μm2). If a diffusion current of 1 mA is required, what must the cross-sectional area (in a direction perpendicular to the page) be? Recall that Dn = 35 cm2/s. 148 Chapter 3 Semiconductors n(x) n0 0 Wx Ans. 56 μA/μm2; 18 μm2 Figure E3.5 3.3.3 Relationship between D and μ A simple but powerful relationship ties the diffusion constant with the mobility, Dn μn = Dp μp = VT (3.21) where VT = kT /q. The parameter VT is known as the thermal voltage. At room temperature, T 300 K and VT = 25.9 mV. We will encounter VT repeatedly throughout this book. The relationship in Eq. (3.21) is known as the Einstein relationship. EXERCISE 3.6 Use the Einstein relationship to find Dn and Dp for intrinsic silicon using μn = 1350 cm2/V · s and μp = 480 cm2/V · s. Ans. 35 cm2/s; 12.4 cm2/s 3.4 The pn Junction Having learned important semiconductor concepts, we are now ready to consider our first practical semiconductor structure—the pn junction. As mentioned previously, the pn junction implements the diode (Chapter 4) and plays the dominant role in the structure and operation of the bipolar junction transistor (BJT, Chapter 6). As well, understanding pn junctions is very important to the study of the MOSFET operation (Chapter 5). Metal contact Metal contact 3.4 The pn Junction 149 Anode p-type silicon n-type silicon Cathode Figure 3.8 Simplified physical structure of the pn junction. (Actual geometries are given in Appendix A.) As the pn junction implements the junction diode, its terminals are labeled anode and cathode. 3.4.1 Physical Structure Figure 3.8 shows a simplified physical structure of the pn junction. It consists of a p-type semiconductor (e.g., silicon) brought into close contact with an n-type semiconductor material (also silicon). In actual practice, both the p and n regions are part of the same silicon crystal; that is, the pn junction is formed within a single silicon crystal by creating regions of different dopings (p and n regions). Appendix A provides a description of the fabrication process of integrated circuits including pn junctions. As indicated in Fig. 3.8, external wire connections are made to the p and n regions through metal (aluminum) contacts. If the pn junction is used as a diode, these constitute the diode terminals and are therefore labeled “anode” and “cathode” in keeping with diode terminology.3 3.4.2 Operation with Open-Circuit Terminals Figure 3.9 shows a pn junction under open-circuit conditions—that is, the external terminals are left open. The “+” signs in the p-type material denote the majority holes. The charge of these holes is neutralized by an equal amount of bound negative charge associated with the acceptor atoms. For simplicity, these bound charges are not shown in the diagram. Also not shown are the minority electrons generated in the p-type material by thermal ionization. In the n-type material the majority electrons are indicated by “–” signs. Here also, the bound positive charge, which neutralizes the charge of the majority electrons, is not shown in order to keep the diagram simple. The n-type material also contains minority holes generated by thermal ionization but not shown in the diagram. The Diffusion Current ID Because the concentration of holes is high in the p region and low in the n region, holes diffuse across the junction from the p side to the n side. Similarly, electrons diffuse across the junction from the n side to the p side. These two current components add together to form the diffusion current ID, whose direction is from the p side to the n side, as indicated in Fig. 3.9. The Depletion Region The holes that diffuse across the junction into the n region quickly recombine with some of the majority electrons present there and thus disappear from the scene. This recombination process results also in the disappearance of some free electrons from the 3This terminology in fact is a carryover from that used with vacuum-tube technology, which was the technology for making diodes and other electronic devices until the invention of the transistor in 1947. This event ushered in the era of solid-state electronics, which changed not only electronics, communications, and computers but indeed the world! 150 Chapter 3 Semiconductors E (b) Figure 3.9 (a) The pn junction with no applied voltage (open-circuited terminals). (b) The potential distribution along an axis perpendicular to the junction. n-type material. Thus some of the bound positive charge will no longer be neutralized by free electrons, and this charge is said to have been uncovered. Since recombination takes place close to the junction, there will be a region close to the junction that is depleted of free electrons and contains uncovered bound positive charge, as indicated in Fig. 3.9. The electrons that diffuse across the junction into the p region quickly recombine with some of the majority holes there, and thus disappear from the scene. This results also in the disappearance of some majority holes, causing some of the bound negative charge to be uncovered (i.e., no longer neutralized by holes). Thus, in the p material close to the junction, there will be a region depleted of holes and containing uncovered bound negative charge, as indicated in Fig. 3.9. From the above it follows that a carrier-depletion region will exist on both sides of the junction, with the n side of this region positively charged and the p side negatively charged. This carrier-depletion region—or, simply, depletion region—is also called the space-charge region. The charges on both sides of the depletion region cause an electric field E to be established across the region in the direction indicated in Fig. 3.9. Hence a potential difference results across the depletion region, with the n side at a positive voltage relative to the p side, as shown in Fig. 3.9(b). Thus the resulting electric field opposes the diffusion of holes into the n region and electrons into the p region. In fact, the voltage drop across the depletion region acts as a barrier that has to be overcome for holes to diffuse into the n region and electrons to diffuse into the p region. The larger the barrier voltage, the smaller the number of carriers that will be able to overcome the barrier, and hence the lower the magnitude of diffusion current. Thus it is the appearance of the barrier voltage V0 that limits the carrier diffusion process. It follows that the diffusion current ID depends strongly on the voltage drop V0 across the depletion region. 3.4 The pn Junction 151 The Drift Current IS and Equilibrium In addition to the current component ID due to majority-carrier diffusion, a component due to minority-carrier drift exists across the junction. Specifically, some of the thermally generated holes in the n material move toward the junction and reach the edge of the depletion region. There, they experience the electric field in the depletion region, which sweeps them across that region into the p side. Similarly, some of the minority thermally generated electrons in the p material move to the edge of the depletion region and get swept by the electric field in the depletion region across that region into the n side. These two current components—electrons moved by drift from p to n and holes moved by drift from n to p—add together to form the drift current IS, whose direction is from the n side to the p side of the junction, as indicated in Fig. 3.9. Since the current IS is carried by thermally generated minority carriers, its value is strongly dependent on temperature; however, it is independent of the value of the depletion-layer voltage V0. This is due to the fact that the drift current is determined by the number of minority carriers that make it to the edge of the depletion region; any minority carriers that manage to get to the edge of the depletion region will be swept across by E irrespective of the value of E or, correspondingly, of V0. Under open-circuit conditions (Fig. 3.9) no external current exists; thus the two opposite currents across the junction must be equal in magnitude: ID = IS This equilibrium condition4 is maintained by the barrier voltage V0. Thus, if for some reason ID exceeds IS, then more bound charge will be uncovered on both sides of the junction, the depletion layer will widen, and the voltage across it (V0) will increase. This in turn causes ID to decrease until equilibrium is achieved with ID = IS. On the other hand, if IS exceeds ID, then the amount of uncovered charge will decrease, the depletion layer will narrow, and the voltage across it (V0) will decrease. This causes ID to increase until equilibrium is achieved with ID = IS. The Junction Built-in Voltage With no external voltage applied, the barrier voltage V0 across the pn junction can be shown to be given by5 V0 = VT ln NAND ni2 (3.22) where NA and ND are the doping concentrations of the p side and n side of the junction, respectively. Thus V0 depends both on doping concentrations and on temperature. It is known as the junction built-in voltage. Typically, for silicon at room temperature, V0 is in the range of 0.6 V to 0.9 V. When the pn junction terminals are left open-circuited, the voltage measured between them will be zero. That is, the voltage V0 across the depletion region does not appear between the junction terminals. This is because of the contact voltages existing at the metal–semiconductor junctions at the terminals, which counter and exactly balance the barrier voltage. If this were not the case, we would have been able to draw energy from the isolated pn junction, which would clearly violate the principle of conservation of energy. Width of and Charge Stored in the Depletion Region Figure 3.10 provides further illustration of the situation that obtains in the pn junction when the junction is in equilibrium. 4In fact, in equilibrum the equality of drift and diffusion currents applies not just to the total currents but also to their individual components. That is, the hole drift current must equal the hole diffusion current and, similarly, the electron drift current must equal the electron diffusion current. 5The derivation of this formula and of a number of others in this chapter can be found in textbooks dealing with devices, such as that by Streetman and Bannerjee (see the reading list in Appendix I). 152 Chapter 3 Semiconductors ID IS ϩϩϩϩ ϩϩϩϩ ؊ ؊ ؉؉ ϪϪϪϪ ϪϪϪϪ ϩ ϩ ϩ ϩ p ϩ ϩ ϩ ϩ ؊ ؊ ؉ ؉ Ϫ ϪnϪ Ϫ ϩϩϩϩ ؊ ϪϪϪϪ ؉؉ ϩϩϩϩ ؊ ϪϪϪϪ Ϫxp 0 xn E (a) pp ϭ NA nn ϭ ND Carrier concentration np0 ϭ ni2 NA W Ϫxp 0 xn (b) pn0 ϭ ni2 ND x Charge density Ϫxp ͦ QϪ ͦ ϭ Aq NAxp W (c) ͦ Qϩ ͦ ϭ Aq NDxn xn x Voltage VO Ϫxp 0 xn (d) Figure 3.10 (a) A pn junction with the terminals open-circuited. (b) Carrier concentrations; note that NA > ND. (c) The charge stored in both sides of the depletion region; QJ = Q+ = Q− . (d) The built-in voltage V0. 3.4 The pn Junction 153 In Fig. 3.10(a) we show a junction in which NA > ND, a typical situation in practice. This is borne out by the carrier concentration on both sides of the junction, as shown in Fig. 3.10(b). Note that we have denoted the minority-carrier concentrations in both sides by np0 and pn0, with the additional subscript “0” signifying equilibrium (i.e., before external voltages are applied, as will be seen in the next section). Observe that the depletion region extends in both the p and n materials and that equal amounts of charge exist on both sides (Q+ and Q− in Fig. 3.10c). However, since usually unequal dopings NA and ND are used, as in the case illustrated in Fig. 3.10, the width of the depletion layer will not be the same on the two sides. Rather, to uncover the same amount of charge, the depletion layer will extend deeper into the more lightly doped material. Specifically, if we denote the width of the depletion region in the p side by xp and in the n side by xn, we can express the magnitude of the charge on the n side of the junction as Q+ = qAxnND (3.23) and that on the p side of the junction as Q− = qAxpNA (3.24) where A is the cross-sectional area of the junction in the plane perpendicular to the page. The charge equality condition can now be written as qAxnND = qAxpNA which can be rearranged to yield xn = NA xp ND (3.25) In actual practice, it is usual for one side of the junction to be much more heavily doped than the other, with the result that the depletion region exists almost entirely on one side (the lightly doped side). The width W of the depletion layer can be shown to be given by W = xn + xp = 2es q 11 + NA ND V0 (3.26) where es is the electrical permittivity of silicon = 11.7e0 = 11.7 × 8.85 × 10−14 F/cm = 1.04 × 10−12 F/cm. Typically W is in the range 0.1 μm to 1 μm. Eqs. (3.25) and (3.26) can be used to obtain xn and xp in terms of W as xn = W NA NA + ND (3.27) xp = W ND NA + ND (3.28) The charge stored on either side of the depletion region can be expressed in terms of W by utilizing Eqs. (3.23) and (3.27) to obtain QJ = Q+ = Q− QJ = Aq NAND NA + ND W (3.29) Finally, we can substitute for W from Eq. (3.26) to obtain QJ = A 2esq NAND NA + ND V0 These expressions for QJ will prove useful in subsequent sections. (3.30) 154 Chapter 3 Semiconductors Example 3.5 Consider a pn junction in equilibrium at room temperature (T = 300 K) for which the dop- ing concentrations are NA = 1018/cm3 and ND = 1016/cm3 and the cross-sectional area A = 10−4 cm2. Calculate pp, np0, nn, pn0, V0, W, xn, xp, and QJ . Use ni = 1.5 × 1010/cm3. Solution pp NA = 1018 cm−3 np0 = ni2 pp ni2 NA = (1.5 × 1010) 2 1018 = 2.25 × 102 cm−3 nn pn0 = ni2 nn ND = 1016 cm−3 ni2 ND = (1.5 × 1010)2 1016 = 2.25 × 104 cm−3 To find V0 we use Eq. (3.22), VO = VT ln NAND ni2 where VT = kT q = 8.62 × 10−5 × 300 (eV) q (e) = 25.9 × 10−3 V Thus, V0 = 25.9 × 10−3 ln 1018 × 1016 2.25 × 1020 = 0.814 V To determine W we use Eq. (3.26): 2 × 1.04 × 10−12 1 1 W= 1.6 × 10−19 1018 + 1016 = 3.27 × 10−5 cm = 0.327 μm × 0.814 To determine xn and xp we use Eqs. (3.27) and (3.28), respectively: xn = W NA NA + ND = 0.327 1018 1018 + 1016 = 0.324 μm xp = W ND NA + ND = 0.327 1016 1018 + 1016 = 0.003 μm Finally, to determine the charge stored on either side of the depletion region, we use Eq. (3.29): QJ = 10−4 × 1.6 × 10−19 1018 × 1016 1018 + 1016 = 5.18 × 10−12 C = 5.18 pC × 0.327 × 10−4 3.5 The pn Junction with an Applied Voltage 155 EXERCISES 3.7 Show that 1q V0 = 2 es NAND NA + ND W2 3.8 Show that for a pn junction in which the p side is much more heavily doped than the n side (i.e., NA ND), referred to as a p+n diode, Eqs. (3.26), (3.27), (3.28), (3.29), and (3.30) can be simplified as follows: W 2es qND V0 xn W xp W / NA/ND QJ AqNDW (3.26 ) (3.27 ) (3.28 ) (3.29 ) QJ A 2esqN DV0 (3.30 ) 3.9 If in the fabrication of the pn junction in Example 3.5, it is required to increase the minority-carrier concentration in the n region by a factor of 2, what must be done? Ans. Lower ND by a factor of 2. 3.5 The pn Junction with an Applied Voltage Having studied the open-circuited pn junction in detail, we are now ready to apply a dc voltage between its two terminals to find its electrical conduction properties. If the voltage is applied so that the p side is made more positive than the n side, it is referred to as a forward-bias6 voltage. Conversely, if our applied dc voltage is such that it makes the n side more positive than the p side, it is said to be a reverse-bias voltage. As will be seen, the pn junction exhibits vastly different conduction properties in its forward and reverse directions. Our plan is as follows. We begin by a simple qualitative description in Section 3.5.1 and then consider an analytical description of the i–v characteristic of the junction in Section 3.5.2. 3.5.1 Qualitative Description of Junction Operation Figure 3.11 shows the pn junction under three different conditions: (a) the open-circuit or equilibrium condition studied in the previous section; (b) the reverse-bias condition, where a dc voltage VR is applied; and (c) the forward-bias condition, where a dc voltage VF is applied. 6For the time being, we take the term bias to refer simply to the application of a dc voltage. We will see in later chapters that it has a deeper meaning in the design of electronic circuits. 156 Chapter 3 Semiconductors ID IS p n VR ID IS p n VF ID IS p n V0 (V0 ϩ VR) (a) Open-circuit (equilibrium) (b) Reverse bias Figure 3.11 The pn junction in: (a) equilibrium; (b) reverse bias; (c) forward bias. (V0 Ϫ VF) (c) Forward bias 3.5 The pn Junction with an Applied Voltage 157 Observe that in the open-circuit case, a barrier voltage V0 develops, making n more positive than p, and limiting the diffusion current ID to a value exactly equal to the drift current IS, thus resulting in a zero current at the junction terminals, as should be the case, since the terminals are open-circuited. Also, as mentioned previously, the barrier voltage V0, though it establishes the current equilibrium across the junction, does not in fact appear between the junction terminals. Consider now the reverse-bias case in (b). The externally applied reverse-bias voltage VR is in the direction to add to the barrier voltage, and it does, thus increasing the effective barrier voltage to (V0 + VR) as shown. This reduces the number of holes that diffuse into the n region and the number of electrons that diffuse into the p region. The end result is that the diffusion current ID is dramatically reduced. As will be seen shortly, a reverse-bias voltage of a volt or so is sufficient to cause ID 0, and the current across the junction and through the external circuit will be equal to IS. Recalling that IS is the current due to the drift across the depletion region of the thermally generated minority carriers, we expect IS to be very small and to be strongly dependent on temperature. We will show this to be the case very shortly. We thus conclude that in the reverse direction, the pn junction conducts a very small and almost-constant current equal to IS. Before leaving the reverse-bias case, observe that the increase in barrier voltage will be accompanied by a corresponding increase in the stored uncovered charge on both sides of the depletion region. This in turn means a wider depletion region, needed to uncover the additional charge required to support the larger barrier voltage (V0 + VR). Analytically, these results can be obtained easily by a simple extension of the results of the equilibrium case. Thus the width of the depletion region can be obtained by replacing V0 in Eq. (3.26) by (V0 + VR), W = xn + xp = 2es q 1+1 NA ND (V0 + VR) (3.31) and the magnitude of the charge stored on either side of the depletion region can be determined by replacing V0 in Eq. (3.30) by (V0 + VR), QJ = A 2esq NAND NA + ND (V0 + VR) (3.32) We next consider the forward-bias case shown in Fig. 3.11(c). Here the applied voltage VF is in the direction that subtracts from the built-in voltage V0, resulting in a reduced barrier voltage (V0 − VF) across the depletion region. This reduced barrier voltage will be accompanied by reduced depletion-region charge and correspondingly narrower depletion-region width W. Most importantly, the lowering of the barrier voltage will enable more holes to diffuse from p to n and more electrons to diffuse from n to p. Thus the diffusion current ID increases substantially and, as will be seen shortly, can become many orders of magnitude larger than the drift current IS. The current I in the external circuit is of course the difference between ID and IS, I = ID − IS and it flows in the forward direction of the junction, from p to n. We thus conclude that the pn junction can conduct a substantial current in the forward-bias region and that current is mostly a diffusion current whose value is determined by the forward-bias voltage VF. 158 Chapter 3 Semiconductors 3.5.2 The Current–Voltage Relationship of the Junction We are now ready to find an analytical expression that describes the current–voltage relationship of the pn junction. In the following we consider a junction operating with a forward applied voltage V and derive an expression for the current I that flows in the forward direction (from p to n). However, our derivation is general and will be seen to yield the reverse current when the applied voltage V is made negative. From the qualitative description above we know that a forward-bias voltage V subtracts from the built-in voltage V0, thus resulting in a lower barrier voltage (V0 − V ). The lowered barrier in turn makes it possible for a greater number of holes to overcome the barrier and diffuse into the n region. A similar statement can be made about electrons from the n region diffusing into the p region. Let us now consider the holes injected into the n region. The concentration of holes in the n region at the edge of the depletion region will increase considerably. In fact, an important result from device physics shows that the steady-state concentration at the edge of the depletion region will be pn(xn) = pn0eV/VT (3.33) That is, the concentration of the minority holes increases from the equilibrium value of pn0 (see Fig. 3.10) to the much larger value determined by the value of V, given by Eq. (3.33). We describe this situation as follows: The forward-bias voltage V results in an excess concentration of minority holes at x = xn, given by Excess concentration = pn0eV/VT − pn0 = pn0 eV/VT − 1 (3.34) The increase in minority-carrier concentration in Eqs. (3.33) and (3.34) occurs at the edge of the depletion region (x = xn). As the injected holes diffuse into the n material, some will recombine with the majority electrons and disappear. Thus, the excess hole concentration will decay exponentially with distance. As a result, the total hole concentration in the n material will be given by pn(x) = pn0 + (Excess concentration)e−(x−xn )/Lp Substituting for the “Excess concentration” from Eq. (3.34) gives pn(x) = pn0 + pn0 eV/VT − 1 e−(x−xn )/Lp (3.35) The exponential decay is characterized by the constant Lp, which is called the diffusion length of holes in the n material. The smaller the value of Lp, the faster the injected holes will recombine with the majority electrons, resulting in a steeper decay of minority-carrier concentration. Figure 3.12 shows the steady-state minority-carrier concentration profiles on both sides of a pn junction in which NA ND. Let’s stay a little longer with the diffusion of holes into the n region. Note that the shaded region under the exponential represents the excess minority carriers (holes). From our study of diffusion in Section 3.3, we know that the establishment of a carrier concentration profile such as that in Fig. 3.12 is essential to support a steady-state diffusion current. In fact, we can now find the value of the hole–diffusion current density by applying Eq. (3.19), Jp (x) = −qDp dpn(x) dx p region pn, np Depletion region 3.5 The pn Junction with an Applied Voltage 159 pn(xn) n region Excess concentration pn(x) np (Ϫxp) np(x) np0 Ϫxp 0 pn0 xn Thermal equilibrium value x Figure 3.12 Minority-carrier distribution in a forward-biased pn junction. It is assumed that the p region is more heavily doped than the n region; NA ND. Substituting for pn(x) from Eq. (3.35) gives Jp(x) = q Dp Lp pn0 eV/VT − 1 e−(x−xn )/Lp (3.36) As expected, Jp(x) is highest at x = xn, Jp(xn) = q Dp Lp pn0 eV/VT − 1 (3.37) and decays exponentially for x > xn, as the minority holes recombine with the majority electrons. This recombination, however, means that the majority electrons will have to be replenished by a current that injects electrons from the external circuit into the n region of the junction. This latter current component has the same direction as the hole current (because electrons moving from right to left give rise to current in the direction from left to right). It follows that as Jp(x) decreases, the electron current component increases by exactly the same amount, making the total current in the n material constant at the value given by Eq. (3.37). An exactly parallel development can be applied to the electrons that are injected from the n to the p region, resulting in an electron diffusion current given by a simple adaptation of Eq. (3.37), Jn −xp = q Dn Ln np0 eV/VT − 1 (3.38) Now, although the currents in Eqs. (3.37) and (3.38) are found at the two edges of the depletion region, their values do not change in the depletion region. Thus we can drop the location descriptors (xn), −xp , add the two current densities, and multiply by the junction area A to 160 Chapter 3 Semiconductors obtain the total current I as I = A Jp + Jn I = Aq Dp Lp pn0 + Dn Ln np0 eV/VT − 1 Substituting for pn0 = ni2/ND and for np0 = ni2/NA gives I = Aqni2 Dp + Dn LpND LnNA eV/VT − 1 (3.39) From this equation we note that for a negative V (reverse bias) with a magnitude of a few times VT (25.9 mV), the exponential term becomes essentially zero, and the current across the junction becomes negative and constant. From our qualitative description in Section 3.5.1, we know that this current must be IS. Thus, I = IS eV/VT − 1 (3.40) where IS = Aqni2 Dp + Dn LpND LnNA (3.41) Figure 3.13 shows the I–V characteristic of the pn junction (Eq. 3.40). Observe that in the reverse direction the current saturates at a value equal to –IS. For this reason, IS is given the name saturation current. From Eq. (3.41) we see that IS is directly proportional to the cross-sectional area A of the junction. Thus, another name for IS, one we prefer to use in this book, is the junction scale current. Typical values for IS, for junctions of various areas, range from 10−18 A to 10−12 A. Besides being proportional to the junction area A, the expression for IS in Eq. (3.41) indicates that IS is proportional to ni2, which is a very strong function of temperature (see Eq. 3.2). I 0 IS V Figure 3.13 The pn junction I–V characteristic. 3.5 The pn Junction with an Applied Voltage 161 Example 3.6 For the pn junction considered in Example 3.5 for which NA = 1018/cm3, ND = 1016/cm3, A = 10−4cm2, and ni = 1.5 × 1010/cm3, let Lp = 5 μm, Ln = 10 μm, Dp (in the n region) = 10 cm2/V· s, and Dn (in the p region) = 18 cm2/V· s. The pn junction is forward biased and conducting a current I = 0.1 mA. Calculate: (a) IS; (b) the forward-bias voltage V; and (c) the component of the current I due to hole injection and that due to electron injection across the junction. Solution (a) Using Eq. (3.41), we find IS as IS = 10−4 × 1.6 × 10−19 × 1.5 × 1010 2 10 18 × 5 × 10−4 × 1016 + 10 × 10−4 × 1018 = 7.3 × 10−15A (b) In the forward direction, Thus, For I = 0.1 mA, I = IS eV/VT − 1 IS eV/VT I V = VT ln IS V = 25.9 × 10−3 ln 0.1 × 10−3 7.3 × 10−15 = 0.605 V (c) The hole-injection component of I can be found using Eq. (3.37) Ip = Aq Dp Lp pn0 eV/VT −1 = Aq Dp ni2 eV/VT − 1 Lp ND Similarly, In can be found using Eq. (3.39), In = Aq Dn Ln ni2 NA eV/VT − 1 Thus, Ip = Dp Ln NA In Dn Lp ND For our case, Ip In = 10 18 × 10 5 × 1018 1016 = 1.11 × 102 = 111 162 Chapter 3 Semiconductors Example 3.6 continued Thus most of the current is conducted by holes injected into the n region. Specifically, 111 Ip = 112 × 0.1 = 0.0991 mA 1 In = 112 × 0.1 = 0.0009 mA This stands to reason, since the p material has a doping concentration 100 times that of the n material. EXERCISES 3.10 Show that if NA ND, IS Aqni2 Dp LpND 3.11 For the pn junction in Example 3.6, find the value of IS and that of the current I at V = 0.605 V (same voltage found in Example 3.6 at a current I = 0.1 mA) if ND is reduced by a factor of 2. Ans. 1.46 × 10−14 A; 0.2 mA 3.12 For the pn junction considered in Examples 3.5 and 3.6, find the width of the depletion region W corresponding to the forward-bias voltage found in Example 3.6. (Hint: Use the formula in Eq. (3.31) with VR replaced with −VF.) Ans. 0.166 μm 3.13 For the pn junction considered in Examples 3.5 and 3.6, find the width of the depletion region W and the charge stored in the depletion region QJ when a 2-V reverse bias is applied. Also find the value of the reverse current I. Ans. 0.608 μm; 9.63 pC; 7.3 × 10−15 A 3.5.3 Reverse Breakdown The description of the operation of the pn junction in the reverse direction, and the I−V relationship of the junction in Eq. (3.40), indicate that at a reverse-bias voltage –V, with V VT , the reverse current that flows across the junction is approximately equal to IS and thus is very small. However, as the magnitude of the reverse-bias voltage V is increased, a value is reached at which a very large reverse current flows as shown in Fig. 3.14. Observe that as V reaches the value VZ, the dramatic increase in reverse current is accompanied by a very small increase in the reverse voltage; that is, the reverse voltage across the junction 3.5 The pn Junction with an Applied Voltage 163 I ϪVZ 0 V Figure 3.14 The I–V characteristic of the pn junction showing the rapid increase in reverse current in the breakdown region. remains very close to the value VZ. The phenomenon that occurs at V = VZ is known as junction breakdown. It is not a destructive phenomenon. That is, the pn junction can be repeatedly operated in the breakdown region without a permanent effect on its characteristics. This, however, is predicated on the assumption that the magnitude of the reverse-breakdown current is limited by the external circuit to a “safe” value. The “safe” value is one that results in the limitation of the power dissipated in the junction to a safe, allowable level. There are two possible mechanisms for pn junction breakdown: the zener effect7 and the avalanche effect. If a pn junction breaks down with a breakdown voltage VZ < 5 V, the breakdown mechanism is usually the zener effect. Avalanche breakdown occurs when VZ is greater than approximately 7 V. For junctions that break down between 5 V and 7 V, the breakdown mechanism can be either the zener or the avalanche effect or a combination of the two. Zener breakdown occurs when the electric field in the depletion layer increases to the point of breaking covalent bonds and generating electron–hole pairs. The electrons generated in this way will be swept by the electric field into the n side and the holes into the p side. Thus these electrons and holes constitute a reverse current across the junction. Once the zener effect starts, a large number of carriers can be generated, with a negligible increase in the junction voltage. Thus the reverse current in the breakdown region will be large and its value must be determined by the external circuit, while the reverse voltage appearing between the diode terminals will remain close to the specified breakdown voltage VZ. The other breakdown mechanism, avalanche breakdown, occurs when the minority carriers that cross the depletion region under the influence of the electric field gain sufficient kinetic energy to be able to break covalent bonds in atoms with which they collide. The carriers liberated by this process may have sufficiently high energy to be able to cause other carriers to be liberated in another ionizing collision. This process keeps repeating in the fashion of an avalanche, with the result that many carriers are created that are able to support any value of 7Named after an early worker in the area. Note that the subscript Z in VZ denotes zener. We will use VZ to denote the breakdown voltage whether the breakdown mechanism is the zener effect or the avalanche effect. 164 Chapter 3 Semiconductors reverse current, as determined by the external circuit, with a negligible change in the voltage drop across the junction. As will be seen in Chapter 4, some pn junction diodes are fabricated to operate specifically in the breakdown region, where use is made of the nearly constant voltage VZ. 3.6 Capacitive Effects in the pn Junction There are two charge-storage mechanisms in the pn junction. One is associated with the charge stored in the depletion region, and the other is associated with the minority-carrier charge stored in the n and p materials as a result of the concentration profiles established by carrier injection. While the first is easier to see when the pn junction is reverse biased, the second is in effect only when the junction is forward biased. 3.6.1 Depletion or Junction Capacitance When a pn junction is reverse biased with a voltage VR, the charge stored on either side of the depletion region is given by Eq. (3.32), QJ = A 2es q NAND NA + ND (V0 + VR ) Thus, for a given pn junction, where α is given by QJ = α V0 + VR (3.42) α=A 2es q NAND NA + ND (3.43) Thus QJ is nonlinearly related to VR, as shown in Fig. 3.15. This nonlinear relationship makes it difficult to define a capacitance that accounts for the need to change QJ whenever VR is Charge stored in depletion layer, QJ Slope ϭ CJ Q Bias point Figure 3.15 The charge stored on either side of the depletion layer as a 0 VQ Reverse voltage,VR function of the reverse voltage VR. 3.6 Capacitive Effects in the pn Junction 165 changed. We can, however, assume that the junction is operating at a point such as Q, as indicated in Fig. 3.15, and define a capacitance Cj that relates the change in the charge QJ to a change in the voltage VR, Cj = dQJ dVR VR =VQ (3.44) This incremental-capacitance approach turns out to be quite useful in electronic circuit design, as we shall see throughout this book. Using Eq. (3.44) together with Eq. (3.42) yields Cj = 2 α V0 + VR (3.45) The value of Cj at zero reverse bias can be obtained from Eq. (3.45) as Cj0 = α 2 V0 (3.46) which enables us to express Cj as Cj = Cj0 1 + VR V0 (3.47) where Cj0 is given by Eq. (3.46) or alternatively if we substitute for α from Eq. (3.43) by Cj0 = A esq NAND 2 NA + ND 1 V0 (3.48) Before leaving the subject of depletion-region or junction capacitance we point out that in the pn junction we have been studying, the doping concentration is made to change abruptly at the junction boundary. Such a junction is known as an abrupt junction. There is another type of pn junction in which the carrier concentration is made to change gradually from one side of the junction to the other. To allow for such a graded junction, the formula for the junction capacitance (Eq. 3.47) can be written in the more general form Cj = Cj0 1 + VR m V0 (3.49) where m is a constant called the grading coefficient, whose value ranges from 1/3 to 1/2 depending on the manner in which the concentration changes from the p to the n side. 166 Chapter 3 Semiconductors EXERCISE 3.14 For the pn junction considered in Examples 3.5 and 3.6, find Cj0 and Cj at VR = 2 V. Recall that V0 = 0.814 V, NA = 1018/cm3, ND = 1016/cm3, and A = 10−4 cm2. Ans. 3.2 pF; 1.7 pF 3.6.2 Diffusion Capacitance Consider a forward-biased pn junction. In steady state, minority-carrier distributions in the p and n materials are established, as shown in Fig. 3.12. Thus a certain amount of excess minority-carrier charge is stored in each of the p and n bulk regions (outside the depletion region). If the terminal voltage V changes, this charge will have to change before a new steady state is achieved. This charge-storage phenomenon gives rise to another capacitive effect, distinctly different from that due to charge storage in the depletion region. To calculate the excess minority-carrier charge, refer to Fig. 3.12. The excess hole charge stored in the n region can be found from the shaded area under the exponential as follows:8 Qp = Aq × shaded area under the pn(x)curve = Aq[pn(xn) − pn0]Lp Substituting for pn(xn) from Eq. (3.33) and using Eq. (3.37) enables us to express Qp as Qp = Lp2 Dp Ip (3.50) The factor Lp2/Dp that relates Qp to Ip is a useful device parameter that has the dimension of time (s) and is denoted τp Thus, τp = Lp2 Dp (3.51) Qp = τpIp (3.52) The time constant τp is known as the excess minority-carrier (hole) lifetime. It is the average time it takes for a hole injected into the n region to recombine with a majority electron. This definition of τp implies that the entire charge Qp disappears and has to be replenished every τp seconds. The current that accomplishes the replenishing is Ip = Qp/τp. This is an alternate derivation for Eq. (3.52). 8Recall that the area under an exponential curve Ae−x/B is equal to AB. 3.6 Capacitive Effects in the pn Junction 167 A relationship similar to that in Eq. (3.52) can be developed for the electron charge stored in the p region, Qn = τnIn (3.53) where τn is the electron lifetime in the p region. The total excess minority-carrier charge can be obtained by adding together Qp and Qn, Q = τpIp + τnIn (3.54) This charge can be expressed in terms of the diode current I = Ip + In as Q = τT I (3.55) where τT is called the mean transit time of the junction. Obviously, τT is related to τp and τn. Furthermore, for most practical devices, one side of the junction is much more heavily doped than the other. For instance, if NA ND, one can show that Ip In, I Ip, Qp Qn, Q Qp, and thus τT τp. For small changes around a bias point, we can define an incremental diffusion capacitance Cd as dQ Cd = dV (3.56) and can show that Cd = τT VT I (3.57) where I is the forward-bias current. Note that Cd is directly proportional to the forward current I and thus is negligibly small when the diode is reverse biased. Also note that to keep Cd small, the transit time τT must be made small, an important requirement for a pn junction intended for high-speed or high-frequency operation. EXERCISES 3.15 Use the definition of Cd in Eq. (3.56) to derive the expression in Eq. (3.57) by means of Eqs. (3.55) and (3.40). 3.16 For the pn junction considered in Examples 3.5 and 3.6 for which Dp = 10 cm2/V · s, and Lp = 5 μm, find τp and Cd at a forward-bias current of 0.1 mA. Recall that for this junction, Ip I. Ans. 25 ns; 96.5 pF 168 Chapter 3 Semiconductors Summary Today’s microelectronics technology is almost entirely based on the semiconductor material silicon. If a circuit is to be fabricated as a monolithic integrated circuit (IC) it is made using a single silicon crystal, no matter how large the circuit is (a recent chip contains 4.31 billion transistors). In a crystal of intrinsic or pure silicon, the atoms are held in position by covalent bonds. At very low temperatures, all the bonds are intact, and no charge carriers are available to conduct electrical current. Thus, at such low temperatures, silicon behaves as an insulator. At room temperature, thermal energy causes some of the covalent bonds to break, thus generating free electrons and holes that become available for current conduction. Current in semiconductors is carried by free electrons and holes. Their numbers are equal and relatively small in intrinsic silicon. The conductivity of silicon can be increased dramatically by introducing small amounts of appropriate impurity materials into the silicon crystal in a process called doping. There are two kinds of doped semiconductor: n-type, in which electrons are abundant, and p-type, in which holes are abundant. There are two mechanisms for the transport of charge carriers in semiconductors: drift and diffusion. Carrier drift results when an electric field E is applied across a piece of silicon. The electric field accelerates the holes in the direction of E and the electrons in the direction opposite to E. These two current components add together to produce a drift current in the direction of E. Carrier diffusion occurs when the concentration of charge carriers is made higher in one part of the silicon crystal than in other parts. To establish a steady-state diffusion current, a carrier concentration gradient must be maintained in the silicon crystal. A basic semiconductor structure is the pn junction. It is fabricated in a silicon crystal by creating a p region in close proximity to an n region. The pn junction is a diode and plays a dominant role in the structure and operation of transistors. When the terminals of the pn junction are left open, no current flows externally. However, two equal and opposite currents, ID and IS, flow across the junction, and equilibrium is maintained by a built-in voltage V0 that develops across the junction, with the n side positive relative to the p side. Note, however, that the voltage across an open junction is 0 V, since V0 is canceled by potentials appearing at the metal-to-semiconductor connection interfaces. The voltage V0 appears across the depletion region, which extends on both sides of the junction. The diffusion current ID is carried by holes diffusing from p to n and electrons diffusing from n to p. ID flows from p to n, which is the forward direction of the junction. Its value depends on V0. The drift current IS is carried by thermally generated minority electrons in the p material that are swept across the depletion layer into the n side, and by thermally generated minority holes in the n side that are swept across the depletion region into the p side. IS flows from n to p, in the reverse direction of the junction, and its value is a strong function of temperature but independent of V0. Forward biasing the pn junction, that is, applying an external voltage V that makes p more positive than n, reduces the barrier voltage to V0 − V and results in an exponential increase in ID while IS remains unchanged. The net result is a substantial current I = ID − IS that flows across the junction and through the external circuit. Applying a negative V reverse biases the junction and increases the barrier voltage, with the result that ID is reduced to almost zero and the net current across the junction becomes the very small reverse current IS. If the reverse voltage is increased in magnitude to a value VZ specific to the particular junction, the junction breaks down, and a large reverse current flows. The value of the reverse current must be limited by the external circuit. Whenever the voltage across a pn junction is changed, some time has to pass before steady state is reached. This is due to the charge-storage effects in the junction, which are modeled by two capacitances: the junction capacitance Cj and the diffusion capacitance Cd. For future reference, we present in Table 3.1 a summary of pertinent relationships and the values of physical constants. Table 3.1 Summary of Important Equations Quantity Carrier concentration in intrinsic silicon (cm−3) Relationship ni = BT 3/2e−Eg /2kT Diffusion current density (A/cm2) Drift current density (A/cm2) Resistivity ( · cm) Relationship between mobility and diffusivity Carrier concentration in n-type silicon (cm−3) Carrier concentration in p-type silicon (cm−3) Junction built-in voltage (V) Width of depletion region (cm) dp Jp = −qDp dx Jn = qDn dn dx Jdrift = q pμp + nμn E ρ = 1/ q pμp + nμn Dn μn = Dp μp = VT nn0 ND pn0 = ni2/ND pp0 NA np0 = ni2/NA V0 = VT ln NA ND ni2 xn = NA xp ND W = xn + xp = 2es 1 + 1 q NA ND V0 + VR Summary 169 Values of Constants and Parameters (for Intrinsic Si at T = 300 K) B = 7.3 × 1015 cm−3K−3/2 Eg = 1.12 eV k = 8.62 × 10−5 eV/K ni = 1.5 × 1010/cm3 q = 1.60 × 10−19 coulomb Dp = 12 cm2/s Dn = 34 cm2/s μp = 480 cm2/V · s μn = 1350 cm2/V · s μp and μn decrease with the increase in doping concentration VT = kT /q 25.9 mV es = 11.7e0 e0 = 8.854 × 10−14 F/cm 170 Chapter 3 Semiconductors Table 3.1 continued Quantity Charge stored in depletion layer (coulomb) Forward current (A) Saturation current (A) I–V relationship Minority-carrier lifetime (s) Minority-carrier charge storage (coulomb) Relationship QJ = q NA ND NA + ND AW I = Ip + In Ip = Aqni2 Dp Lp ND eV/VT − 1 In = Aqni2 Dn Ln NA eV/VT − 1 IS = Aqni2 Dp + Dn LpND LnNA I = IS eV/VT − 1 τp = Lp2/Dp τn = Ln2/Dn Qp = τpIp Qn = τnIn Q = Qp + Qn = τT I Depletion capacitance (F) Diffusion capacitance (F) Cj0 = A es q 2 NAND 1 NA + N D V0 Cj = Cj0 1 + VR m V0 Cd = τT VT I Values of Constants and Parameters (for Intrinsic Si at T = 300 K) Lp, Ln = 1 μm to 100 μm τp, τn = 1 ns to 104 ns m = 1 to 1 32 PROBLEMS If in the following problems the need arises for the values of particular parameters or physical constants that are not stated, please consult Table 3.1. Section 3.1: Intrinsic Semiconductors 3.1 Find values of the intrinsic carrier concentration ni for silicon at −55°C, 0°C, 20°C, 75°C, and 125°C. At each temperature, what fraction of the atoms is ionized? Recall that a silicon crystal has approximately 5 × 1022 atoms/cm3. 3.2 Calculate the value of ni for gallium arsenide (GaAs) at T = 300 K. The constant B = 3.56 × 1014 cm−3K−3/2 and the bandgap voltage Eg = 1.42 eV. of 3 V is imposed. Let μn = 1350 cm2/V · s and μp = 480 cm2/V · s· 3.8 Find the current that flows in a silicon bar of 10-μm length having a 5-μm × 4-μm cross-section and having free-electron and hole densities of 104/cm3 and 1016/cm3, respectively, when a 1 V is applied end-to-end. Use μn = 1200 cm2/V · s and μp = 500 cm2/V · s. 3.9 In a 10-μm-long bar of donor-doped silicon, what donor concentration is needed to realize a current density of 2 mA/μm2 in response to an applied voltage of 1 V? (Note: Although the carrier mobilities change with doping concentration, as a first approximation you may assume μn to be constant and use 1350 cm2/V · s, the value for intrinsic silicon.) Section 3.2: Doped Semiconductors 3.3 For a p-type silicon in which the dopant concentration NA = 5 × 1018/cm3, find the hole and electron concentrations at T = 300 K. 3.4 For a silicon crystal doped with phosphorus, what must ND be if at T = 300 K the hole concentration drops below the intrinsic level by a factor of 108? 3.5 In a phosphorus-doped silicon layer with impurity concentration of 1017/cm3, find the hole and electron concentrations at 27°C and 125°C. 3.10 Holes are being steadily injected into a region of n-type silicon (connected to other devices, the details of which are not important for this question). In the steady state, the excess-hole concentration profile shown in Fig. P3.10 is established in the n-type silicon region. Here “excess” means over and above the thermal-equilibrium concentration (in the absence of hole injection), denoted pn0. If ND = 1016/cm3, ni = 1.5 × 1010/cm3, Dp = 12 cm2/s, and W = 50 nm, find the density of the current that will flow in the x direction. Section 3.3: Current Flow in Semiconductors 3.6 A young designer, aiming to develop intuition concerning conducting paths within an integrated circuit, examines the end-to-end resistance of a connecting bar 10-μm long, 3-μm wide, and 1 μm thick, made of various materials. The designer considers: 108 pn0 pn(x) pn0 n region (a) intrinsic silicon (b) n-doped silicon with ND = 5 × 1016/cm3 (c) n-doped silicon with ND = 5 × 1018/cm3 (d) p-doped silicon with NA = 5 × 1016/cm3 (e) aluminum with resistivity of 2.8 μ · cm 0 Figure P3.10 W x Find the resistance in each case. For intrinsic silicon, use the data in Table 3.1. For doped silicon, assume μn = 3μp = 1200 cm2/V · s. (Recall that R = ρL/A.) 3.7 Contrast the electron and hole drift velocities through a 10-μm layer of intrinsic silicon across which a voltage 3.11 Both the carrier mobility and the diffusivity decrease as the doping concentration of silicon is increased. Table P3.11 provides a few data points for μn and μp versus doping concentration. Use the Einstein relationship to obtain the corresponding values for Dn and Dp. = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 3 PROBLEMS 172 Chapter 3 Semiconductors Table P3.11 Doping Concentration (carriers/cm3) Intrinsic 1016 1017 1018 μn (cm2/V · s) 1350 1200 750 380 μp (cm2/V · s) 480 400 260 160 Dn (cm2/s) Dp (cm2/s) Section 3.4: The pn Junction 3.12 Calculate the built-in voltage of a junction in which the p and n regions are doped equally with 5 × 1016 atoms/cm3. Assume ni = 1.5 × 1010/cm3. With the terminals left open, what is the width of the depletion region, and how far does it extend into the p and n regions? If the cross-sectional area of the junction is 20 μm2, find the magnitude of the charge stored on either side of the junction. 3.13 If, for a particular junction, the acceptor concentration is 1017/cm3 and the donor concentration is 1016/cm3, find the junction built-in voltage. Assume ni = 1.5 × 1010/cm3. Also, find the width of the depletion region (W) and its extent in each of the p and n regions when the junction terminals are left open. Calculate the magnitude of the charge stored on either side of the junction. Assume that the junction area is 100 μm2. 3.14 Estimate the total charge stored in a 0.1-μm depletion layer on one side of a 10-μm × 10-μm junction. The doping concentration on that side of the junction is 1018/cm3. 3.15 In a pn junction for which NA ND, and the depletion layer exists mostly on the shallowly doped side with W = 0.2 μm, find V0 if ND = 1016/cm3. Also calculate QJ for the case A = 10 μm2. 3.16 By how much does V0 change if NA or ND is increased by a factor of 10? Section 3.5: The pn Junction with an Applied Voltage 3.17 If a 3-V reverse-bias voltage is applied across the junction specified in Problem 3.13, find W and QJ . 3.18 Show that for a pn junction reverse-biased with a voltage VR, the depletion-layer width W and the charge stored on either side of the junction, QJ , can be expressed as W = W0 1 + VR V0 QJ = QJ0 1 + VR V0 where W0 and QJ0 are the values in equilibrium. 3.19 In a forward-biased pn junction show that the ratio of the current component due to hole injection across the junction to the component due to electron injection is given by Ip = Dp Ln NA In Dn Lp ND Evaluate this ratio for the case NA = 1018/cm3, ND = 1016/cm3, Lp = 5 μm, Ln = 10 μm, Dp = 10 cm2/s, and Dn = 20 cm2/s, and hence find Ip and In for the case in which the pn junction is conducting a forward current I = 100 μA. 3.20 Calculate IS and the current I for V = 750 mV for a pn junction for which NA = 1017/cm3, ND = 1016/cm3, A = 100 μm2, ni = 1.5 × 1010/cm3, Lp = 5 μm, Ln = 10 μm, Dp = 10 cm2/s, and Dn = 18 cm2/s. 3.21 Assuming that the temperature dependence of IS arises mostly because IS is proportional to ni2, use the expression for ni in Eq. (3.2) to determine the factor by which ni2 changes as T changes from 300 K to 305 K. This will be approximately the same factor by which IS changes for a 5°C rise in temperature. What is the factor? = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 3 PROBLEMS Problems 173 3.22 A p+n junction is one in which the doping concentration in the p region is much greater than that in the n region. In such a junction, the forward current is mostly due to hole injection across the junction. Show that I Ip = Aqni2 Dp LpND eV/VT − 1 For the specific case in which ND = 1017/cm3, Dp = 10 cm2/s, Lp = 10 μm, and A = 104 μm2, find IS and the voltage V obtained when I = 1 mA. Assume operation at 300 K where ni = 1.5 × 1010/cm3. 3.23 A pn junction for which the breakdown voltage is 12 V has a rated (i.e., maximum allowable) power dissipation of 0.25 W. What continuous current in the breakdown region will raise the dissipation to half the rated value? If breakdown occurs for only 10 ms in every 20 ms, what average breakdown current is allowed? Section 3.6: Capacitive Effects in the pn Junction 3.24 For the pn junction specified in Problem 3.13, find Cj0 and Cj at VR = 3 V. 3.25 For a particular junction for which Cj0 = 0.4 pF, V0 = 0.75 V, and m = 1/3, find Cj at reverse-bias voltages of 1 V and 10 V. to have at I = 0.1 mA? What is the mean transit time for this junction? 3.28 For the p+n junction specified in Problem 3.22, find τp and calculate the excess minority-carrier charge and the value of the diffusion capacitance at I = 0.1 mA. *3.29 A short-base diode is one where the widths of the p and n regions are much smaller than Ln and Lp, respectively. As a result, the excess minority-carrier distribution in each region is a straight line rather than the exponentials shown in Fig. 3.12. (a) For the short-base diode, sketch a figure corresponding to Fig. 3.12 and assume as in Fig. 3.12 that NA ND. (b) Following a derivation similar to that given in Section 3.5.2, show that if the widths of the p and n regions are denoted Wp and Wn then I = Aqni2 Dp + Dn Wn − xn ND Wp − xp NA eV/VT − 1 and 1 Qp = 2 Wn − xn Dp 2 Ip 1 2 Wn2 Dp Ip , for Wn xn 3.26 The junction capacitance Cj can be thought of as that of a parallel-plate capacitor and thus given by Cj = eA W Show that this approach leads to a formula identical to that obtained by combining Eqs. (3.43) and (3.45) [or equivalently, by combining Eqs. (3.47) and (3.48)]. 3.27 A pn junction operating in the forward-bias region with a current I of 1 mA is found to have a diffusion capacitance of 5 pF. What diffusion capacitance do you expect this junction (c) Also, assuming Q Qp, I Ip, show that Cd = τT VT I where τT = 1 2 Wn2 Dp (d) If a designer wishes to limit Cd to 8 pF at I = 1 mA, what should Wn be? Assume Dp = 10 cm2/s. = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 4 Diodes Introduction 175 4.1 The Ideal Diode 176 4.2 Terminal Characteristics of Junction Diodes 184 4.3 Modeling the Diode Forward Characteristic 190 4.4 Operation in the Reverse Breakdown Region—Zener Diodes 202 4.5 Rectifier Circuits 207 4.6 Limiting and Clamping Circuits 221 4.7 Special Diode Types 227 Summary 229 Problems 230 IN THIS CHAPTER YOU WILL LEARN 1. The characteristics of the ideal diode and how to analyze and design circuits containing multiple ideal diodes together with resistors and dc sources to realize useful and interesting nonlinear functions. 2. The details of the i–v characteristic of the junction diode (which was derived in Chapter 3) and how to use it to analyze diode circuits operating in the various bias regions: forward, reverse, and breakdown. 3. A simple but effective model of the diode i–v characteristic in the forward direction: the constant-voltage-drop model. 4. A powerful technique for the application and modeling of the diode (and in later chapters, transistors): dc-biasing the diode and modeling its operation for small signals around the dc operating point by means of the small-signal model. 5. The use of a string of forward-biased diodes and of diodes operating in the breakdown region (zener diodes), to provide constant dc voltages (voltage regulators). 6. Application of the diode in the design of rectifier circuits, which convert ac voltages to dc as needed for powering electronic equipment. 7. A number of other practical and important applications of diodes. Introduction In Chapters 1 and 2 we dealt almost entirely with linear circuits; any nonlinearity, such as that introduced by amplifier output saturation, was treated as a problem to be solved by the circuit designer. However, there are many other signal-processing functions that can be implemented only by nonlinear circuits. Examples include the generation of dc voltages from the ac power supply, and the generation of signals of various waveforms (e.g., sinusoids, square waves, pulses). Also, digital logic and memory circuits constitute a special class of nonlinear circuits. The simplest and most fundamental nonlinear circuit element is the diode. Just like a resistor, the diode has two terminals; but unlike the resistor, which has a linear (straight-line) relationship between the current flowing through it and the voltage appearing across it, the diode has a nonlinear i–v characteristic. This chapter is concerned with the study of diodes. In order to understand the essence of the diode function, we begin with a fictitious element, the ideal diode. We then introduce the silicon junction diode, explain its terminal characteristics, and provide techniques for the 175 176 Chapter 4 Diodes analysis of diode circuits. The latter task involves the important subject of device modeling. Our study of modeling the diode characteristics will lay the foundation for our study of modeling transistor operation in the next three chapters. Of the many applications of diodes, their use in the design of rectifiers (which convert ac to dc) is the most common. Therefore we shall study rectifier circuits in some detail and briefly look at a number of other diode applications. Further nonlinear circuits that utilize diodes and other devices will be found throughout the book, but particularly in Chapter 18. The junction diode is nothing more than the pn junction we studied in Chapter 3, and most of this chapter is concerned with the study of silicon pn-junction diodes. In the last section, however, we briefly consider some specialized diode types, including the photodiode and the light-emitting diode. 4.1 The Ideal Diode 4.1.1 Current–Voltage Characteristic The ideal diode may be considered to be the most fundamental nonlinear circuit element. It is a two-terminal device having the circuit symbol of Fig. 4.1(a) and the i–v characteristic shown in Fig. 4.1(b). The terminal characteristic of the ideal diode can be interpreted as follows: If a negative voltage (relative to the reference direction indicated in Fig. 4.1a) is applied to the diode, no current flows and the diode behaves as an open circuit (Fig. 4.1c). Diodes operated in this mode are said to be reverse biased, or operated in the reverse direction. An ideal diode has zero current when operated in the reverse direction and is said to be cut off, or simply off. On the other hand, if a positive current (relative to the reference direction indicated in Fig. 4.1(a) is applied to the ideal diode, zero voltage drop appears across the diode. In other words, the ideal diode behaves as a short circuit in the forward direction (Fig. 4.1d); it passes any current with zero voltage drop. A forward-biased diode is said to be turned on, or simply on. From the above description it should be noted that the external circuit must be designed to limit the forward current through a conducting diode, and the reverse voltage across a cutoff diode, to predetermined values. Figure 4.2 shows two diode circuits that illustrate this point. In the circuit of Fig. 4.2(a) the diode is obviously conducting. Thus its voltage drop will be zero, and the current through it will be determined by the +10-V supply and the 1-k resistor as 10 mA. The diode in the circuit of Fig. 4.2(b) is obviously cut off, and thus its current will be zero, which in turn means that the entire 10-V supply will appear as reverse bias across the diode. The positive terminal of the diode is called the anode and the negative terminal the cathode, a carryover from the days of vacuum-tube diodes. The i–v characteristic of the ideal diode (conducting in one direction and not in the other) should explain the choice of its arrow-like circuit symbol. As should be evident from the preceding description, the i–v characteristic of the ideal diode is highly nonlinear; although it consists of two straight-line segments, they are at 90° to one another. A nonlinear curve that consists of straight-line segments is said to be piecewise linear. If a device having a piecewise-linear characteristic is used in a particular application in such a way that the signal across its terminals swings along only one of the linear segments, then the device can be considered a linear circuit element as far as that particular circuit 4.1 The Ideal Diode 177 i ϩ vϪ Figure 4.1 The ideal diode: (a) diode circuit symbol; (b) i–v characteristic; (c) equivalent circuit in the reverse direction; (d) equivalent circuit in the forward direction. Figure 4.2 The two modes of operation of ideal diodes and the use of an external circuit to limit (a) the forward current and (b) the (a) (b) reverse voltage. application is concerned. On the other hand, if signals swing past one or more of the break points in the characteristic, linear analysis is no longer possible. 4.1.2 A Simple Application: The Rectifier A fundamental application of the diode, one that makes use of its severely nonlinear i–v curve, is the rectifier circuit shown in Fig. 4.3(a). The circuit consists of the series connection of a diode D and a resistor R. Let the input voltage vI be the sinusoid shown in Fig. 4.3(b), and assume the diode to be ideal. During the positive half-cycles of the input sinusoid, the positive 178 Chapter 4 Diodes vI will cause current to flow through the diode in its forward direction. It follows that the diode voltage vD will be very small—ideally zero. Thus the circuit will have the equivalent shown in Fig. 4.3(c), and the output voltage vO will be equal to the input voltage vI. On the other hand, during the negative half-cycles of vI, the diode will not conduct. Thus the circuit will have the equivalent shown in Fig. 4.3(d), and vO will be zero. Thus the output voltage will have the waveform shown in Fig. 4.3(e). Note that while vI alternates in polarity and has a zero average value, vO is unidirectional and has a finite average value or a dc component. Thus the circuit of Fig. 4.3(a) rectifies the signal and hence is called a rectifier. It can be used to generate dc from ac. We will study rectifier circuits in Section 4.5. D (a) (b) vI Ն 0 (c) (e) vI Յ 0 (d) Figure 4.3 (a) Rectifier circuit. (b) Input waveform. (c) Equivalent circuit when vI ≥ 0. (d) Equivalent circuit when vI ≤ 0. (e) Output waveform. 4.1 The Ideal Diode 179 EXERCISES 4.1 For the circuit in Fig. 4.3(a), sketch the transfer characteristic vO versus vI . Ans. See Fig. E4.1 Figure E4.1 4.2 For the circuit in Fig. 4.3(a), sketch the waveform of vD. Ans. vD = vI − vO, resulting in the waveform in Fig. E4.2 vD 0 t –Vp Figure E4.2 4.3 In the circuit of Fig. 4.3(a), let vI have a peak value of 10 V and R = 1 k . Find the peak value of iD and the dc component of vO. (Hint: The average value of half-sine waves is Vp/π .) Ans. 10 mA; 3.18 V 180 Chapter 4 Diodes Example 4.1 Figure 4.4(a) shows a circuit for charging a 12-V battery. If vS is a sinusoid with 24-V peak amplitude, find the fraction of each cycle during which the diode conducts. Also, find the peak value of the diode current and the maximum reverse-bias voltage that appears across the diode. (a) (b) Figure 4.4 Circuit and waveforms for Example 4.1. Solution The diode conducts when vS exceeds 12 V, as shown in Fig. 4.4(b). The conduction angle is 2θ , where θ is given by 24 cos θ = 12 Thus θ = 60° and the conduction angle is 120°, or one-third of a cycle. The peak value of the diode current is given by Id = 24 − 12 100 = 0.12 A The maximum reverse voltage across the diode occurs when vS is at its negative peak and is equal to 24 + 12 = 36 V. 4.1.3 Another Application: Diode Logic Gates Diodes together with resistors can be used to implement digital logic functions. Figure 4.5 shows two diode logic gates. To see how these circuits function, consider a positive-logic system in which voltage values close to 0 V correspond to logic 0 (or low) and voltage values close to +5 V correspond to logic 1 (or high). The circuit in Fig. 4.5(a) has three inputs, vA, vB, and vC. It is easy to see that diodes connected to +5-V inputs will conduct, thus clamping the output vY to a value equal to +5 V. This positive voltage at the output will keep the diodes whose inputs are low (around 0 V) cut off. Thus the output will be high if one or more of the inputs are high. The circuit therefore implements the logic OR function, which in Boolean 4.1 The Ideal Diode 181 (a) (b) Figure 4.5 Diode logic gates: (a) OR gate; (b) AND gate (in a positive-logic system). notation is expressed as Y =A+B+C Similarly, the reader is encouraged to show that using the same logic system mentioned above, the circuit of Fig. 4.5(b) implements the logic AND function, Y =A·B·C Example 4.2 Assuming the diodes to be ideal, find the values of I and V in the circuits of Fig. 4.6. D D (a) Figure 4.6 Circuits for Example 4.2. D D (b) 182 Chapter 4 Diodes Example 4.2 continued Solution In these circuits it might not be obvious at first sight whether none, one, or both diodes are conducting. In such a case, we make a plausible assumption, proceed with the analysis, and then check whether we end up with a consistent solution. For the circuit in Fig. 4.6(a), we shall assume that both diodes are conducting. It follows that VB = 0 and V = 0. The current through D2 can now be determined from ID2 = 10 − 10 0 = 1 mA Writing a node equation at B, I + 1 = 0 − (−10) 5 results in I = 1 mA. Thus D1 is conducting as originally assumed, and the final result is I = 1 mA and V = 0 V. For the circuit in Fig. 4.6(b), if we assume that both diodes are conducting, then VB = 0 and V = 0. The current in D2 is obtained from 10 − 0 ID2 = 5 = 2 mA The node equation at B is I + 2 = 0 − (−10) 10 which yields I = −1 mA. Since this is not possible, our original assumption is not correct. We start again, assuming that D1 is off and D2 is on. The current ID2 is given by ID2 = 10 − (−10) 15 = 1.33 mA and the voltage at node B is VB = −10 + 10 × 1.33 = +3.3 V Thus D1 is reverse biased as assumed, and the final result is I = 0 and V = 3.3 V. EXERCISES 4.4 Find the values of I and V in the circuits shown in Fig. E4.4. 4.1 The Ideal Diode 183 (a) (b) (c) (d) (e) (f ) Figure E4.4 Ans. (a) 2 mA, 0 V; (b) 0 mA, 5 V; (c) 0 mA, 5 V; (d) 2 mA, 0 V; (e) 3 mA, +3 V; (f) 4 mA, +1 V 4.5 Figure E4.5 shows a circuit for an ac voltmeter. It utilizes a moving-coil meter that gives a full-scale reading when the average current flowing through it is 1 mA. The moving-coil meter has a 50resistance. 184 Chapter 4 Diodes Moving-coil meter Figure E4.5 Find the value of R that results in the meter indicating a full-scale reading when the input sine-wave voltage vI is 20 V peak-to-peak. (Hint: The average value of half-sine waves is Vp/π .) Ans. 3.133 k 4.2 Terminal Characteristics of Junction Diodes The most common implementation of the diode utilizes a pn junction. We have studied the physics of the pn junction and derived its i–v characteristic in Chapter 3. That the pn junction is used to implement the diode function should come as no surprise: the pn junction can conduct substantial current in the forward direction and almost no current in the reverse direction. In this section we study the i–v characteristic of the pn junction diode in detail in order to prepare ourselves for diode circuit applications. Figure 4.7 shows the i–v characteristic of a silicon junction diode. The same characteristic is shown in Fig. 4.8 with some scales expanded and others compressed to reveal details. Note that the scale changes have resulted in the apparent discontinuity at the origin. As indicated, the characteristic curve consists of three distinct regions: 1. The forward-bias region, determined by v > 0 2. The reverse-bias region, determined by v < 0 3. The breakdown region, determined by v < −VZK These three regions of operation are described in the following sections. 4.2.1 The Forward-Bias Region The forward-bias—or simply forward—region of operation is entered when the terminal voltage v is positive. In the forward region the i–v relationship is closely approximated by i = IS ev/VT − 1 (4.1) In this equation1 IS is a constant for a given diode at a given temperature. A formula for IS in terms of the diode’s physical parameters and temperature was given in Eq. (3.41). The current 1Equation (4.1), the diode equation, is sometimes written to include a constant n in the exponential, i = IS(ev/nVT − 1) with n having a value between 1 and 2, depending on the material and the physical structure of the diode. Diodes using the standard integrated-circuit fabrication process exhibit n = 1 when operated under normal conditions. For simplicity, we shall use n = 1 throughout this book, unless otherwise specified. 4.2 Terminal Characteristics of Junction Diodes 185 Figure 4.7 The i–v characteristic of a silicon junction diode. Figure 4.8 The diode i–v relationship with some scales expanded and others compressed in order to reveal details. 186 Chapter 4 Diodes IS is usually called the saturation current (for reasons that will become apparent shortly). Another name for IS, and one that we will occasionally use, is the scale current. This name arises from the fact that IS is directly proportional to the cross-sectional area of the diode. Thus doubling of the junction area results in a diode with double the value of IS and, as the diode equation indicates, double the value of current i for a given forward voltage v. For “small-signal” diodes, which are small-size diodes intended for low-power applications, IS is on the order of 10−15 A. The value of IS is, however, a very strong function of temperature. As a rule of thumb, IS doubles in value for every 5°C rise in temperature. The voltage VT in Eq. (4.1) is a constant called the thermal voltage and is given by kT VT = q (4.2) where k = Boltzmann’s constant = 8.62 × 10−5 eV/K = 1.38 × 10−23 joules/kelvin T = the absolute temperature in kelvins = 273 + temperature in °C q = the magnitude of electronic charge = 1.60 × 10−19 coulomb Substituting k = 8.62 × 10−5 eV/K into Eq. (4.2) gives VT = 0.0862T , mV (4.2a) Thus, at room temperature (20°C) the value of VT is 25.3 mV. In rapid approximate circuit analysis we shall use VT 25 mV at room temperature.2 For appreciable current i in the forward direction, specifically for i IS, Eq. (4.1) can be approximated by the exponential relationship i ISev /VT (4.3) This relationship can be expressed alternatively in the logarithmic form i v = VT ln IS (4.4) where ln denotes the natural (base e) logarithm. The exponential relationship of the current i to the voltage v holds over many decades of current (a span of as many as seven decades—i.e., a factor of 107—can be found). This is quite a remarkable property of junction diodes, one that is also found in bipolar junction transistors and that has been exploited in many interesting applications. Let us consider the forward i–v relationship in Eq. (4.3) and evaluate the current I1 corresponding to a diode voltage V1: I1 = IS eV1 /VT Similarly, if the voltage is V2, the diode current I2 will be I2 = IS eV2 /VT 2A slightly higher ambient temperature (25°C or so) is usually assumed for electronic equipment operating inside a cabinet. At this temperature, VT 25.8 mV. Nevertheless, for the sake of simplicity and to promote rapid circuit analysis, we shall use the more arithmetically convenient value of VT 25 mV throughout this book. 4.2 Terminal Characteristics of Junction Diodes 187 These two equations can be combined to produce which can be rewritten as or, in terms of base-10 logarithms, I = e 2 (V2 −V1 )/VT I1 V2 − V1 = VT ln I2 I1 V2 − V1 = 2.3 VT log I2 I1 (4.5) This equation simply states that for a decade (factor of 10) change in current, the diode voltage drop changes by 2.3VT , which is approximately 60 mV. This also suggests that the diode i–v relationship is most conveniently plotted on semilog paper. Using the vertical, linear axis for v and the horizontal, log axis for i, one obtains a straight line with a slope of 60 mV per decade of current. A glance at the i–v characteristic in the forward region (Fig. 4.8) reveals that the current is negligibly small for v smaller than about 0.5 V. This value is usually referred to as the cut-in voltage. It should be emphasized, however, that this apparent threshold in the characteristic is simply a consequence of the exponential relationship. Another consequence of this relationship is the rapid increase of i. Thus, for a “fully conducting” diode, the voltage drop lies in a narrow range, approximately 0.6 V to 0.8 V. This gives rise to a simple “model” for the diode where it is assumed that a conducting diode has approximately a 0.7-V drop across it. Diodes with different current ratings (i.e., different areas and correspondingly different IS) will exhibit the 0.7-V drop at different currents. For instance, a small-signal diode may be considered to have a 0.7-V drop at i = 1 mA, while a higher-power diode may have a 0.7-V drop at i = 1 A. We will study the topics of diode-circuit analysis and diode models in the next section. Example 4.3 A silicon diode said to be a 1-mA device displays a forward voltage of 0.7 V at a current of 1 mA. Evaluate the junction scaling constant IS. What scaling constants would apply for a 1-A diode of the same manufacture that conducts 1 A at 0.7 V? Solution Since then i = ISev/VT IS = ie−v/VT 188 Chapter 4 Diodes Example 4.3 continued For the 1-mA diode: IS = 10−3e−700/25 = 6.9 × 10−16 A The diode conducting 1 A at 0.7 V corresponds to one-thousand 1-mA diodes in parallel with a total junction area 1000 times greater. Thus IS is also 1000 times greater, IS = 6.9 × 10−13 A Since both IS and VT are functions of temperature, the forward i–v characteristic varies with temperature, as illustrated in Fig. 4.9. At a given constant diode current, the voltage drop across the diode decreases by approximately 2 mV for every 1°C increase in temperature. The change in diode voltage with temperature has been exploited in the design of electronic thermometers. Ϫ2 mVր°C Figure 4.9 Temperature dependence of the diode forward characteristic. At a constant current, the voltage drop decreases by approx- imately 2 mV for every 1°C increase in temperature. EXERCISES 4.6 Find the change in diode voltage if the current changes from 0.1 mA to 10 mA. Ans. 120 mV 4.7 A silicon junction diode has v = 0.7 V at i = 1 mA. Find the voltage drop at i = 0.1 mA and i = 10 mA. Ans. 0.64 V; 0.76 V 4.8 Using the fact that a silicon diode has IS = 10−14 A at 25°C and that IS increases by 15% per °C rise in temperature, find the value of IS at 125°C. Ans. 1.17 × 10−8 A 4.2 Terminal Characteristics of Junction Diodes 189 4.2.2 The Reverse-Bias Region The reverse-bias region of operation is entered when the diode voltage v is made negative. Equation (4.1) predicts that if v is negative and a few times larger than VT (25 mV) in magnitude, the exponential term becomes negligibly small compared to unity, and the diode current becomes i −IS That is, the current in the reverse direction is constant and equal to IS. This constancy is the reason behind the term saturation current. Real diodes exhibit reverse currents that, though quite small, are much larger than IS. For instance, a small-signal diode whose IS is on the order of 10−14 A to 10−15 A could show a reverse current on the order of 1 nA. The reverse current also increases somewhat with the increase in magnitude of the reverse voltage. Note that because of the very small magnitude of the current, these details are not clearly evident on the diode i–v characteristic of Fig. 4.8. A large part of the reverse current is due to leakage effects. These leakage currents are proportional to the junction area, just as IS is. Their dependence on temperature, however, is different from that of IS. Thus, whereas IS doubles for every 5°C rise in temperature, the corresponding rule of thumb for the temperature dependence of the reverse current is that it doubles for every 10°C rise in temperature. EXERCISE 4.9 The diode in the circuit of Fig. E4.9 is a large high-current device whose reverse leakage is reasonably independent of voltage. If V = 1 V at 20°C, find the value of V at 40°C and at 0°C. Figure E4.9 Ans. 4 V; 0.25 V 190 Chapter 4 Diodes 4.2.3 The Breakdown Region The third distinct region of diode operation is the breakdown region, which can be easily identified on the diode i–v characteristic in Fig. 4.8. The breakdown region is entered when the magnitude of the reverse voltage exceeds a threshold value that is specific to the particular diode, called the breakdown voltage. This is the voltage at the “knee” of the i–v curve in Fig. 4.8 and is denoted VZK , where the subscript Z stands for zener (see Section 3.5.3) and K denotes knee. As can be seen from Fig. 4.8, in the breakdown region the reverse current increases rapidly, with the associated increase in voltage drop being very small. Diode breakdown is normally not destructive, provided the power dissipated in the diode is limited by external circuitry to a “safe” level. This safe value is normally specified on the device data sheets. It therefore is necessary to limit the reverse current in the breakdown region to a value consistent with the permissible power dissipation. The fact that the diode i–v characteristic in breakdown is almost a vertical line enables it to be used in voltage regulation. This subject will be studied in Section 4.5. 4.3 Modeling the Diode Forward Characteristic Having studied the diode terminal characteristics we are now ready to consider the analysis of circuits employing forward-conducting diodes. Figure 4.10 shows such a circuit. It consists of a dc source VDD, a resistor R, and a diode. We wish to analyze this circuit to determine the diode voltage VD and current ID. To aid in our analysis, we need to represent the diode with a model. There are a variety of diode models, of which we now know two: the ideal-diode model and the exponential model. In the following discussion we shall assess the suitability of these two models in various analysis situations. Also, we shall develop and comment on other models. This material, besides being useful in the analysis and design of diode circuits, establishes a foundation for the modeling of transistor operation that we will study in the next three chapters. 4.3.1 The Exponential Model The most accurate description of the diode operation in the forward region is provided by the exponential model. Unfortunately, however, its severely nonlinear nature makes this model the most difficult to use. To illustrate, let’s analyze the circuit in Fig. 4.10 using the exponential diode model. Assuming that VDD is greater than 0.5 V or so, the diode current will be much greater than IS, and we can represent the diode i–v characteristic by the exponential relationship, ID + VD – Figure 4.10 A simple circuit used to illustrate the analysis of circuits in which the diode is forward conducting. 4.3 Modeling the Diode Forward Characteristic 191 resulting in ID = IS eVD /VT (4.6) The other equation that governs circuit operation is obtained by writing a Kirchhoff loop equation, resulting in ID = VDD − VD R (4.7) Assuming that the diode parameter IS is known, Eqs. (4.6) and (4.7) are two equations in the two unknown quantities ID and VD. Two alternative ways for obtaining the solution are graphical analysis and iterative analysis. 4.3.2 Graphical Analysis Using the Exponential Model Graphical analysis is performed by plotting the relationships of Eqs. (4.6) and (4.7) on the i–v plane. The solution can then be obtained as the coordinates of the point of intersection of the two graphs. A sketch of the graphical construction is shown in Fig. 4.11. The curve represents the exponential diode equation (Eq. 4.6), and the straight line represents Eq. (4.7). Such a straight line is known as the load line, a name that will become more meaningful in later chapters. The load line intersects the diode curve at point Q, which represents the operating point of the circuit. Its coordinates give the values of ID and VD. Graphical analysis aids in the visualization of circuit operation. However, the effort involved in performing such an analysis, particularly for complex circuits, is too great to be justified in practice. Figure 4.11 Graphical analysis of the circuit in Fig. 4.10 using the exponential diode model. 4.3.3 Iterative Analysis Using the Exponential Model Equations (4.6) and (4.7) can be solved using a simple iterative procedure, as illustrated in the following example. 192 Chapter 4 Diodes Example 4.4 Determine the current ID and the diode voltage VD for the circuit in Fig. 4.10 with VDD = 5 V and R = 1 k . Assume that the diode has a current of 1 mA at a voltage of 0.7 V. Solution To begin the iteration, we assume that VD = 0.7 V and use Eq. (4.7) to determine the current, ID = VDD − VD R = 5 − 0.7 = 4.3 mA 1 We then use the diode equation to obtain a better estimate for VD. This can be done by employing Eq. (4.5), namely, V2 − V1 = 2.3VT log I2 I1 Substituting 2.3VT = 60 mV, we have V2 = V1 + 0.06 log I2 I1 Substituting V1 = 0.7 V, I1 = 1 mA, and I2 = 4.3 mA results in V2 = 0.738 V. Thus the results of the first iteration are ID = 4.3 mA and VD = 0.738 V. The second iteration proceeds in a similar manner: 5 − 0.738 ID = 1 = 4.262 mA V2 = 0.738 + 0.06 log 4.262 4.3 = 0.738 V Thus the second iteration yields ID = 4.262 mA and VD = 0.738 V. Since these values are very close to the values obtained after the first iteration, no further iterations are necessary, and the solution is ID = 4.262 mA and VD = 0.738 V. 4.3.4 The Need for Rapid Analysis The iterative analysis procedure utilized in the example above is simple and yields accurate results after two or three iterations. Nevertheless, there are situations in which the effort and time required are still greater than can be justified. Specifically, if one is doing a pencil-and-paper design of a relatively complex circuit, rapid circuit analysis is a necessity. 4.3 Modeling the Diode Forward Characteristic 193 Through quick analysis, the designer is able to evaluate various possibilities before deciding on a suitable circuit design. To speed up the analysis process, one must be content with less precise results. This, however, is seldom a problem, because the more accurate analysis can be postponed until a final or almost-final design is obtained. Accurate analysis of the almost-final design can be performed with the aid of a computer circuit-analysis program such as SPICE (see Appendix B and the website). The results of such an analysis can then be used to further refine or “fine-tune” the design. To speed up the analysis process, we must find a simpler model for the diode forward characteristic. 4.3.5 The Constant-Voltage-Drop Model The simplest and most widely used diode model is the constant-voltage-drop model. This model is based on the observation that a forward-conducting diode has a voltage drop that varies in a relatively narrow range, say, 0.6 to 0.8 V. The model assumes this voltage to be constant at a value, say, 0.7 V. This development is illustrated in Fig. 4.12. The constant-voltage-drop model is the one most frequently employed in the initial phases of analysis and design. This is especially true if at these stages one does not have detailed information about the diode characteristics, which is often the case. i i 0.7 V v (a) 0 0.7 V v (b) i ϩ vD Ϫ i Ͼ 0, vD ϭ 0.7 V (c) Figure 4.12 Development of the diode constant-voltage-drop model: (a) the exponential characteristic; (b) approximating the exponential characteristic by a constant voltage, usually about 0.7 Vi; (c) the resulting model of the forward-conducting diodes. 194 Chapter 4 Diodes Finally, note that if we employ the constant-voltage-drop model to solve the problem in Example 4.4, we obtain VD = 0.7 V and ID = VDD − 0.7 R = 5 − 0.7 = 4.3 mA 1 which are not very different from the values obtained before with the more elaborate exponential model. 4.3.6 The Ideal-Diode Model In applications that involve voltages much greater than the diode voltage drop (0.6 V–0.8 V), we may neglect the diode voltage drop altogether while calculating the diode current. The result is the ideal-diode model, which we studied in Section 4.1. For the circuit in Example 4.4 (i.e., Fig. 4.10 with VDD = 5 V and R = 1 k ), utilization of the ideal-diode model leads to VD = 0 V ID = 5 − 1 0 = 5 mA which for a very quick analysis would not be bad as a gross estimate. However, with almost no additional work, the 0.7-V-drop model yields much more realistic results. We note, however, that the greatest utility of the ideal-diode model is in determining which diodes are on and which are off in a multidiode circuit, such as those considered in Section 4.1. EXERCISES 4.10 For the circuit in Fig. 4.10, find ID and VD for the case VDD = 5 V and R = 10 k . Assume that the diode has a voltage of 0.7 V at 1-mA current. Use (a) iteration and (b) the constant-voltage-drop model with VD = 0.7 V. Ans. (a) 0.43 mA, 0.68 V; (b) 0.43 mA, 0.7 V D4.11 Design the circuit in Fig. E4.11 to provide an output voltage of 2.4 V. Assume that the diodes available have 0.7-V drop at 1 mA. 4.3 Modeling the Diode Forward Characteristic 195 Figure E4.11 Ans. R = 139 4.12 Repeat Exercise 4.4 using the 0.7-V-drop model to obtain better estimates of I and V than those found in Exercise 4.4 (using the ideal-diode model). Ans. (a) 1.72 mA, 0.7 V; (b) 0 mA, 5 V; (c) 0 mA, 5 V; (d) 1.72 mA, 0.7 V; (e) 2.3 mA, +2.3 V; (f) 3.3 mA, +1.7 V 4.3.7 The Small-Signal Model Consider the situation in Fig. 4.13(a), where a dc voltage VDD establishes a dc current ID through the series combination of a resistance R and a diode D. The resulting diode voltage is denoted VD. As mentioned above, values of ID and VD can be obtained by solving the circuit using the diode exponential characteristic or, much more quickly, approximate values can be found using the diode constant-voltage-drop model. Next, consider the situation of VDD undergoing a small change VDD, as shown in Fig. 4.13(b). As indicated, the current ID changes by an increment ID, and the diode voltage VD changes by an increment VD. We wish to find a quick way to determine the values of these incremental changes. Toward that end, we develop a “small-signal” model for the diode. R ID R ID + ⌬ID ϩ ϩ VDD D VD ⌬VDD ϩ Ϫ D VD + ⌬VD Ϫ VDD Ϫ (a) (b) Figure 4.13 (a) A simple diode circuit; (b) the situation when VDD changes by VDD. 196 Chapter 4 Diodes Figure 4.14 Development of the diode small-signal model. Here the word signal emphasizes that in general, VDD can be a time-varying quantity. The qualifier “small” indicates that this diode model applies only when VD is kept sufficiently small, with “sufficiently” to be quantified shortly. To develop the diode small-signal model, refer to Fig. 4.14. We express the voltage across the diode as the sum of the dc voltage VD and the time-varying signal vd(t), vD(t) = VD + vd(t) (4.8) Correspondingly, the total instantaneous diode current iD(t) will be iD(t) = ISevD /VT (4.9) Substituting for vD from Eq. (4.8) gives iD(t) = ISe(VD +vd )/VT which can be rewritten iD(t) = ISeVD /VT evd /VT (4.10) 4.3 Modeling the Diode Forward Characteristic 197 In the absence of the signal vd(t), the diode voltage is equal to VD, and the diode current is ID, given by ID = IS eVD /VT (4.11) Thus, iD(t) in Eq. (4.10) can be expressed as iD(t) = IDevd /VT (4.12) Now if the amplitude of the signal vd(t) is kept sufficiently small such that vd 1 VT (4.13) then we may expand the exponential of Eq. (4.12) in a series and truncate the series after the first two terms to obtain the approximate expression iD(t) ID 1 + vd VT (4.14) This is the small-signal approximation. It is valid for signals whose amplitudes are smaller than about 5 mV (see Eq. 4.13, and recall that VT = 25 mV).3 From Eq. (4.14) we have iD(t) = ID + ID VT vd (4.15) Thus, superimposed on the dc current ID, we have a signal current component directly proportional to the signal voltage vd. That is, iD = ID + id (4.16) where id = ID VT vd (4.17) The quantity relating the signal current id to the signal voltage vd has the dimensions of conductance, mhos ( ), and is called the diode small-signal conductance. The inverse of this parameter is the diode small-signal resistance, or incremental resistance, rd, rd = VT ID (4.18) Note that the value of rd is inversely proportional to the bias current ID. 3For vd = 5 mV, vd/VT = 0.2. Thus the next term in the series expansion of the exponential will be 1 × 0.22 = 0.02, a factor of 10 lower than the linear term we kept. 2 198 Chapter 4 Diodes Additional insight into the small-signal approximation and the small-signal diode model can be obtained by considering again the graphical construction in Fig. 4.14. Here the diode is seen to be operating at a dc bias point Q characterized by the dc voltage VD and the corresponding dc current ID. Superimposed on VD we have a signal vd(t), assumed (arbitrarily) to have a triangular waveform. It is easy to see that using the small-signal approximation is equivalent to assuming that the signal amplitude is sufficiently small such that the excursion along the i–v curve is limited to a short almost-linear segment. The slope of this segment, which is equal to the slope of the tangent to the i–v curve at the operating point Q, is equal to the small-signal conductance. The reader is encouraged to prove that the slope of the i–v curve at i = ID is equal to ID/VT , which is 1/rd; that is, rd = 1 ∂ iD ∂v D iD = ID (4.19) From the preceding we conclude that superimposed on the quantities VD and ID that define the dc bias point, or quiescent point, of the diode will be the small-signal quantities vd(t) and id(t), which are related by the diode small-signal resistance rd evaluated at the bias point (Eq. 4.18). Thus the small-signal analysis can be performed separately from the dc bias analysis, a great convenience that results from the linearization of the diode characteristics inherent in the small-signal approximation. Specifically, after the dc analysis is performed, the small-signal equivalent circuit is obtained by eliminating all dc sources (i.e., short-circuiting dc voltage sources and open-circuiting dc current sources) and replacing the diode by its small-signal resistance. Thus, for the circuit in Fig. 4.13(b), the dc analysis is obtained by using the circuit in Fig. 4.13(a), while the incremental quantities ID and VD can be determined by using the small-signal equivalent circuit shown in Fig. 4.15. The following example should further illustrate the application of the small-signal model. ⌬VDD ϩ Ϫ R ⌬ID ϩ rd ⌬VD Ϫ Figure 4.15 Circuit for determining the incremental quantities ID and VD for the circuit in Figure 4.13(b). Note that replacing the diode with its small-signal resistance rd results in a linear circuit. Example 4.5 Consider the circuit shown in Fig. 4.16(a) for the case in which R = 10 k . The power supply V + has a dc value of 10 V on which is superimposed a 60-Hz sinusoid of 1-V peak amplitude. (This “signal” component of the power-supply voltage is an imperfection in the power-supply design. It is known as the power-supply ripple. More on this later.) Calculate both the dc voltage of the diode and the amplitude of the sine-wave signal appearing across it. Assume the diode to have a 0.7-V drop at 1-mA current. 4.3 Modeling the Diode Forward Characteristic 199 10 V ID R R ϩ VD vs ϩ Ϫ Ϫ ϩ rd vd Ϫ (a) (b) (c) Figure 4.16 (a) Circuit for Example 4.5. (b) Circuit for calculating the dc operating point. (c) Small-signal equivalent circuit. Solution Considering dc quantities only, we assume VD 0.7 V and calculate the diode dc current ID = 10 − 0.7 10 = 0.93 mA Since this value is very close to 1 mA, the diode voltage will be very close to the assumed value of 0.7 V. At this operating point, the diode incremental resistance rd is rd = VT ID = 25 0.93 = 26.9 The signal voltage across the diode can be found from the small-signal equivalent circuit in Fig. 4.16(c). Here vs denotes the 60-Hz 1-V peak sinusoidal component of V +, and vd is the corresponding signal across the diode. Using the voltage divider rule provides the peak amplitude of vd as follows: v d (peak) = Vˆs R rd + rd = 0.0269 1 10 + 0.0269 = 2.68 mV Finally, we note that since this value is quite small, our use of the small-signal model of the diode is justified. From the above we see that for a diode circuit that involves both dc and signal quantities, a small-signal equivalent circuit can be obtained by eliminating the dc sources and replacing each diode with its small-signal resistance rd. Such a circuit is linear and can be solved using linear circuit analysis. Finally, we note that while rd models the small-signal operation of the diode at low frequencies, its dynamic operation is modeled by the capacitances Cj and Cd, which we 200 Chapter 4 Diodes studied in Section 3.6 and which also are small-signal parameters. A complete model of the diode includes Cj and Cd in parallel with rd. 4.3.8 Use of the Diode Forward Drop in Voltage Regulation A further application of the diode small-signal model is found in a popular diode application, namely, the use of diodes to create a regulated voltage. A voltage regulator is a circuit whose purpose is to provide a constant dc voltage between its output terminals. The output voltage is required to remain as constant as possible in spite of (a) changes in the load current drawn from the regulator output terminal and (b) changes in the dc power-supply voltage that feeds the regulator circuit. Since the forward-voltage drop of the diode remains almost constant at approximately 0.7 V while the current through it varies by relatively large amounts, a forward-biased diode can make a simple voltage regulator. For instance, we have seen in Example 4.5 that while the 10-V dc supply voltage had a ripple of 2 V peak-to-peak (a ±10% variation), the corresponding ripple in the diode voltage was only about ±2.7 mV (a ±0.4% variation). Regulated voltages greater than 0.7 V can be obtained by connecting a number of diodes in series. For example, the use of three forward-biased diodes in series provides a voltage of about 2 V. One such circuit is investigated in the following example, which utilizes the diode small-signal model to quantify the efficacy of the voltage regulator that is realized. Example 4.6 Consider the circuit shown in Fig. 4.17. A string of three diodes is used to provide a constant voltage of about 2.1 V. We want to calculate the percentage change in this regulated voltage caused by (a) a ±10% change in the power-supply voltage, and (b) connection of a 1-k load resistance. 10Ϯ 1 V R = 1 k⍀ ϩ vO RL = 1 k⍀ Ϫ Figure 4.17 Circuit for Example 4.6. Solution With no load, the nominal value of the current in the diode string is given by I = 10 − 2.1 = 7.9 mA 1 4.3 Modeling the Diode Forward Characteristic 201 Thus each diode will have an incremental resistance of rd = VT I Thus, rd = 25 7.9 = 3.2 The three diodes in series will have a total incremental resistance of r = 3rd = 9.6 This resistance, along with the resistance R, forms a voltage divider whose ratio can be used to calculate the change in output voltage due to a ±10% (i.e., ±1-V) change in supply voltage. Thus the peak-to-peak change in output voltage will be vO = 2 r r + R = 0.0096 2 0.0096 + 1 = 19 mV peak-to-peak That is, corresponding to the ±1-V (±10%) change in supply voltage, the output voltage will change by ±9.5 mV or ±0.5%. Since this implies a change of about ±3.2 mV per diode, our use of the small-signal model is justified. When a load resistance of 1 k is connected across the diode string, it draws a current of approximately 2.1 mA. Thus the current in the diodes decreases by 2.1 mA, resulting in a decrease in voltage across the diode string given by vO = −2.1 × r = −2.1 × 9.6 = −20 mV Since this implies that the voltage across each diode decreases by about 6.7 mV, our use of the small-signal model is not entirely justified. Nevertheless, a detailed calculation of the voltage change using the exponential model results in vO = − 23 mV, which is not too different from the approximate value obtained using the incremental model. EXERCISES 4.13 Find the value of the diode small-signal resistance rd at bias currents of 0.1 mA, 1 mA, and 10 mA. Ans. 250 ; 25 ; 2.5 4.14 Consider a diode biased at 1 mA. Find the change in current as a result of changing the voltage by (a) –10 mV, (b) –5 mV, (c) +5 mV, and (d) +10 mV. In each case, do the calculations (i) using the small-signal model and (ii) using the exponential model. Ans. (a) –0.40, –0.33 mA; (b) –0.20, –0.18 mA; (c) +0.20, +0.22 mA; (d) +0.40, +0.49 mA D4.15 Design the circuit of Fig. E4.15 so that VO = 3 V when IL = 0, and VO changes by 20 mV per 1 mA of load current. (a) Use the small-signal model of the diode to find the value of R. (b) Specify the value of IS of each of the diodes. 202 Chapter 4 Diodes (c) For this design, use the diode exponential model to determine the actual change in VO when a current IL = 1 mA is drawn from the regulator. ϩ15 V R VO IL Figure E4.15 Ans. (a) R = 2.4 k ; (b) IS = 4.7 × 10−16 A; (c) –23 mV 4.4 Operation in the Reverse Breakdown Region—Zener Diodes The very steep i–v curve that the diode exhibits in the breakdown region (Fig. 4.8) and the almost-constant voltage drop that this indicates suggest that diodes operating in the breakdown region can be used in the design of voltage regulators. From the previous section, the reader will recall that voltage regulators are circuits that provide a constant dc output voltage in the face of changes in their load current and in the system power-supply voltage. This in fact turns out to be an important application of diodes operating in the reverse breakdown region, and special diodes are manufactured to operate specifically in the breakdown region. Such diodes are called breakdown diodes or, more commonly, as noted earlier, zener diodes. Figure 4.18 shows the circuit symbol of the zener diode. In normal applications of zener diodes, current flows into the cathode, and the cathode is positive with respect to the anode. Thus IZ and VZ in Fig. 4.18 have positive values. IZ VZ Figure 4.18 Circuit symbol for a zener diode. 4.4 Operation in the Reverse Breakdown Region—Zener Diodes 203 4.4.1 Specifying and Modeling the Zener Diode Figure 4.19 shows details of the diode i–v characteristic in the breakdown region. We observe that for currents greater than the knee current IZK (specified on the data sheet of the zener diode), the i–v characteristic is almost a straight line. The manufacturer usually specifies the voltage across the zener diode VZ at a specified test current, IZT . We have indicated these parameters in Fig. 4.19 as the coordinates of the point labeled Q. Thus a 6.8-V zener diode will exhibit a 6.8-V drop at a specified test current of, say, 10 mA. As the current through the zener deviates from IZT , the voltage across it will change, though only slightly. Figure 4.19 shows that corresponding to current change I the zener voltage changes by V, which is related to I by V = rz I where rz is the inverse of the slope of the almost-linear i–v curve at point Q. Resistance rz is the incremental resistance of the zener diode at operating point Q. It is also known as the dynamic resistance of the zener, and its value is specified on the device data sheet. Typically, rz is in the range of a few ohms to a few tens of ohms. Obviously, the lower the value of rz is, the more constant the zener voltage remains as its current varies, and thus the more ideal its performance becomes in the design of voltage regulators. In this regard, we observe from Fig. 4.19 that while rz remains low and almost constant over a wide range of current, its value increases considerably in the vicinity of the knee. Therefore, as a general design guideline, one should avoid operating the zener in this low-current region. Zener diodes are fabricated with voltages VZ in the range of a few volts to a few hundred volts. In addition to specifying VZ (at a particular current IZT ), rz, and IZK , the manufacturer ϪVZ ϪVZ0 ϪVZK i 0ϪIZK v Slope ϭ 1 rz Q ⌬V ⌬I ϪIZT (test current) ⌬V ϭ ⌬I rz Figure 4.19 The diode i–v characteristic with the breakdown region shown in some detail. 204 Chapter 4 Diodes Figure 4.20 Model for the zener diode. also specifies the maximum power that the device can safely dissipate. Thus a 0.5-W, 6.8-V zener diode can operate safely at currents up to a maximum of about 70 mA. The almost-linear i–v characteristic of the zener diode suggests that the device can be modeled as indicated in Fig. 4.20. Here VZ0 denotes the point at which the straight line of slope 1/rz intersects the voltage axis (refer to Fig. 4.19). Although VZ0 is shown in Fig. 4.19 to be slightly different from the knee voltage VZK , in practice their values are almost equal. The equivalent circuit model of Fig. 4.20 can be analytically described by VZ = VZ0 + rzIZ and it applies for IZ > IZK and, obviously, VZ > VZ0. (4.20) 4.4.2 Use of the Zener as a Shunt Regulator We now illustrate, by way of an example, the use of zener diodes in the design of shunt regulators, so named because the regulator circuit appears in parallel (shunt) with the load. Example 4.7 The 6.8-V zener diode in the circuit of Fig. 4.21(a) is specified to have VZ = 6.8 V at IZ = 5 mA, rz = 20 , and IZK = 0.2 mA. The supply voltage V + is nominally 10 V but can vary by ±1 V. (a) Find VO with no load and with V + at its nominal value. (b) Find the change in VO resulting from the ±1-V change in V +. Note that in mV/V, is known as line regulation. VO/ V + , usually expressed (c) Find the change in VO resulting from connecting a load resistance RL that draws a current IL = 1 mA, and hence find the load regulation VO/ IL in mV/mA. (d) Find the change in VO when RL = 2 k . (e) Find the value of VO when RL = 0.5 k . (f) What is the minimum value of RL for which the diode still operates in the breakdown region? 4.4 Operation in the Reverse Breakdown Region—Zener Diodes 205 1 V) I IZ ϩ IL VO Ϫ (a) (b) Figure 4.21 (a) Circuit for Example 4.7. (b) The circuit with the zener diode replaced with its equivalent circuit model. Solution First we must determine the value of the parameter VZ0 of the zener diode model. Substituting VZ = 6.8 V, IZ = 5 mA, and rz = 20 in Eq. (4.20) yields VZ0 = 6.7 V. Figure 4.21(b) shows the circuit with the zener diode replaced with its model. (a) With no load connected, the current through the zener is given by IZ = I = V + − VZ0 R + rz = 10 − 6.7 0.5 + 0.02 = 6.35 mA Thus, VO = VZ0 + IZ rz = 6.7 + 6.35 × 0.02 = 6.83 V (b) For a ±1-V change in V +, the change in output voltage can be found from VO = V + rz R + rz = ±1 × 20 500 + 20 = ±38.5 mV Thus, Line regulation = 38.5 mV/V 206 Chapter 4 Diodes Example 4.7 continued (c) When a load resistance RL that draws a load current IL = 1 mA is connected, the zener current will decrease by 1 mA. The corresponding change in zener voltage can be found from VO = rz IZ = 20 × −1 = −20 mV Thus the load regulation is Load regulation ≡ VO = −20 mV/mA IL (d) When a load resistance of 2 k is connected, the load current will be approximately 6.8 V/2 k = 3.4 mA. Thus the change in zener current will be IZ = −3.4 mA, and the corresponding change in zener voltage (output voltage) will thus be VO = rz IZ = 20 × −3.4 = −68 mV This value could have been obtained by multiplying the load regulation by the value of IL (3.4 mA). (e) An RL of 0.5 k would draw a load current of 6.8/0.5 = 13.6 mA. This is not possible, because the current I supplied through R is only 6.4 mA (for V + = 10 V). Therefore, the zener must be cut off. If this is indeed the case, then VO is determined by the voltage divider formed by RL and R (Fig. 4.21a), VO = V + R RL + RL = 10 0.5 0.5 + 0.5 = 5 V Since this voltage is lower than the breakdown voltage of the zener, the diode is indeed no longer operating in the breakdown region. (f) For the zener to be at the edge of the breakdown region, IZ = IZK = 0.2 mA and VZ VZK 6.7 V. At this point the lowest (worst-case) current supplied through R is (9 − 6.7)/0.5 = 4.6 mA, and thus the load current is 4.6 −0.2 = 4.4 mA. The corresponding value of RL is RL = 6.7 4.4 1.5 k 4.4.3 Temperature Effects The dependence of the zener voltage VZ on temperature is specified in terms of its temperature coefficient TC, or temco as it is commonly known, which is usually expressed in mV/°C. The value of TC depends on the zener voltage, and for a given diode the TC varies with the operating current. Zener diodes whose VZ are lower than about 5 V exhibit a negative TC. On the other hand, zeners with higher voltages exhibit a positive TC. The TC of a zener diode with a VZ of about 5 V can be made zero by operating the diode at a specified current. Another commonly used technique for obtaining a reference voltage with low temperature coefficient 4.5 Rectifier Circuits 207 is to connect a zener diode with a positive temperature coefficient of about 2 mV/°C in series with a forward-conducting diode. Since the forward-conducting diode has a voltage drop of 0.7 V and a TC of about –2 mV/°C, the series combination will provide a voltage of (VZ + 0.7) with a TC of about zero. EXERCISES 4.16 A zener diode whose nominal voltage is 10 V at 10 mA has an incremental resistance of 50 . What voltage do you expect if the diode current is halved? Doubled? What is the value of VZ0 in the zener model? Ans. 9.75 V; 10.5 V; 9.5 V 4.17 A zener diode exhibits a constant voltage of 5.6 V for currents greater than five times the knee current. IZK is specified to be 1 mA. The zener is to be used in the design of a shunt regulator fed from a 15-V supply. The load current varies over the range of 0 mA to 15 mA. Find a suitable value for the resistor R. What is the maximum power dissipation of the zener diode? Ans. 470 ; 112 mW 4.18 A shunt regulator utilizes a zener diode whose voltage is 5.1 V at a current of 50 mA and whose incremental resistance is 7 . The diode is fed from a supply of 15-V nominal voltage through a 200- resistor. What is the output voltage at no load? Find the line regulation and the load regulation. Ans. 5.1 V; 33.8 mV/V; –7 mV/mA 4.4.4 A Final Remark Though simple and useful, zener diodes have lost a great deal of their popularity in recent years. They have been virtually replaced in voltage-regulator design by specially designed integrated circuits (ICs) that perform the voltage-regulation function much more effectively and with greater flexibility than zener diodes. 4.5 Rectifier Circuits One of the most important applications of diodes is in the design of rectifier circuits. A diode rectifier forms an essential building block of the dc power supplies required to power electronic equipment. A block diagram of such a power supply is shown in Fig. 4.22. As indicated, the power supply is fed from the 120-V (rms) 60-Hz ac line, and it delivers a dc voltage VO (usually in the range of 4 V to 20 V) to an electronic circuit represented by the load block. The dc voltage VO is required to be as constant as possible in spite of variations in the ac line voltage and in the current drawn by the load. The first block in a dc power supply is the power transformer. It consists of two separate coils wound around an iron core that magnetically couples the two windings. The primary winding, having N1 turns, is connected to the 120-V ac supply, and the secondary winding, having N2 turns, is connected to the circuit of the dc power supply. Thus an ac voltage vS of 120(N2/N1) V (rms) develops between the two terminals of the secondary winding. By 208 Chapter 4 Diodes t Figure 4.22 Block diagram of a dc power supply. selecting an appropriate turns ratio (N1/N2) for the transformer, the designer can step the line voltage down to the value required to yield the particular dc voltage output of the supply. For instance, a secondary voltage of 8-V rms may be appropriate for a dc output of 5 V. This can be achieved with a 15:1 turns ratio. In addition to providing the appropriate sinusoidal amplitude for the dc power supply, the power transformer provides electrical isolation between the electronic equipment and the power-line circuit. This isolation minimizes the risk of electric shock to the equipment user. The diode rectifier converts the input sinusoid vS to a unipolar output, which can have the pulsating waveform indicated in Fig. 4.22. Although this waveform has a nonzero average or a dc component, its pulsating nature makes it unsuitable as a dc source for electronic circuits, hence the need for a filter. The variations in the magnitude of the rectifier output are considerably reduced by the filter block in Fig. 4.22. In this section we shall study a number of rectifier circuits and a simple implementation of the output filter. The output of the rectifier filter, though much more constant than without the filter, still contains a time-dependent component, known as ripple. To reduce the ripple and to stabilize the magnitude of the dc output voltage against variations caused by changes in load current, a voltage regulator is employed. Such a regulator can be implemented using the zener shunt regulator configuration studied in Section 4.4. Alternatively, and much more commonly at present, an integrated-circuit regulator can be used. 4.5.1 The Half-Wave Rectifier The half-wave rectifier utilizes alternate half-cycles of the input sinusoid. Figure 4.23(a) shows the circuit of a half-wave rectifier. This circuit was analyzed in Section 4.1 (see Fig. 4.3) assuming an ideal diode. Using the more realistic constant-voltage-drop diode model, we obtain vO = 0, vO = vS − VD, vS < VD vS ≥ VD (4.21a) (4.21b) The transfer characteristic represented by these equations is sketched in Fig. 4.23(b), where VD = 0.7 V or 0.8 V. Figure 4.23(c) shows the output voltage obtained when the input vS is a sinusoid. In selecting diodes for rectifier design, two important parameters must be specified: the current-handling capability required of the diode, determined by the largest current the diode is expected to conduct, and the peak inverse voltage (PIV) that the diode must be able to vS ϩ Ϫ D ϩ R vO Ϫ (a) v VD Vs VD 4.5 Rectifier Circuits 209 vO 0 VD (b) Slope ϭ1 vS vS vO t (c) Figure 4.23 (a) Half-wave rectifier. (b) Transfer characteristic of the rectifier circuit. (c) Input and output waveforms. withstand without breakdown, determined by the largest reverse voltage that is expected to appear across the diode. In the rectifier circuit of Fig. 4.23(a), we observe that when vS is negative the diode will be cut off and vO will be zero. It follows that the PIV is equal to the peak of vS, PIV = Vs (4.22) It is usually prudent, however, to select a diode that has a reverse breakdown voltage at least 50% greater than the expected PIV. Before leaving the half-wave rectifier, the reader should note two points. First, it is possible to use the diode exponential characteristic to determine the exact transfer characteristic of the rectifier (see Problem 4.68). However, the amount of work involved is usually too great to be justified in practice. Of course, such an analysis can be easily done using a computer circuit-analysis program such as SPICE. Second, whether we analyze the circuit accurately or not, it should be obvious that this circuit does not function properly when the input signal is small. For instance, this circuit cannot be used to rectify an input sinusoid of 100-mV amplitude. For such an application one 210 Chapter 4 Diodes resorts to a so-called precision rectifier, a circuit utilizing diodes in conjunction with op amps. One such circuit is presented in Section 4.5.5. EXERCISE 4.19 For the half-wave rectifier circuit in Fig. 4.23(a), show the following: (a) For the half-cycles during which the diode conducts, conduction begins at an angle θ = sin−1 VD/Vs and terminates at (π – θ ), for a total conduction angle of (π – 2θ ). (b) The average value (dc component) of vO is VO (1/π )Vs −VD/2. (c) The peak diode current is Vs − VD /R. Find numerical values for these quantities for the case of 12-V (rms) sinusoidal input, VD 0.7 V, and R = 100 . Also, give the value for PIV. Ans. (a) θ = 2.4°, conduction angle = 175°; (b) 5.05 V; (c) 163 mA; 17 V 4.5.2 The Full-Wave Rectifier The full-wave rectifier utilizes both halves of the input sinusoid. To provide a unipolar output, it inverts the negative halves of the sine wave. One possible implementation is shown in Fig. 4.24(a). Here the transformer secondary winding is center-tapped to provide two equal voltages vS across the two halves of the secondary winding with the polarities indicated. Note that when the input line voltage (feeding the primary) is positive, both of the signals labeled vS will be positive. In this case D1 will conduct and D2 will be reverse biased. The current through D1 will flow through R and back to the center tap of the secondary. The circuit then behaves like a half-wave rectifier, and the output during the positive half-cycles when D1 conducts will be identical to that produced by the half-wave rectifier. Now, during the negative half-cycle of the ac line voltage, both of the voltages labeled vS will be negative. Thus D1 will be cut off while D2 will conduct. The current conducted by D2 will flow through R and back to the center tap. It follows that during the negative half-cycles while D2 conducts, the circuit behaves again as a half-wave rectifier. The important point, however, is that the current through R always flows in the same direction, and thus vO will be unipolar, as indicated in Fig. 4.24(c). The output waveform shown is obtained by assuming that a conducting diode has a constant voltage drop VD. Thus the transfer characteristic of the full-wave rectifier takes the shape shown in Fig. 4.24(b). The full-wave rectifier obviously produces a more “energetic” waveform than that provided by the half-wave rectifier. In almost all rectifier applications, one opts for a full-wave type of some kind. To find the PIV of the diodes in the full-wave rectifier circuit, consider the situation during the positive half-cycles. Diode D1 is conducting, and D2 is cut off. The voltage at the cathode of D2 is vO, and that at its anode is –vS. Thus the reverse voltage across D2 will be (vO + vS), which will reach its maximum when vO is at its peak value of (Vs – VD), and vS is at its peak value of Vs; thus, PIV = 2Vs − VD which is approximately twice that for the case of the half-wave rectifier. ϩ ac line voltage Ϫ D1 ϩ Center vS tap Ϫ ϩ vS Ϫ D2 (a) ϩ R vO Ϫ Slope Ӎ Ϫ1 4.5 Rectifier Circuits 211 vO ϪVD 0 VD (b) Slope Ӎ 1 vS v VD Vs vS ϪvS vO t (c) Figure 4.24 Full-wave rectifier utilizing a transformer with a center-tapped secondary winding: (a) circuit; (b) transfer characteristic assuming a constant-voltage-drop model for the diodes; (c) input and output waveforms. EXERCISE 4.20 For the full-wave rectifier circuit in Fig. 4.24(a), show the following: (a) The output is zero for an angle of 2 sin−1 VD/Vs centered around the zero-crossing points of the sine-wave input. (b) The average value (dc component) of vO is VO (2/π )Vs − VD. (c) The peak current through each diode is Vs − VD /R. Find the fraction (percentage) of each cycle during which vO > 0, the value of VO, the peak diode current, and the value of PIV, all for the case in which vS is a 12-V (rms) sinusoid, VD 0.7 V, and R = 100 . Ans. 97.4%; 10.1 V; 163 mA; 33.2 V 212 Chapter 4 Diodes 4.5.3 The Bridge Rectifier An alternative implementation of the full-wave rectifier is shown in Fig. 4.25(a). This circuit, known as the bridge rectifier because of the similarity of its configuration to that of the Wheatstone bridge, does not require a center-tapped transformer, a distinct advantage over the full-wave rectifier circuit of Fig. 4.24. The bridge rectifier, however, requires four diodes as compared to two in the previous circuit. This is not much of a disadvantage, because diodes are inexpensive and one can buy a diode bridge in one package. The bridge-rectifier circuit operates as follows: During the positive half-cycles of the input voltage, vS is positive, and thus current is conducted through diode D1, resistor R, and diode D2. Meanwhile, diodes D3 and D4 will be reverse biased. Observe that there are two diodes in series in the conduction path, and thus vO will be lower than vS by two diode drops (compared to one drop in the circuit previously discussed). This is somewhat of a disadvantage of the bridge rectifier. Next, consider the situation during the negative half-cycles of the input voltage. The secondary voltage vS will be negative, and thus −vS will be positive, forcing current through D3, R, and D4. Meanwhile, diodes D1 and D2 will be reverse biased. The important point to note, though, is that during both half-cycles, current flows through R in the same direction (from right to left), and thus vO will always be positive, as indicated in Fig. 4.25(b). To determine the peak inverse voltage (PIV) of each diode, consider the circuit during the positive half-cycles. The reverse voltage across D3 can be determined from the loop formed ϩ Ϫ ac line voltage v Vs ϩ vS Ϫ (a) 2 VD D4 D1 Ϫ vO ϩ R D2 D3 vS ϪvS vO t (b) Figure 4.25 The bridge rectifier: (a) circuit; (b) input and output waveforms. 4.5 Rectifier Circuits 213 by D3, R, and D2 as vD3(reverse) = vO + vD2(forward) Thus the maximum value of vD3 occurs at the peak of vO and is given by PIV = Vs − 2VD + VD = Vs − VD Observe that here the PIV is about half the value for the full-wave rectifier with a center-tapped transformer. This is another advantage of the bridge rectifier. Yet one more advantage of the bridge-rectifier circuit over that utilizing a center-tapped transformer is that only about half as many turns are required for the secondary winding of the transformer. Another way of looking at this point can be obtained by observing that each half of the secondary winding of the center-tapped transformer is utilized for only half the time. These advantages have made the bridge rectifier the most popular rectifier circuit configuration. EXERCISE 4.21 For the bridge-rectifier circuit of Fig. 4.25(a), use the constant-voltage-drop diode model to show that (a) the average (or dc component) of the output voltage is VO (2/π )Vs − 2VD and (b) the peak diode current is (Vs − 2VD)/R. Find numerical values for the quantities in (a) and (b) and the PIV for the case in which vS is a 12-V (rms) sinusoid, VD 0.7 V, and R = 100 . Ans. 9.4 V; 156 mA; 16.3 V 4.5.4 The Rectifier with a Filter Capacitor—The Peak Rectifier The pulsating nature of the output voltage produced by the rectifier circuits discussed above makes it unsuitable as a dc supply for electronic circuits. A simple way to reduce the variation of the output voltage is to place a capacitor across the load resistor. It will be shown that this filter capacitor serves to reduce substantially the variations in the rectifier output voltage. To see how the rectifier circuit with a filter capacitor works, consider first the simple circuit shown in Fig. 4.26. Let the input vI be a sinusoid with a peak value Vp, and assume the diode to be ideal. As vI goes positive, the diode conducts and the capacitor is charged so that vO = vI . This situation continues until vI reaches its peak value Vp. Beyond the peak, as vI decreases, the diode becomes reverse biased and the output voltage remains constant at the value Vp. In fact, theoretically speaking, the capacitor will retain its charge and hence its voltage indefinitely, because there is no way for the capacitor to discharge. Thus the circuit provides a dc voltage output equal to the peak of the input sine wave. This is a very encouraging result in view of our desire to produce a dc output. Next, we consider the more practical situation where a load resistance R is connected across the capacitor C, as depicted in Fig. 4.27(a). However, we will continue to assume the diode to be ideal. As before, for a sinusoidal input, the capacitor charges to the peak of the input Vp. Then the diode cuts off, and the capacitor discharges through the load resistance R. The capacitor discharge will continue for almost the entire cycle, until the time at which vI 214 Chapter 4 Diodes D (a) 0 (b) Figure 4.26 (a) A simple circuit used to illustrate the effect of a filter capacitor. (b) Input and output waveforms assuming an ideal diode. Note that the circuit provides a dc voltage equal to the peak of the input sine wave. The circuit is therefore known as a peak rectifier or a peak detector. exceeds the capacitor voltage. Then the diode turns on again and charges the capacitor up to the peak of vI, and the process repeats itself. Observe that to keep the output voltage from decreasing too much during capacitor discharge, one selects a value for C so that the time constant CR is much greater than the discharge interval. We are now ready to analyze the circuit in detail. Figure 4.27(b) shows the steady-state input and output voltage waveforms under the assumption that CR T , where T is the period of the input sinusoid. The waveforms of the load current iL = vO/R and of the diode current (when it is conducting) (4.23) iD = iC + iL = C dvI dt + iL are shown in Fig. 4.27(c). The following observations are in order: (4.24) (4.25) 1. The diode conducts for a brief interval, t, near the peak of the input sinusoid and supplies the capacitor with charge equal to that lost during the much longer discharge interval. The latter is approximately equal to the period T. vI ϩ Ϫ iD D iC iL ϩ C R vO Ϫ (a) T Vr Vp vI t1 t2 Conduction interval ⌬t vO vI ⌬t (b) ⌬t iD 4.5 Rectifier Circuits 215 t iL t (c) Figure 4.27 Voltage and current waveforms in the peak-rectifier circuit with CR T . The diode is assumed ideal. 2. Assuming an ideal diode, the diode conduction begins at time t1, at which the input vI equals the exponentially decaying output vO. Conduction stops at t2 shortly after the peak of vI ; the exact value of t2 can be determined by setting iD = 0 in Eq. (4.25). 3. During the diode-off interval, the capacitor C discharges through R, and thus vO decays exponentially with a time constant CR. The discharge interval begins just past the peak of vI. At the end of the discharge interval, which lasts for almost the entire period T, vO = Vp – Vr, where Vr is the peak-to-peak ripple voltage. When CR T , the value of Vr is small. 216 Chapter 4 Diodes 4. When Vr is small, vO is almost constant and equal to the peak value of vI . Thus the dc output voltage is approximately equal to Vp. Similarly, the current iL is almost constant, and its dc component IL is given by IL = Vp R (4.26) If desired, a more accurate expression for the output dc voltage can be obtained by taking the average of the extreme values of vO, VO = Vp − 1 2 Vr (4.27) With these observations in hand, we now derive expressions for Vr and for the average and peak values of the diode current. During the diode-off interval, vO can be expressed as vO = Vpe−t/CR At the end of the discharge interval we have Vp − Vr Vpe−T/CR Now, since CR T , we can use the approximation e−T/CR 1 − T /CR to obtain T Vr Vp CR (4.28) We observe that to keep Vr small we must select a capacitance C so that CR T . The ripple voltage Vr in Eq. (4.28) can be expressed in terms of the frequency f = 1/T as Vr = Vp fCR Using Eq. (4.26) we can express Vr by the alternate expression Vr = IL fC (4.29a) (4.29b) Note that an alternative interpretation of the approximation made above is that the capacitor discharges by means of a constant current IL = Vp/R. This approximation is valid as long as Vr Vp. Assuming that diode conduction ceases almost at the peak of vI, we can determine the conduction interval t from Vp cos(ω t) = Vp − Vr where ω = 2π f = 2π /T is the angular frequency of vI. Since (ω t) is a small angle, we can employ the approximation cos(ω t) 1 − 1 (ω t)2 to obtain 2 ω t 2Vr/Vp (4.30) We note that when Vr Vp, the conduction angle ω t will be small, as assumed. To determine the average diode current during conduction, iDav, we equate the charge that the diode supplies to the capacitor, Qsupplied = iCav t 4.5 Rectifier Circuits 217 where from Eq. (4.24), iCav = iDav − IL to the charge that the capacitor loses during the discharge interval, Qlost = CVr to obtain, using Eqs. (4.30) and (4.29a), iDav = IL 1 + π 2Vp/Vr (4.31) Observe that when Vr Vp, the average diode current during conduction is much greater than the dc load current. This is not surprising, since the diode conducts for a very short interval and must replenish the charge lost by the capacitor during the much longer interval in which it is discharged by IL. The peak value of the diode current, iDmax, can be determined by evaluating the expression in Eq. (4.25) at the onset of diode conduction—that is, at t = t1 = − t (where t = 0 is at the peak). Assuming that iL is almost constant at the value given by Eq. (4.26), we obtain iDmax = IL 1 + 2π 2Vp/Vr (4.32) From Eqs. (4.31) and (4.32), we see that for Vr Vp, iDmax 2iDav, which correlates with the fact that the waveform of iD is almost a right-angle triangle (see Fig. 4.27c). Example 4.8 Consider a peak rectifier fed by a 60-Hz sinusoid having a peak value Vp = 100 V. Let the load resistance R = 10 k . Find the value of the capacitance C that will result in a peak-to-peak ripple of 2 V. Also, calculate the fraction of the cycle during which the diode is conducting and the average and peak values of the diode current. Solution From Eq. (4.29a) we obtain the value of C as The conduction angle ω C = Vp Vr f R = 100 2 × 60 × 10 × 103 = 83.3 μF t is found from Eq. (4.30) as √ ω t = 2 × 2/100 = 0.2 rad Thus the diode conducts for (0.2/2π ) × 100 = 3.18% of the cycle. The average diode current is obtained from Eq. (4.31), where IL = 100/10 = 10 mA, as √ iDav = 10 1 + π 2 × 100/2 = 324 mA The peak diode current is found using Eq. (4.32), √ iDmax = 10 1 + 2π 2 × 100/2 = 638 mA 218 Chapter 4 Diodes Figure 4.28 Waveforms in the full-wave peak rectifier. The circuit of Fig. 4.27(a) is known as a half-wave peak rectifier. The full-wave rectifier circuits of Figs. 4.24(a) and 4.25(a) can be converted to peak rectifiers by including a capacitor across the load resistor. As in the half-wave case, the output dc voltage will be almost equal to the peak value of the input sine wave (Fig. 4.28). The ripple frequency, however, will be twice that of the input. The peak-to-peak ripple voltage, for this case, can be derived using a procedure identical to that above but with the discharge period T replaced by T /2, resulting in Vr = Vp 2 fCR (4.33) While the diode conduction interval, t, will still be given by Eq. (4.30), the average and peak currents in each of the diodes will be given by iDav = IL 1 + π Vp/2Vr (4.34) iDmax = IL 1 + 2π Vp/2Vr (4.35) Comparing these expressions with the corresponding ones for the half-wave case, we note that for the same values of Vp, f , R, and Vr (and thus the same IL), we need a capacitor half the size of that required in the half-wave rectifier. Also, the current in each diode in the full-wave rectifier is approximately half that which flows in the diode of the half-wave circuit. The analysis above assumed ideal diodes. The accuracy of the results can be improved by taking the diode voltage drop into account. This can be easily done by replacing the peak voltage Vp to which the capacitor charges with (Vp – VD) for the half-wave circuit and the full-wave circuit using a center-tapped transformer and with (Vp – 2VD) for the bridge-rectifier case. We conclude this section by noting that peak-rectifier circuits find application in signal-processing systems where it is required to detect the peak of an input signal. In such a case, the circuit is referred to as a peak detector. A particularly popular application of the peak detector is in the design of a demodulator for amplitude-modulated (AM) signals. We shall not discuss this application further here. 4.5 Rectifier Circuits 219 EXERCISES 4.22 Derive the expressions in Eqs. (4.33), (4.34), and (4.35). 4.23 Consider a bridge-rectifier circuit with a filter capacitor C placed across the load resistor R for the case in which the transformer secondary delivers a sinusoid of 12 V (rms) having a 60-Hz frequency and assuming VD = 0.8 V and a load resistance R = 100 . Find the value of C that results in a ripple voltage no larger than 1 V peak-to-peak. What is the dc voltage at the output? Find the load current. Find the diodes’ conduction angle. Provide the average and peak diode currents. What is the peak reverse voltage across each diode? Specify the diode in terms of its peak current and its PIV. Ans. 1281 μF; 15.4 V or (a better estimate) 14.9 V; 0.15 A; 0.36 rad (20.7°); 1.45 A; 2.74 A; 16.2 V. Thus select a diode with 3.5-A to 4-A peak current and a 20-V PIV rating. THE EARLIEST SEMICONDUCTOR DIODE: The cat’s whisker or crystal detector was the first electronic diode to be commercialized as an envelope detector for the radio-frequency signals used in radio telephony. The earliest diode, invented in Germany by Karl Ferdinand Braun, consisted of a small slab of galena (lead sulfide) to which contact was made by sharpened spring wire, which could be adjusted. For this and other contributions to early radios, Braun received the Nobel Prize in Physics in 1909. The silicon-based point-contact diode, later refined and packaged, was an important solid-state component of radar equipment during World War II. 4.5.5 Precision Half-Wave Rectifier—The Superdiode4 The rectifier circuits studied thus far suffer from having one or two diode drops in the signal paths. Thus these circuits work well only when the signal to be rectified is much larger than the voltage drop of a conducting diode (0.7 V or so). In such a case, the details of the diode forward characteristics or the exact value of the diode voltage do not play a prominent role in determining circuit performance. This is indeed the case in the application of rectifier circuits in power-supply design. There are other applications, however, where the signal to be rectified is small (e.g., on the order of 100 mV or so) and thus clearly insufficient to turn on a diode. Also, in instrumentation applications, the need arises for rectifier circuits with very precise and predictable transfer characteristics. For these applications, a class of circuits has been developed utilizing op amps (Chapter 2) together with diodes to provide precision rectification. In the following discussion, we study one such circuit. A comprehensive study of op amp–diode circuits is available on the website. 4This section requires knowledge of operational amplifiers (Chapter 2). 220 Chapter 4 Diodes (a) (b) Figure 4.29 (a) The “superdiode” precision half-wave rectifier and (b) its almost-ideal transfer characteristic. Note that when vI > 0 and the diode conducts, the op amp supplies the load current, and the source is conveniently buffered, an added advantage. Not shown are the op-amp power supplies. Figure 4.29(a) shows a precision half-wave rectifier circuit consisting of a diode placed in the negative-feedback path of an op amp, with R being the rectifier load resistance. The op amp, of course, needs power supplies for its operation. For simplicity, these are not shown in the circuit diagram. The circuit works as follows: If vI goes positive, the output voltage vA of the op amp will go positive and the diode will conduct, thus establishing a closed feedback path between the op amp’s output terminal and the negative input terminal. This negative-feedback path will cause a virtual short circuit to appear between the two input terminals of the op amp. Thus the voltage at the negative input terminal, which is also the output voltage vO, will equal (to within a few millivolts) that at the positive input terminal, which is the input voltage vI , vO = vI vI ≥ 0 Note that the offset voltage ( 0.7 V) exhibited in the simple half-wave rectifier circuit of Fig. 4.23 is no longer present. For the op-amp circuit to start operation, vI has to exceed only a negligibly small voltage equal to the diode drop divided by the op amp’s open-loop gain. In other words, the straight-line transfer characteristic vO–vI almost passes through the origin. This makes this circuit suitable for applications involving very small signals. Consider now the case when vI goes negative. The op amp’s output voltage vA will tend to follow and go negative. This will reverse-bias the diode, and no current will flow through resistance R, causing vO to remain equal to 0 V. Thus, for vI < 0, vO = 0. Since in this case the diode is off, the op amp will be operating in an open-loop fashion, and its output will be at its negative saturation level. The transfer characteristic of this circuit will be that shown in Fig. 4.29(b), which is almost identical to the ideal characteristic of a half-wave rectifier. The nonideal diode characteristics have been almost completely masked by placing the diode in the negative-feedback path of an op amp. This is another dramatic application of negative feedback, a subject we will study formally in Chapter 11. The combination of diode and op amp, shown in the dashed box in Fig. 4.29(a), is appropriately referred to as a “superdiode.” 4.6 Limiting and Clamping Circuits 221 EXERCISES 4.24 Consider the operational rectifier or superdiode circuit of Fig. 4.29(a), with R = 1 k . For vI = 10 mV, 1 V, and –1 V, what are the voltages that result at the rectifier output and at the output of the op amp? Assume that the op amp is ideal and that its output saturates at ±12 V. The diode has a 0.7-V drop at 1-mA current. Ans. 10 mV, 0.59 V; 1 V, 1.7 V; 0 V, –12 V 4.25 If the diode in the circuit of Fig. 4.29(a) is reversed, find the transfer characteristic vO as a function of vI. Ans. vO = 0 for vI ≥ 0; vO = vI for vI ≤ 0 4.6 Limiting and Clamping Circuits In this section, we shall present additional nonlinear circuit applications of diodes. 4.6.1 Limiter Circuits Figure 4.30 shows the general transfer characteristic of a limiter circuit. As indicated, for inputs in a certain range, L−/K ≤ vI ≤ L+/K, the limiter acts as a linear circuit, providing an output proportional to the input, vO = KvI. Although in general K can be greater than 1, the Figure 4.30 General transfer characteristic for a limiter circuit. 222 Chapter 4 Diodes circuits discussed in this section have K ≤ 1 and are known as passive limiters. (Examples of active limiters will be presented in Chapter 18.) If vI exceeds the upper threshold L+/K , the output voltage is limited or clamped to the upper limiting level L+. On the other hand, if vI is reduced below the lower limiting threshold L−/K , the output voltage vO is limited to the lower limiting level L−. The general transfer characteristic of Fig. 4.30 describes a double limiter—that is, a limiter that works on both the positive and negative peaks of an input waveform. Single limiters, of course, exist. Finally, note that if an input waveform such as that shown in Fig. 4.31 is fed to a double limiter, its two peaks will be clipped off. Limiters therefore are sometimes referred to as clippers. The limiter whose characteristics are depicted in Fig. 4.30 is described as a hard limiter. Soft limiting is characterized by smoother transitions between the linear region and the saturation regions and a slope greater than zero in the saturation regions, as illustrated in Fig. 4.32. Depending on the application, either hard or soft limiting may be preferred. Limiters find application in a variety of signal-processing systems. One of their simplest applications is in limiting the voltage between the two input terminals of an op amp to a value lower than the breakdown voltage of the transistors that make up the input stage of the op-amp circuit. We will have more to say on this and other limiter applications at later points in this book. Diodes can be combined with resistors to provide simple realizations of the limiter function. A number of examples are depicted in Fig. 4.33. In each part of the figure both the circuit and its transfer characteristic are given. The transfer characteristics are obtained using the constant-voltage-drop (VD = 0.7 V) diode model but assuming a smooth transition between the linear and saturation regions of the transfer characteristic. The circuit in Fig. 4.33(a) is that of the half-wave rectifier except that here the output is taken across the diode. For vI < 0.5 V, the diode is cut off, no current flows, and the voltage drop across R is zero; thus vO = vI . As vI exceeds 0.5 V, the diode turns on, eventually limiting Figure 4.31 Applying a sine wave to a limiter can result in clipping off its two peaks. 0 Figure 4.32 Soft limiting. 4.6 Limiting and Clamping Circuits 223 Figure 4.33 A variety of basic limiting circuits. vO to one diode drop (0.7 V). The circuit of Fig. 4.33(b) is similar to that in Fig. 4.33(a) except that the diode is reversed. Double limiting can be implemented by placing two diodes of opposite polarity in parallel, as shown in Fig. 4.33(c). Here the linear region of the characteristic is obtained for −0.5V ≤ vI ≤ 0.5 V. For this range of vI , both diodes are off and vO = vI . As vI exceeds 0.5 V, D1 turns on and eventually limits vO to +0.7 V. Similarly, as vI goes more negative than –0.5 V, D2 turns on and eventually limits vO to –0.7 V. The thresholds and saturation levels of diode limiters can be controlled by using strings of diodes and/or by connecting a dc voltage in series with the diode(s). The latter idea is illustrated in Fig. 4.33(d). Finally, rather than strings of diodes, we may use two zener diodes in series, as shown in Fig. 4.33(e). In this circuit, limiting occurs in the positive direction at a voltage of VZ2 + 0.7, where 0.7 V represents the voltage drop across zener diode Z1 when conducting in the forward direction. For negative inputs, Z1 acts as a zener, while Z2 conducts 224 Chapter 4 Diodes in the forward direction. It should be mentioned that pairs of zener diodes connected in series are available commercially for applications of this type under the name double-anode zener. More flexible limiter circuits are possible if op amps are combined with diodes and resistors. Examples of such circuits are discussed in Chapter 18. EXERCISE 4.26 Assuming the diodes to be ideal, describe the transfer characteristic of the circuit shown in Fig. E4.26. ϩ Ϫ Figure E4.26 Ans. vO = vI vO = 1 2 v I − 2.5 vO = 1 2 v I + 2.5 for −5 ≤ vI ≤ +5 for vI ≤ −5 for vI ≥ +5 4.6.2 The Clamped Capacitor or DC Restorer If in the basic peak-rectifier circuit, the output is taken across the diode rather than across the capacitor, an interesting circuit with important applications results. The circuit, called a dc restorer, is shown in Fig. 4.34 fed with a square wave. Because of the polarity in which the diode is connected, the capacitor will charge to a voltage vC with the polarity indicated in Fig. 4.34 and equal to the magnitude of the most negative peak of the input signal. Subsequently, the diode turns off and the capacitor retains its voltage indefinitely. If, Ϫ vC ϩ (a) (b) (c) Figure 4.34 The clamped capacitor or dc restorer with a square-wave input and no load. 4.6 Limiting and Clamping Circuits 225 for instance, the input square wave has the arbitrary levels –6 V and +4 V, then vC will be equal to 6 V. Now, since the output voltage vO is given by vO = vI + vC it follows that the output waveform will be identical to that of the input, except that it is shifted upward by vC volts. In our example the output will thus be a square wave with levels of 0 V and +10 V. Another way of visualizing the operation of the circuit in Fig. 4.34 is to note that because the diode is connected across the output with the polarity shown, it prevents the output voltage from going below 0 V (by conducting and charging up the capacitor, thus causing the output to rise to 0 V), but this connection will not constrain the positive excursion of vO. The output waveform will therefore have its lowest peak clamped to 0 V, which is why the circuit is called a clamped capacitor. It should be obvious that reversing the diode polarity will provide an output waveform whose highest peak is clamped to 0 V. In either case, the output waveform will have a finite average value or dc component. This dc component is entirely unrelated to the average value of the input waveform. As an application, consider a pulse signal being transmitted through a capacitively coupled or ac-coupled system. The capacitive coupling will cause the pulse train to lose whatever dc component it originally had. Feeding the resulting pulse waveform to a clamping circuit provides it with a well-determined dc component, a process known as dc restoration. This is why the circuit is also called a dc restorer. Restoring dc is useful because the dc component or average value of a pulse waveform is an effective measure of its duty cycle.5 The duty cycle of a pulse waveform can be modulated (in a process called pulsewidth modulation) and made to carry information. In such a system, detection or demodulation could be achieved simply by feeding the received pulse waveform to a dc restorer and then using a simple RC low-pass filter to separate the average of the output waveform from the superimposed pulses. When a load resistance R is connected across the diode in a clamping circuit, as shown in Fig. 4.35, the situation changes significantly. While the output is above ground, a current must flow in R. Since at this time the diode is off, this current obviously comes from the capacitor, (b) (a) (c) Figure 4.35 The clamped capacitor with a load resistance R. 5The duty cycle of a pulse waveform is the proportion of each cycle occupied by the pulse. In other words, it is the pulse width expressed as a fraction of the pulse period. 226 Chapter 4 Diodes thus causing the capacitor to discharge and the output voltage to fall. This is shown in Fig. 4.35 for a square-wave input. During the interval t0 to t1, the output voltage falls exponentially with time constant CR. At t1 the input decreases by Va volts, and the output attempts to follow. This causes the diode to conduct heavily and to quickly charge the capacitor. At the end of the interval t1 to t2, the output voltage would normally be a few tenths of a volt negative (e.g., –0.5 V). Then, as the input rises by Va volts (at t2), the output follows, and the cycle repeats itself. In the steady state the charge lost by the capacitor during the interval t0 to t1 is recovered during the interval t1 to t2. This charge equilibrium enables us to calculate the average diode current as well as the details of the output waveform. 4.6.3 The Voltage Doubler Figure 4.36(a) shows a circuit composed of two sections in cascade: a clamped capacitor formed by C1 and D1, and a peak rectifier formed by D2 and C2. When excited by a sinusoid of amplitude Vp the clamping section provides the voltage waveform vD1 shown, assuming ideal diodes, in Fig. 4.36(b). Note that while the positive peaks are clamped to 0 V, the negative peak reaches –2Vp. In response to this waveform, the peak-detector section provides across capacitor C2 a dc voltage equal to the negative peak of vD1, that is, −2Vp. Because the output voltage is double the input peak, the circuit is known as a voltage doubler. The technique can be extended to provide output dc voltages that are higher multiples of Vp. D vI = Vp sin vt D ϩ vO = 2Vp Ϫ (a) Vp vI 0 t vD1 ϪVp vO Ϫ2Vp (b) Figure 4.36 Voltage doubler: (a) circuit; (b) waveforms of the input voltage, the voltage across D1, and the output voltage vo = −2Vp. 4.7 Special Diode Types 227 EXERCISE 4.27 If the diode in the circuit of Fig. 4.34 is reversed, what will the dc component of vO become? Ans. –5 V 4.7 Special Diode Types In this section, we discuss briefly some important special types of diodes. 4.7.1 The Schottky-Barrier Diode (SBD) The Schottky-barrier diode (SBD) is formed by bringing metal into contact with a moderately doped n-type semiconductor material. The resulting metal–semiconductor junction behaves like a diode, conducting current in one direction (from the metal anode to the semiconductor cathode) and acting as an open circuit in the other, and is known as the Schottky-barrier diode or simply the Schottky diode. In fact, the current–voltage characteristic of the SBD is remarkably similar to that of a pn-junction diode, with two important exceptions: 1. In the SBD, current is conducted by majority carriers (electrons). Thus the SBD does not exhibit the minority-carrier charge-storage effects found in forward-biased pn junctions. As a result, Schottky diodes can be switched from on to off, and vice versa, much faster than is possible with pn-junction diodes. 2. The forward voltage drop of a conducting SBD is lower than that of a pn-junction diode. For example, an SBD made of silicon exhibits a forward voltage drop of 0.3 V to 0.5 V, compared to the 0.6 V to 0.8 V found in silicon pn-junction diodes. SBDs can also be made of gallium arsenide (GaAs) and, in fact, play an important role in the design of GaAs circuits.6 Gallium-arsenide SBDs exhibit forward voltage drops of about 0.7 V. Apart from GaAs circuits, Schottky diodes find application in the design of a special form of bipolar-transistor logic circuits, known as Schottky-TTL, where TTL stands for transistor–transistor logic. Before leaving the subject of Schottky-barrier diodes, it is important to note that not every metal–semiconductor contact is a diode. In fact, metal is commonly deposited on the semiconductor surface in order to make terminals for the semiconductor devices and to connect different devices in an integrated-circuit chip. Such metal–semiconductor contacts are known as ohmic contacts to distinguish them from the rectifying contacts that result in SBDs. Ohmic contacts are usually made by depositing metal on very heavily doped (and thus low-resistivity) semiconductor regions. (Recall that SBDs use moderately doped material.) 6The website accompanying this text contains material on GaAs circuits. 228 Chapter 4 Diodes 4.7.2 Varactors In Chapter 3 we learned that reverse-biased pn junctions exhibit a charge-storage effect that is modeled with the depletion-layer or junction capacitance Cj. As Eq. (3.49) indicates, Cj is a function of the reverse-bias voltage VR. This dependence turns out to be useful in a number of applications, such as the automatic tuning of radio receivers. Special diodes are therefore fabricated to be used as voltage-variable capacitors known as varactors. These devices are optimized to make the capacitance a strong function of voltage by arranging that the grading coefficient m is 3 or 4. 4.7.3 Photodiodes If a reverse-biased pn junction is illuminated—that is, exposed to incident light—the photons impacting the junction cause covalent bonds to break, and thus electron-hole pairs are generated in the depletion layer. The electric field in the depletion region then sweeps the liberated electrons to the n side and the holes to the p side, giving rise to a reverse current across the junction. This current, known as photocurrent, is proportional to the intensity of the incident light. Such a diode, called a photodiode, can be used to convert light signals into electrical signals. Photodiodes are usually fabricated using a compound semiconductor7 such as gallium arsenide. The photodiode is an important component of a growing family of circuits known as optoelectronics or photonics. As the name implies, such circuits utilize an optimum combination of electronics and optics for signal processing, storage, and transmission. Usually, electronics is the preferred means for signal processing, whereas optics is most suited for transmission and storage. Examples include fiber-optic transmission of telephone and television signals and the use of optical storage in CD-ROM computer discs. Optical transmission provides very wide bandwidths and low signal attenuation. Optical storage allows vast amounts of data to be stored reliably in a small space. Finally, we should note that without reverse bias, the illuminated photodiode functions as a solar cell. Usually fabricated from low-cost silicon, a solar cell converts light to electrical energy. 4.7.4 Light-Emitting Diodes (LEDs) The light-emitting diode (LED) performs the inverse of the function of the photodiode; it converts a forward current into light. The reader will recall from Chapter 3 that in a forward-biased pn junction, minority carriers are injected across the junction and diffuse into the p and n regions. The diffusing minority carriers then recombine with the majority carriers. Such recombination can be made to give rise to light emission. This can be done by fabricating the pn junction using a semiconductor of the type known as direct-bandgap materials. Gallium arsenide belongs to this group and can thus be used to fabricate light-emitting diodes. The light emitted by an LED is proportional to the number of recombinations that take place, which in turn is proportional to the forward current in the diode. 7Whereas an elemental semiconductor, such as silicon, uses an element from column IV of the periodic table, a compound semiconductor uses a combination of elements from columns III and V or II and VI. For example, GaAs is formed of gallium (column III) and arsenic (column V) and is thus known as a III-V compound. Summary 229 LEDs are very popular devices. They find application in the design of numerous types of displays, including the displays of laboratory instruments such as digital voltmeters. They can be made to produce light in a variety of colors. Furthermore, LEDs can be designed so as to produce coherent light with a very narrow bandwidth. The resulting device is a laser diode. Laser diodes find application in optical communication systems and in DVD players, among other things. Combining an LED with a photodiode in the same package results in a device known as an optoisolator. The LED converts an electrical signal applied to the optoisolator into light, which the photodiode detects and converts back to an electrical signal at the output of the optoisolator. Use of the optoisolator provides complete electrical isolation between the electrical circuit that is connected to the isolator’s input and the circuit that is connected to its output. Such isolation can be useful in reducing the effect of electrical interference on signal transmission within a system, and thus optoisolators are frequently employed in the design of digital systems. They can also be used in the design of medical instruments to reduce the risk of electrical shock to patients. Note that the optical coupling between an LED and a photodiode need not be accomplished inside a small package. Indeed, it can be implemented over a long distance using an optical fiber, as is done in fiber-optic communication links. FROM INDICATION TO ILLUMINATION: Light-emitting diodes (LEDs), which once served only as low-powered status indicators, are now lighting our way! Increasingly, automotive lighting uses LEDs; increasingly, too, LED bulbs of higher and higher power are replacing both incandescent and fluorescent lighting in homes and offices. Incandescent bulbs are only 5% efficient in the conversion of electricity into light—the other 95% is dissipated as heat. The light conversion efficiency of LEDs, however, is 60%. Moreover, LEDs last 25 times longer (25,000 hours) than incandescent bulbs and 3 times longer than fluorescents. Summary In the forward direction, the ideal diode conducts any current forced by the external circuit while displaying a zero voltage drop. The ideal diode does not conduct in the reverse direction; any applied voltage appears as reverse bias across the diode. The unidirectional-current-flow property makes the diode useful in the design of rectifier circuits. The forward conduction of practical silicon-junction diodes is accurately characterized by the relationship i = ISev/VT . A silicon diode conducts a negligible current until the forward voltage is at least 0.5 V. Then the current increases rapidly, with the voltage drop increasing by 60 mV for every decade of current change. In the reverse direction, a silicon diode conducts a current on the order of 10−9 A. This current is much greater than IS because of leakage effects and increases with the magnitude of reverse voltage. Beyond a certain value of reverse voltage (that depends on the diode), breakdown occurs, and current increases rapidly with a small corresponding increase in voltage. Diodes designed to operate in the breakdown region are called zener diodes. They are employed in the design of voltage regulators whose function is to provide a constant dc voltage that varies little with variations in power-supply voltage and/or load current. 230 Chapter 4 Diodes In many applications, a conducting diode is modeled as having a constant voltage drop, usually approximately 0.7 V. A diode biased to operate at a dc current ID has a small-signal resistance rd = VT /ID. Rectifiers convert ac voltages into unipolar voltages. Half-wave rectifiers do this by passing the voltage in half of each cycle and blocking the opposite-polarity voltage in the other half of the cycle. Full-wave rectifiers accomplish the task by passing the voltage in half of each cycle and inverting the voltage in the other half-cycle. The bridge-rectifier circuit is the preferred full-wave rectifier configuration. The variation of the output waveform of the rectifier is reduced considerably by connecting a capacitor C across the output load resistance R. The resulting circuit is the peak rectifier. The output waveform then consists of a dc voltage almost equal to the peak of the input sine wave, Vp, on which is superimposed a ripple component of frequency 2f (in the full-wave case) and of peak-to-peak amplitude Vr = Vp/2fCR. To reduce this ripple voltage further, a voltage regulator is employed. Combination of diodes, resistors, and possibly reference voltages can be used to design voltage limiters that prevent one or both extremities of the output waveform from going beyond predetermined values, the limiting level(s). Applying a time-varying waveform to a circuit consisting of a capacitor in series with a diode and taking the output across the diode provides a clamping function. Specifically, depending on the polarity of the diode, either the positive or negative peaks of the signal will be clamped to the voltage at the other terminal of the diode (usually ground). In this way the output waveform has a nonzero average or dc component, and the circuit is known as a dc restorer. By cascading a clamping circuit with a peak-rectifier circuit, a voltage doubler is realized. PROBLEMS Computer Simulation Problems Problems identified by the Multisim/PSpice icon are intended to demonstrate the value of using SPICE simulation to verify hand analysis and design, and to investigate important issues such as allowable signal swing and amplifier nonlinear distortion. Instructions to assist in setting up PSPice and Multisim simulations for all the indicated problems can be found in the corresponding files on the website. Note that if a particular parameter value is not specified in the problem statement, you are to make a reasonable assumption. to the terminals of an ideal diode. Describe two possible situations that result. What are the diode current and terminal voltage when (a) the connection is between the diode cathode and the positive terminal of the battery and (b) the anode and the positive terminal are connected? 4.2 For the circuits shown in Fig. P4.2 using ideal diodes, find the values of the voltages and currents indicated. 4.3 For the circuits shown in Fig. P4.3 using ideal diodes, find the values of the labeled voltages and currents. Section 4.1: The Ideal Diode 4.1 An AA flashlight cell, whose The´venin equivalent is a voltage source of 1.5 V and a resistance of 1 , is connected 4.4 In each of the ideal-diode circuits shown in Fig. P4.4, vI is a 1-kHz, 5-V peak sine wave. Sketch the waveform resulting at vO. What are its positive and negative peak values? = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem 3 3 3 3 Problems 231 CHAPTER 4 PROBLEMS 3 (a) Figure P4.2 D 3 (b) 3 (c) 3 (d) 3 2 D 2 Figure P4.3 3 (a) 2 D 2 D (b) D1 vI D1 D2 D1 D2 vO vI vO vI vO 1 k⍀ 1 k⍀ 1 k⍀ (a) (b) D1 vI D2 D2 vO vI 1 k⍀ D1 (c) D3 D1 vO vI 1 k⍀ D2 (d) (e) (f ) Figure P4.4 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem vO 1 k⍀ CHAPTER 4 PROBLEMS 232 Chapter 4 Diodes 1 k⍀ vI vO vI 1 k⍀ D1 D1 vI vO D2 1 k⍀ vO 1 k⍀ D1 (g) (h) (i) ϩ15 V vI Figure P4.4 continued D1 1 k⍀ ( j) 1 mA vO 1 k⍀ vO 1 k⍀ D1 D2 vI (k) 4.5 The circuit shown in Fig. P4.5 is a model for a battery charger. Here vI is a 6-V peak sine wave, D1 and D2 are ideal diodes, I is a 60-mA current source, and B is a 3-V battery. Sketch and label the waveform of the battery current iB. What is its peak value? What is its average value? If the peak value of vI is reduced by 10%, what do the peak and average values of iB become? 4.6 The circuits shown in Fig. P4.6 can function as logic gates for input voltages that are either high or low. Using “1” to denote the high value and “0” to denote the low value, prepare a table with four columns including all possible input combinations and the resulting values of X and Y. What logic function is X of A and B? What logic function is Y of A and B? For what values of A and B do X and Y have the same value? For what values of A and B do X and Y have opposite values? vI D1 I iB D2 vO B Figure P4.5 D1 A B D2 (a) Figure P4.6 A I B X D3 Y D4 I (b) = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 4 PROBLEMS Problems 233 D 4.7 For the logic gate of Fig. 4.5(a), assume ideal diodes and input voltage levels of 0 V and +5 V. Find a suitable value for R so that the current required from each of the input signal sources does not exceed 0.2 mA. D 4.8 Repeat Problem 4.7 for the logic gate of Fig. 4.5(b). 4.9 Assuming that the diodes in the circuits of Fig. P4.9 are ideal, find the values of the labeled voltages and currents. be ideal. Select a suitable value for R so that the peak diode current does not exceed 40 mA. What is the greatest reverse voltage that will appear across the diode? 4.12 Consider the rectifier circuit of Fig. 4.3(a) in the event that the input source vI has a source resistance Rs. For the case Rs = R and assuming the diode to be ideal, sketch and clearly label the transfer characteristic vO versus vI . ϩ3V 12 k⍀ ϩ3V 6 k⍀ 4.13 A symmetrical square wave of 5-V peak-to-peak amplitude and zero average is applied to a circuit resembling that in Fig. 4.3(a) and employing a 100- resistor. What is the peak output voltage that results? What is the average output voltage that results? What is the peak diode current? What is the average diode current? What is the maximum reverse voltage across the diode? DD DD 4.14 Repeat Problem 4.13 for the situation in which the average voltage of the square wave is 1 V, while its peak-to-peak value remains at 5 V. 6 k⍀ 12 k⍀ Ϫ3V (a) Ϫ3V (b) Figure P4.9 4.10 Assuming that the diodes in the circuits of Fig. P4.10 are ideal, utilize The´venin’s theorem to simplify the circuits and thus find the values of the labeled currents and voltages. D *4.15 Design a battery-charging circuit, resembling that in Fig. 4.4(a) and using an ideal diode, in which current flows to the 12-V battery 25% of the time with an average value of 100 mA. What peak-to-peak sine-wave voltage is required? What resistance is required? What peak diode current flows? What peak reverse voltage does the diode endure? If resistors can be specified to only one significant digit, and the peak-to-peak voltage only to the nearest volt, what design would you choose to guarantee the required charging current? What fraction of the cycle does diode current flow? What is the average diode current? What is the peak diode current? What peak reverse voltage does the diode endure? ϩ5V ϩ5 V ϩ3 V 10 k⍀ 10 k⍀ I 4.16 The circuit of Fig. P4.16 can be used in a signaling system using one wire plus a common ground return. At any moment, the input has one of three values: +3 V, 0 V, –3 V. ϪV ϩ 10 10 k⍀ 10 k⍀ D D (a) (b) Figure P4.10 D 4.11 For the rectifier circuit of Fig. 4.3(a), let the input sine wave have 120-V rms value and assume the diode to Figure P4.16 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 4 PROBLEMS 234 Chapter 4 Diodes What is the status of the lamps for each input value? (Note that the lamps can be located apart from each other and that there may be several of each type of connection, all on one wire!) Section 4.2: Terminal Characteristics of Junction Diodes 4.17 Calculate the value of the thermal voltage, VT , at –55°C, 0°C, +40°C, and +125°C. At what temperature is VT exactly 25 mV? 4.18 At what forward voltage does a diode conduct a current equal to 10,000IS? In terms of IS, what current flows in the same diode when its forward voltage is 0.7 V? 4.19 A diode for which the forward voltage drop is 0.7 V at 1.0 mA is operated at 0.5 V. What is the value of the current? 4.20 A particular diode is found to conduct 1 mA with a junction voltage of 0.7 V. What current will flow in this diode if the junction voltage is raised to 0.71 V? To 0.8 V? If the junction voltage is lowered to 0.69 V? To 0.6 V? What change in junction voltage will increase the diode current by a factor of 10? 4.21 The following measurements are taken on particular junction diodes for which V is the terminal voltage and I is the diode current. For each diode, estimate values of IS and the terminal voltage at 10% of the measured current. (a) V = 0.700 V at I = 1.00 A (b) V = 0.650 V at I = 1.00 mA (c) V = 0.650 V at I = 10 μA (d) V = 0.700 V at I = 100 mA 4.22 Listed below are the results of measurements taken on several different junction diodes. For each diode, the data provided are the diode current I and the corresponding diode voltage V. In each case, estimate IS, and the diode voltage at 10I and I/10. Figure P4.23 4.24 A junction diode is operated in a circuit in which it is supplied with a constant current I. What is the effect on the forward voltage of the diode if an identical diode is connected in parallel? 4.25 Two diodes with saturation currents IS1 and IS2 are connected in parallel with their cathodes joined together and connected to grounds. The two anodes are joined together and fed with a constant current I. Find the currents ID1 and ID2 that flow through the two diodes, and the voltage VD that appears across their parallel combination. 4.26 Four diodes are connected in parallel: anodes joined together and fed with a constant current I, and cathodes joined together and connected to ground. What relative junction areas should these diodes have if their currents must have binary-weighted ratios, with the smallest being 0.1 mA? What value of I is needed? 4.27 In the circuit shown in Fig. P4.27, D1 has 10 times the junction area of D2. What value of V results? To obtain a value for V of 60 mV, what current I2 is needed? (a) 10.0 mA, 700 mV (b) 1.0 mA, 700 mV (c) 10 A, 800 mV (d) 1 mA, 700 mV (e) 10 μA, 600 mV I1 10 mA D2 D1 4.23 The circuit in Fig. P4.23 utilizes three identical diodes having IS = 10−14 A. Find the value of the current I required to obtain an output voltage VO = 2.0 V. If a current of 1 mA is drawn away from the output terminal by a load, what is the change in output voltage? I2 3 mA ϩ V Ϫ Figure P4.27 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 4 PROBLEMS Problems 235 4.28 For the circuit shown in Fig. P4.28, both diodes are identical. Find the value of R for which V = 50 mV. junction temperature? What is the power dissipated in the diode in its final state? What is the temperature rise per watt of power dissipation? (This is called the thermal resistance.) I 10 mA D2 D1 ϩ R V *4.32 A designer of an instrument that must operate over a wide supply-voltage range, noting that a diode’s junction-voltage drop is relatively independent of junction current, considers the use of a large diode to establish a small relatively constant voltage. A power diode, for which the nominal current at 0.8 V is 10 A, is available. If the current source feeding the diode changes in the range 1 mA to 3 mA and if, in addition, the temperature changes by ±20°C, what is the expected range of diode voltage? Ϫ Figure P4.28 4.29 A diode fed with a constant current I = 1 mA has a voltage V = 690 mV at 20°C. Find the diode voltage at −20°C and at +85°C. 4.30 In the circuit shown in Fig. P4.30, D1 is a large-area, high-current diode whose reverse leakage is high and independent of applied voltage, while D2 is a much smaller, low-current diode. At an ambient temperature of 20°C, resistor R1 is adjusted to make VR1 = V2 = 520 mV. Subsequent measurement indicates that R1 is 520 k . What do you expect the voltages VR1 and V2 to become at 0°C and at 40°C? *4.33 As an alternative to the idea suggested in Problem 4.32, the designer considers a second approach to producing a relatively constant small voltage from a variable current supply: It relies on the ability to make quite accurate copies of any small current that is available (using a process called current mirroring). The designer proposes to use this idea to supply two diodes of different junction areas with equal currents and to measure their junction-voltage difference. Two types of diodes are available: for a forward voltage of 700 mV, one conducts 0.1 mA, while the other conducts 1 A. Now, for identical currents in the range of 1 mA to 3 mA supplied to each, what range of difference voltages result? What is the effect of a temperature change of ±20°C on this arrangement? ϩ10 V Section 4.3: Modeling the Diode Forward Characteristic R1 ϩ D1 V1 Ϫ ϩ D2 V2 Ϫ Figure P4.30 4.31 When a 10-A current is applied to a particular diode, it is found that the junction voltage immediately becomes 700 mV. However, as the power being dissipated in the diode raises its temperature, it is found that the voltage decreases and eventually reaches 600 mV. What is the apparent rise in *4.34 Consider the graphical analysis of the diode circuit of Fig. 4.10 with VDD = 1 V, R = 1 k , and a diode having IS = 10−15 A. Calculate a small number of points on the diode characteristic in the vicinity of where you expect the load line to intersect it, and use a graphical process to refine your estimate of diode current. What value of diode current and voltage do you find? Analytically, find the voltage corresponding to your estimate of current. By how much does it differ from the graphically estimated value? 4.35 Use the iterative-analysis procedure to determine the diode current and voltage in the circuit of Fig. 4.10 for VDD = 1 V, R = 1 k , and a diode having IS = 10−15 A. 4.36 A “1-mA diode” (i.e., one that has vD = 0.7 V at iD = 1 mA) is connected in series with a 500- resistor to a 1.0 V supply. = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem 236 Chapter 4 Diodes CHAPTER 4 PROBLEMS (a) Provide a rough estimate of the diode current you would expect. (b) Estimate the diode current more closely using iterative analysis. D 4.37 Assuming the availability of diodes for which vD = 0.75 V at iD = 1 mA, design a circuit that utilizes four diodes connected in series, in series with a resistor R connected to a 15-V power supply. The voltage across the string of diodes is to be 3.3 V. 4.38 A diode operates in a series circuit with a resistance R and a dc source V. A designer, considering using a constant-voltage model, is uncertain whether to use 0.7 V or 0.6 V for VD. For what value of V is the difference in the calculated values of current only 1%? For V = 3 V and R = 1 k , what two current estimates would result from the use of the two values of VD? What is their percentage difference? 4.39 A designer has a supply of diodes for which a current of 2 mA flows at 0.7 V. Using a 1-mA current source, the designer wishes to create a reference voltage of 1.3 V. Suggest a combination of series and parallel diodes that will do the job as well as possible. How many diodes are needed? What voltage is actually achieved? 4.40 Solve the problems in Example 4.2 using the constant-voltage-drop (VD = 0.7 V) diode model. 4.41 For the circuits shown in Fig. P4.2, using the constant-voltage-drop (VD = 0.7 V) diode model, find the voltages and currents indicated. 4.42 For the circuits shown in Fig. P4.3, using the constant-voltage-drop (VD = 0.7 V) diode model, find the voltages and currents indicated. 4.46 The small-signal model is said to be valid for voltage variations of about 5 mV. To what percentage current change does this correspond? (Consider both positive and negative signals.) What is the maximum allowable voltage signal (positive or negative) if the current change is to be limited to 10%? 4.47 In a particular circuit application, ten “20-mA diodes” (a 20-mA diode is a diode that provides a 0.7-V drop when the current through it is 20 mA) connected in parallel operate at a total current of 0.1 A. For the diodes closely matched, what current flows in each? What is the corresponding small-signal resistance of each diode and of the combination? Compare this with the incremental resistance of a single diode conducting 0.1 A. If each of the 20-mA diodes has a series resistance of 0.2 associated with the wire bonds to the junction, what is the equivalent resistance of the 10 parallel-connected diodes? What connection resistance would a single diode need in order to be totally equivalent? (Note: This is why the parallel connection of real diodes can often be used to advantage.) 4.48 In the circuit shown in Fig. P4.48, I is a dc current and vs is a sinusoidal signal. Capacitors C1 and C2 are very large; their function is to couple the signal to and from the diode but block the dc current from flowing into the signal source or the load (not shown). Use the diode small-signal model to show that the signal component of the output voltage is vo = vs VT VT + IRs If vs = 10 mV, find vo for I = 1 mA, 0.1 mA, and 1 μA. Let Rs = 1 k . At what value of I does vo become one-half of vs? Note that this circuit functions as a signal attenuator with the attenuation factor controlled by the value of the dc current I. 4.43 For the circuits in Fig. P4.9, using the constant-voltage-drop (VD = 0.7 V) diode model, find the values of the labeled currents and voltages. C1 C2 4.44 For the circuits in Fig. P4.10, utilize The´venin’s theo- ϩ rem to simplify the circuits and find the values of the labeled currents and voltages. Assume that conducting diodes can be vo represented by the constant-voltage-drop model (VD = 0.7 V). Ϫ D 4.45 Repeat Problem 4.11, representing the diode by the constant-voltage-drop (VD = 0.7 V) model. How different is the resulting design? Figure P4.48 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 4 PROBLEMS Problems 237 4.49 In the attenuator circuit of Fig. P4.48, let Rs = 10 k . The diode is a 1-mA device; that is, it exhibits a voltage drop of 0.7 V at a dc current of 1 mA. For small input signals, what value of current I is needed for vo/vs = 0.50? 0.10? 0.01? 0.001? In each case, what is the largest input signal that can be used while ensuring that the signal component of the diode current is limited to ±10% of its dc current? What output signals correspond? 4.50 In the capacitor-coupled attenuator circuit shown in Fig. P4.50, I is a dc current that varies from 0 mA to 1 mA, and C1 and C2 are large coupling capacitors. For very small input signals, so that the diodes can be represented by their etshmqauatilvvlv-aoislie=gnntIac,liwrrcheuseiitrseatanIndicstehisnusmrsdAh1 o.awFnidtnhdartdv2vvo,/oivg=iivfoerrd1tIrh+d=e2r0ds2mμaaAnlld,-s1hiegμnnAcael, 10 μA, 100 μA, 500 μA, 600 μA, 900 μA, 990 μA, and 1 mA. Note that this is a signal attenuator whose transmission is linearly controlled by the dc current I. (b) For a forward-conducting diode, what is the largest signal-voltage magnitude that it can support while the corresponding signal current is limited to 10% of the dc bias current? Now, for the circuit in Fig. P4.51, for 10-mV peak input, what is the smallest value of I for which the diode currents remain within ±10% of their dc values? (c) For I = 1 mA, what is the largest possible output signal for which the diode currents deviate by at most 10% of their dc values? What is the corresponding peak input? What is the total current in each diode? I D1 D3 vo vi ϩ Ϫ D2 D4 10 k⍀ 1 mA C2 I vo D1 D2 C1 vi I Figure P4.50 *4.51 In the circuit shown in Fig. P4.51, diodes D1 through D4 are identical, and each exhibits a voltage drop of 0.7 V at a 1-mA current. (a) For small input signals (e.g., 10-mV peak), find the small-signal equivalent circuit and use it to determine values of the small-signal transmission vo/vi for various values of I: 0 μA, 1 μA, 10 μA, 100 μA, 1 mA, and 10 mA. Figure P4.51 **4.52 In Problem 4.51 we investigated the operation of the circuit in Fig. P4.51 for small input signals. In this problem we wish to find the voltage-transfer characteristic (VTC) vO versus vI for −12 V ≤ vI ≤ 12 V for the case I = 1 mA and each of the diodes exhibits a voltage drop of 0.7 V at a current of 1 mA. Toward this end, use the diode exponential characteristic to construct a table that gives the values of: the current iO in the 10-k resistor, the current in each of the four diodes, the voltage drop across each of the four diodes, and the input voltage vI , for vO = 0, +1 V, +2 V, +5 V, +9 V, +9.9 V, +9.99 V, +10.5 V, +11 V, and +12 V. Use these data, with extrapolation to negative values of vI and vO, to sketch the required VTC. Also sketch the VTC that results if I is reduced to 0.5 mA. (Hint: From symmetry, observe that as vO increases and iO correspondingly increases, iD3 and iD2 increase by equal amounts and iD4 and iD1 decrease by (the same) equal amounts.) = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem 238 Chapter 4 Diodes CHAPTER 4 PROBLEMS *4.53 In the circuit shown in Fig. P4.53, I is a dc current and vi is a sinusoidal signal with small amplitude (less than 10 mV) and a frequency of 100 kHz. Representing the diode by its small-signal resistance rd, which is a function of I, sketch the small-signal equivalent circuit and use it to determine the sinusoidal output voltage Vo, and thus find the phase shift between Vi and Vo. Find the value of I that will provide a phase shift of –45°, and find the range of phase shift achieved as I is varied over the range of 0.1 times to 10 times this value. I (b) Generalize the expression above for the case of m diodes connected in series and the value of R adjusted so that the voltage across each diode is 0.7 V (and VO = 0.7m V). (c) Calculate the value of line regulation for the case V + = 15 V (nominally) and (i) m = 1 and (ii) m = 4. *4.55 Consider the voltage-regulator circuit shown in Fig P4.54 under the condition that a load current IL is drawn from the output terminal. (a) If the value of IL is sufficiently small that the corresponding change in regulator output voltage VO is small enough to justify using the diode small-signal model, show that vi ϩ Ϫ vo C 10 nF Figure P4.53 VO IL =− rd R This quantity is known as the load regulation and is usually expressed in mV/mA. (b) If the value of R is selected such that at no load the voltage across the diode is 0.7 V and the diode current is ID, show that the expression derived in (a) becomes *4.54 Consider the voltage-regulator circuit shown in Fig. P4.54. The value of R is selected to obtain an output voltage VO (across the diode) of 0.7 V. V R VO Figure P4.54 (a) Use the diode small-signal model to show that the change in output voltage corresponding to a change of 1 V in V + is VO = VT V + V + + VT − 0.7 This quantity is known as the line regulation and is usually expressed in mV/V. VO = − VT V +−0.7 IL ID V + − 0.7 + VT Select the lowest possible value for ID that results in a load regulation whose magnitude is ≤ 5 mV/mA. If V + is nominally 15 V, what value of R is required? Also, specify the diode required in terms of its IS. (c) Generalize the expression derived in (b) for the case of m diodes connected in series and R adjusted to obtain VO = 0.7m V at no load. D *4.56 Design a diode voltage regulator to supply 1.5 V to a 1.5-k load. Use two diodes specified to have a 0.7-V drop at a current of 1 mA. The diodes are to be connected to a +5-V supply through a resistor R. Specify the value for R. What is the diode current with the load connected? What is the increase resulting in the output voltage when the load is disconnected? What change results if the load resistance is reduced to 1 k ? To 750 ? To 500 ? (Hint: Use the small-signal diode model to calculate all changes in output voltage.) D *4.57 A voltage regulator consisting of two diodes in series fed with a constant-current source is used as a replacement for a single carbon–zinc cell (battery) of nominal voltage 1.5 V. The regulator load current varies from 2 mA to = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem Problems 239 CHAPTER 4 PROBLEMS 7 mA. Constant-current supplies of 5 mA, 10 mA, and 15 mA are available. Which would you choose, and why? What change in output voltage would result when the load current varies over its full range? **4.58 A particular design of a voltage regulator is shown in Fig. P4.58. Diodes D1 and D2 are 10-mA units; that is, each has a voltage drop of 0.7 V at a current of 10 mA. Use the diode exponential model and iterative analysis to answer the following questions: ϩ5 V (c) rz = 2 , VZ = 6.8 V, and VZK = 6.6 V (d) VZ = 18 V, IZT = 5 mA, and VZK = 17.6 V (e) IZT = 200 mA, VZ = 7.5 V, and rz = 1.5 Assuming that the power rating of a breakdown diode is established at about twice the specified zener current (IZT ), what is the power rating of each of the diodes described above? D 4.60 A designer requires a shunt regulator of approximately 20 V. Two kinds of zener diodes are available: 6.8-V devices with rz of 10 and 5.1-V devices with rz of 25 . For the two major choices possible, find the load regulation. In this calculation neglect the effect of the regulator resistance R. 200 ⍀ ϩ D1 VO 150 ⍀ D2 Ϫ 4.61 A shunt regulator utilizing a zener diode with an incremental resistance of 8 is fed through an 82- resistor. If the raw supply changes by 1.0 V, what is the corresponding change in the regulated output voltage? 4.62 A 9.1-V zener diode exhibits its nominal voltage at a test current of 20 mA. At this current the incremental resistance is specified as 10 . Find VZ0 of the zener model. Find the zener voltage at a current of 10 mA and at 50 mA. Figure P4.58 (a) What is the regulator output voltage VO with the 150load connected? (b) Find VO with no load. (c) With the load connected, to what value can the 5-V supply be lowered while maintaining the loaded output voltage within 0.1 V of its nominal value? (d) What does the loaded output voltage become when the 5-V supply is raised by the same amount as the drop found in (c)? (e) For the range of changes explored in (c) and (d), by what percentage does the output voltage change for each percentage change of supply voltage in the worst case? Section 4.4: Operation in the Reverse Breakdown Region—Zener Diodes 4.59 Partial specifications of a collection of zener diodes are provided below. For each, identify the missing parameter and estimate its value. Note from Fig. 4.19 that VZK VZ0 and IZK is very small. D 4.63 Design a 7.5-V zener regulator circuit using a 7.5-V zener specified at 10 mA. The zener has an incremental resistance rz = 30 and a knee current of 0.5 mA. The regulator operates from a 10-V supply and has a 1.5-k load. What is the value of R you have chosen? What is the regulator output voltage when the supply is 10% high? Is 10% low? What is the output voltage when both the supply is 10% high and the load is removed? What is the smallest possible load resistor that can be used while the zener operates at a current no lower than the knee current while the supply is 10% low? What is the load voltage in this case? D 4.64 Provide two designs of shunt regulators utilizing the 1N5235 zener diode, which is specified as follows: VZ = 6.8 V and rz = 5 for IZ = 20 mA; at IZ = 0.25 mA (nearer the knee), rz = 750 . For both designs, the supply voltage is nominally 9 V and varies by ±1 V. For the first design, assume that the availability of supply current is not a problem, and thus operate the diode at 20 mA. For the second design, assume that the current from the raw supply is limited, and therefore you are forced to operate the diode at 0.25 mA. For the purpose of these initial designs, assume no load. For each design find the value of R and the line regulation. (a) VZ = 10.0 V, VZK = 9.6 V, and IZT = 50 mA (b) IZT = 10 mA, VZ = 9.1 V, and rz = 30 D *4.65 A zener shunt regulator employs a 9.1-V zener diode for which VZ = 9.1 V at IZ = 9 mA, with rz = 40 and = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 4 PROBLEMS 240 Chapter 4 Diodes IZK = 0.5 mA. The available supply voltage of 15 V can vary as much as ±10%. For this diode, what is the value of VZ0? For a nominal load resistance RL of 1 k and a nominal zener current of 10 mA, what current must flow in the supply resistor R? For the nominal value of supply voltage, select a value for resistor R, specified to one significant digit, to provide at least that current. What nominal output voltage results? For a ±10% change in the supply voltage, what variation in output voltage results? If the load current is reduced by 50%, what increase in VO results? What is the smallest value of load resistance that can be tolerated while maintaining regulation when the supply voltage is low? What is the lowest possible output voltage that results? Calculate values for the line regulation and for the load regulation for this circuit using the numerical results obtained in this problem. D *4.66 It is required to design a zener shunt regulator to provide a regulated voltage of about 10 V. The available 10-V, 1-W zener of type 1N4740 is specified to have a 10-V drop at a test current of 25 mA. At this current, its rz is 7 . The raw supply, VS, available has a nominal value of 20 V but can vary by as much as ±25%. The regulator is required to supply a load current of 0 mA to 20 mA. Design for a minimum zener current of 5 mA. (a) Find VZ0. (b) Calculate the required value of R. (c) Find the line regulation. What is the change in VO expressed as a percentage, corresponding to the ±25% change in VS? (d) Find the load regulation. By what percentage does VO change from the no-load to the full-load condition? (e) What is the maximum current that the zener in your design is required to conduct? What is the zener power dissipation under this condition? Section 4.5: Rectifier Circuits 4.67 Consider the half-wave rectifier circuit of Fig. 4.23(a) with the diode reversed. Let vS be a sinusoid with 10-V peak amplitude, and let R = 1 k . Use the constant-voltage-drop diode model with VD = 0.7 V. (a) Sketch the transfer characteristic. (b) Sketch the waveform of vO. (c) Find the average value of vO. (d) Find the peak current in the diode. (e) Find the PIV of the diode. 4.68 Using the exponential diode characteristic, show that for vS and vO both greater than zero, the circuit of Fig. 4.23(a) has the transfer characteristic vO = vS − vD at iD = 1 mA − VT ln vO/R where vS and vO are in volts and R is in kilohms. Note that this relationship can be used to obtain the voltage transfer characteristic vO vs. vS by finding vS corresponding to various values of vO. 4.69 Consider a half-wave rectifier circuit with a triangular-wave input of 5-V peak-to-peak amplitude and zero average, and with R = 1 k . Assume that the diode can be represented by the constant-voltage-drop model with VD = 0.7 V. Find the average value of vO. 4.70 A half-wave rectifier circuit with a 1-k load operates from a 120-V (rms) 60-Hz household supply through a 12-to-1 step-down transformer. It uses a silicon diode that can be modeled to have a 0.7-V drop for any current. What is the peak voltage of the rectified output? For what fraction of the cycle does the diode conduct? What is the average output voltage? What is the average current in the load? 4.71 A full-wave rectifier circuit with a 1-k load operates from a 120-V (rms) 60-Hz household supply through a 6-to-1 transformer having a center-tapped secondary winding. It uses two silicon diodes that can be modeled to have a 0.7-V drop for all currents. What is the peak voltage of the rectified output? For what fraction of a cycle does each diode conduct? What is the average output voltage? What is the average current in the load? 4.72 A full-wave bridge-rectifier circuit with a 1-k load operates from a 120-V (rms) 60-Hz household supply through a 12-to-1 step-down transformer having a single secondary winding. It uses four diodes, each of which can be modeled to have a 0.7-V drop for any current. What is the peak value of the rectified voltage across the load? For what fraction of a cycle does each diode conduct? What is the average voltage across the load? What is the average current through the load? D 4.73 It is required to design a full-wave rectifier circuit using the circuit of Fig. 4.24 to provide an average output voltage of: (a) 10 V (b) 100 V = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem Problems 241 CHAPTER 4 PROBLEMS D D D D Figure P4.76 In each case find the required turns ratio of the transformer. Assume that a conducting diode has a voltage drop of 0.7 V. The ac line voltage is 120 V rms. D 4.74 Repeat Problem 4.73 for the bridge-rectifier circuit of Fig. 4.25. D 4.75 Consider the full-wave rectifier in Fig. 4.24 when the transformer turns ratio is such that the voltage across the entire secondary winding is 20 V rms. If the input ac line voltage (120 V rms) fluctuates by as much as ±10%, find the required PIV of the diodes. (Remember to use a factor of safety in your design.) 4.76 The circuit in Fig. P4.76 implements a complementary-output rectifier. Sketch and clearly label the waveforms of v + O and v − O . Assume a 0.7-V drop across each conducting diode. If the magnitude of the average of each output is to be 12 V, find the required amplitude of the sine wave across the entire secondary winding. What is the PIV of each diode? 4.77 Augment the rectifier circuit of Problem 4.70 with a capacitor chosen to provide a peak-to-peak ripple voltage of (i) 10% of the peak output and (ii) 1% of the peak output. In each case: (a) What average output voltage results? (b) What fraction of the cycle does the diode conduct? (c) What is the average diode current? (d) What is the peak diode current? 4.78 Repeat Problem 4.77 for the rectifier in Problem 4.71. 4.79 Repeat Problem 4.77 for the rectifier in Problem 4.72. D *4.80 It is required to use a peak rectifier to design a dc power supply that provides an average dc output voltage of 12 V on which a maximum of ±1-V ripple is allowed. The rectifier feeds a load of 200 . The rectifier is fed from the line voltage (120 V rms, 60 Hz) through a transformer. The diodes available have 0.7-V drop when conducting. If the designer opts for the half-wave circuit: (a) Specify the rms voltage that must appear across the transformer secondary. (b) Find the required value of the filter capacitor. (c) Find the maximum reverse voltage that will appear across the diode, and specify the PIV rating of the diode. (d) Calculate the average current through the diode during conduction. (e) Calculate the peak diode current. D *4.81 Repeat Problem 4.80 for the case in which the designer opts for a full-wave circuit utilizing a center-tapped transformer. D *4.82 Repeat Problem 4.80 for the case in which the designer opts for a full-wave bridge-rectifier circuit. D *4.83 Consider a half-wave peak rectifier fed with a voltage vS having a triangular waveform with 24-V peak-to-peak amplitude, zero average, and 1-kHz frequency. Assume that the diode has a 0.7-V drop when conducting. Let the load resistance R = 100 and the filter capacitor C = 100 μF. Find the average dc output voltage, the time interval during which the diode conducts, the average diode current during conduction, and the maximum diode current. = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 4 PROBLEMS 242 Chapter 4 Diodes D *4.84 Consider the circuit in Fig. P4.76 with two equal filter capacitors placed across the load resistors R. Assume that the diodes available exhibit a 0.7-V drop when conducting. Design the circuit to provide ±12-V dc output voltages with a peak-to-peak ripple no greater than 1 V. Each supply should be capable of providing 100-mA dc current to its load resistor R. Completely specify the capacitors, diodes, and the transformer. 4.85 The op amp in the precision rectifier circuit of Fig. P4.85 is ideal with output saturation levels of ±13 V. Assume that when conducting the diode exhibits a constant voltage drop of 0.7 V. Find v−, vO, and vA for: (a) vI = +1 V (b) vI = +3 V (c) vI = −1 V (d) vI = −3 V Also, find the average output voltage obtained when vI is a symmetrical square wave of 1-kHz frequency, 5-V amplitude, and zero average. (c) vI = −1 V (d) vI = −3 V R vI vϪ R D1 Ϫ D2 vO ϩ vA Figure P4.86 Section 4.6: Limiting and Clamping Circuits 4.87 Sketch the transfer characteristic vO versus vI for the limiter circuits shown in Fig. P4.87. All diodes begin conducting at a forward voltage drop of 0.5 V and have voltage drops of 0.7 V when conducting a current iD ≥ 1 mA. vI ϩ Ϫ D vO vA ϩ3 V vϪ R R Figure P4.85 RL vI vO 1 k⍀ (a) ϩ3 V 4.86 The op amp in the circuit of Fig. P4.86 is ideal with output saturation levels of ±12 V. The diodes exhibit a constant 0.7-V drop when conducting. Find v−, vA, and vO vI vO for: 1 k⍀ (a) vI = +1 V (b) vI = +3 V (b) Figure P4.87 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 4 PROBLEMS 1 k⍀ vI vO Ϫ3 V (c) 1 k⍀ vI vO Problems 243 the diodes can be represented by the constant-voltage-drop model with VD = 0.7 V. Also assume that the zener voltage is 6.8 V and that rz is negligibly small. *4.91 Plot the transfer characteristic of the circuit in Fig. P4.91 by evaluating vI corresponding to vO = 0.5 V, 0.6 V, 0.7 V, 0.8 V, 0 V, –0.5 V, –0.6 V, –0.7 V, and –0.8 V. Use the exponential model for the diodes, and assume that they have 0.7-V drops at 1-mA currents. Characterize the circuit as a hard or soft limiter. What is the value of K? Estimate L+ and L−. Ϫ3 V (d) Figure P4.87 continued 4.88 The circuits in Fig. P4.87(a) and (d) are connected as follows: The two input terminals are tied together, and the two output terminals are tied together. Sketch the transfer characteristic of the circuit resulting, assuming that the cut-in voltage of the diodes is 0.5 V and their voltage drop when conducting a current iD ≥ 1 mA is 0.7 V. 4.89 Repeat Problem 4.88 for the two circuits in Fig. P4.87(a) and (b) connected together as follows: The two input terminals are tied together, and the two output terminals are tied together. Figure P4.91 4.92 Design limiter circuits using only diodes and 10-k resistors to provide an output signal limited to the range: 4.90 Sketch and clearly label the transfer characteristic of the circuit in Fig. P4.90 for −15 V ≤ vI ≤ +15 V. Assume that (a) –0.7 V and above (b) +2.1 V and below (c) ±1.4 V D D D D Figure P4.90 Assume that each diode has a 0.7-V drop when conducting. 4.93 Design a two-sided limiting circuit using a resistor, two diodes, and two power supplies to feed a 1-k load with nominal limiting levels of ±2.2 V. Use diodes modeled by a constant 0.7 V. In the nonlimiting region, the voltage gain should be at least 0.94 V/V. **4.94 In the circuit shown in Fig. P4.94, the diodes exhibit a 0.7-V drop at 0.1 mA. For inputs over the range of ±5 V, use the diode exponential model to provide a calibrated sketch of the voltages at outputs B and C versus vA. For a 5-V peak, = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 4 PROBLEMS 244 Chapter 4 Diodes 100-Hz sinusoid applied at A, sketch the signals at nodes B and C. 5 k⍀ A B vI D2 D3 D1 D4 C 1 k⍀ ϩ1 V 3 k⍀ 1 k⍀ D1 vO D2 D3 1 k⍀ Figure P4.94 Ϫ2 V Figure P4.95 **4.95 Sketch and label the voltage-transfer characteristic vO versus vI of the circuit shown in Fig. P4.95 over a ±10-V range of input signals. Use the diode exponential model and assume that all diodes are 1-mA units (i.e., each exhibits a 0.7-V drop at a current of 1 mA). What are the slopes of the characteristics at the extreme ±10-V levels? 4.96 A clamped capacitor using an ideal diode with cathode grounded is supplied with a sine wave of 5-V rms. What is the average (dc) value of the resulting output? *4.97 For the circuits in Fig. P4.97, each utilizing an ideal diode (or diodes), sketch the output for the input shown. Label the most positive and most negative output levels. Assume CR T . = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem Problems 245 CHAPTER 4 PROBLEMS (a) (b) (c) (d) (e) (f ) Figure P4.97 (g) (h) CHAPTER 5 MOS Field-Effect Transistors (MOSFETs) Introduction 247 5.1 Device Structure and Physical Operation 248 5.2 Current–Voltage Characteristics 264 5.3 MOSFET Circuits at DC 276 5.4 The Body Effect and Other Topics 288 Summary 291 Problems 292 IN THIS CHAPTER YOU WILL LEARN 1. The physical structure of the MOS transistor and how it works. 2. How the voltage between two terminals of the transistor controls the current that flows through the third terminal, and the equations that describe these current–voltage characteristics. 3. How to analyze and design circuits that contain MOS transistors, resistors, and dc sources. Introduction Having studied the junction diode, which is the most basic two-terminal semiconductor device, we now turn our attention to three-terminal semiconductor devices. Three-terminal devices are far more useful than two-terminal ones because they can be used in a multitude of applications, ranging from signal amplification to digital logic and memory. The basic principle involved is the use of the voltage between two terminals to control the current flowing in the third terminal. In this way a three-terminal device can be used to realize a controlled source, which as we have learned in Chapter 1 is the basis for amplifier design. Also, in the extreme, the control signal can be used to cause the current in the third terminal to change from zero to a large value, thus allowing the device to act as a switch. As we shall see in Chapter 14, the switch is the basis for the realization of the logic inverter, the basic element of digital circuits. There are two major types of three-terminal semiconductor devices: the metal-oxidesemiconductor field-effect transistor (MOSFET), which is studied in this chapter, and the bipolar junction transistor (BJT), which we shall study in Chapter 6. Although each of the two transistor types offers unique features and areas of application, the MOSFET has become by far the most widely used electronic device, especially in the design of integrated circuits (ICs), which are entire circuits fabricated on a single silicon chip. Compared to BJTs, MOSFETs can be made quite small (i.e., requiring a small area on the silicon IC chip), and their manufacturing process is relatively simple (see Appendix A). Also, their operation requires comparatively little power. Furthermore, circuit designers have found ingenious ways to implement digital and analog functions utilizing MOSFETs almost exclusively (i.e., with very few or no resistors). All of these properties have made it possible to pack large numbers of MOSFETs (as many as 4 billion!) on a single IC chip to implement very sophisticated, very-large-scale-integrated (VLSI) digital circuits such as those for memory and microprocessors. Analog circuits such as amplifiers and filters can also be implemented in MOS technology, albeit in smaller, less-dense chips. Also, both analog and digital functions are increasingly being implemented on the same IC chip, in what is known as mixed-signal design. 247 248 Chapter 5 MOS Field-Effect Transistors (MOSFETs) The objective of this chapter is to develop in the reader a high degree of familiarity with the MOSFET: its physical structure and operation, terminal characteristics, and dc circuit applications. This will provide a solid foundation for the application of the MOSFET in amplifier design (Chapter 7) and in digital circuit design (Chapter 14). Although discrete MOS transistors exist, and the material studied in this chapter will enable the reader to design discrete MOS circuits, our study of the MOSFET is strongly influenced by the fact that most of its applications are in integrated-circuit design. The design of IC analog and digital MOS circuits occupies a large proportion of the remainder of this book. THE FIRST FIELD-EFFECT DEVICES: In 1925 a patent for solid-state electric-field-controlled conductor was filed in Canada by Julius E. Lilienfeld, a physicist at the University of Leipzig, Germany. Other patent refinements followed in the United States in 1926 and 1928. Regrettably, no research papers were published. Consequently, in 1934 Oskar Heil, a German physicist working at the University of Cambridge, U.K., filed a patent on a similar idea. But all these early concepts of electric-field control of a semiconducting path languished because suitable technology was not available. The invention of the bipolar transistor in 1947 at Bell Telephone Laboratories resulted in the speedy development of bipolar devices, a circumstance that further delayed the development of field-effect transistors. Although the field-effect device was described in a paper by William Shockley in 1952, it was not until 1960 that a patent on an insulated-gate field-effect device, the MOSFET, was filed by Dawon Kahng and Martin Atalla, also at Bell Labs. Clearly, the idea of field-effect control for amplification and switching has changed the world. With integrated-circuit chips today containing billions of MOS devices, MOS dominates the electronics world! 5.1 Device Structure and Physical Operation The enhancement-type MOSFET is the most widely used field-effect transistor. Except for the last section, this chapter is devoted to the study of the enhancement-type MOSFET. We begin in this section by learning about its structure and physical operation. This will lead to the current–voltage characteristics of the device, studied in the next section. 5.1.1 Device Structure Figure 5.1 shows the physical structure of the n-channel enhancement-type MOSFET. The meaning of the names “enhancement” and “n-channel” will become apparent shortly. The transistor is fabricated on a p-type substrate, which is a single-crystal silicon wafer that provides physical support for the device (and for the entire circuit in the case of an integrated circuit). Two heavily doped n-type regions, indicated in the figure as the n+ source1 and the n+ drain regions, are created in the substrate. A thin layer of silicon dioxide (SiO2) of thickness tox (typically 1 nm to 10 nm),2 which is an excellent electrical insulator, is grown on the surface of the substrate, covering the area between the source and drain regions. Metal is deposited on top of the oxide layer to form the gate electrode of the device. Metal contacts are 1The notation n+ indicates heavily doped n-type silicon. Conversely, n− is used to denote lightly doped n-type silicon. Similar notation applies for p-type silicon. 2A nanometer (nm) is 10−9 m or 0.001 μm. A micrometer (μm), or micron, is 10−6 m. Sometimes the oxide thickness is expressed in angstroms. An angstrom (Å) is 10−1 nm, or 10−10 m. W Source region nϩ 5.1 Device Structure and Physical Operation 249 S Metal G D Oxide (SiO2) p-type substrate (Body) L nϩ Channel region B Drain region Source (S) (a) Gate (G) Drain (D) Oxide (SiO2) (thickness = tox) Metal nϩ Channel region nϩ L p-type substrate (Body) Body (B) (b) Figure 5.1 Physical structure of the enhancement-type NMOS transistor: (a) perspective view; (b) cross section. Typically L = 0.03 μm to 1 μm, W = 0.05 μm to 100 μm, and the thickness of the oxide layer (tox) is in the range of 1 to 10 nm. also made to the source region, the drain region, and the substrate, also known as the body.3 Thus four terminals are brought out: the gate terminal (G), the source terminal (S), the drain terminal (D), and the substrate or body terminal (B). 3In Fig. 5.1, the contact to the body is shown on the bottom of the device. This will prove helpful in Section 5.4 in explaining a phenomenon known as the “body effect.” It is important to note, however, that in actual ICs, contact to the body is made at a location on the top of the device. 250 Chapter 5 MOS Field-Effect Transistors (MOSFETs) At this point it should be clear that the name of the device (metal-oxide-semiconductor FET) is derived from its physical structure. The name, however, has become a general one and is used also for FETs that do not use metal for the gate electrode. In fact, most modern MOSFETs are fabricated using a process known as silicon-gate technology, in which a certain type of silicon, called polysilicon, is used to form the gate electrode (see Appendix A). Our description of MOSFET operation and characteristics applies irrespective of the type of gate electrode. Another name for the MOSFET is the insulated-gate FET or IGFET. This name also arises from the physical structure of the device, emphasizing the fact that the gate electrode is electrically insulated from the device body (by the oxide layer). It is this insulation that causes the current in the gate terminal to be extremely small (of the order of 10−15 A). Observe that the substrate forms pn junctions with the source and drain regions. In normal operation these pn junctions are kept reverse biased at all times. Since, as we shall see shortly, the drain will always be at a positive voltage relative to the source, the two pn junctions can be effectively cut off by simply connecting the substrate terminal to the source terminal. We shall assume this to be the case in the following description of MOSFET operation. Thus, here, the substrate will be considered as having no effect on device operation, and the MOSFET will be treated as a three-terminal device, with the terminals being the gate (G), the source (S), and the drain (D). It will be shown that a voltage applied to the gate controls current flow between source and drain. This current will flow in the longitudinal direction from drain to source in the region labeled “channel region.” Note that this region has a length L and a width W, two important parameters of the MOSFET. Typically, L is in the range of 0.03 μm to 1 μm, and W is in the range of 0.05 μm to 100 μm. Finally, note that the MOSFET is a symmetrical device; thus its source and drain can be interchanged with no change in device characteristics. 5.1.2 Operation with Zero Gate Voltage With zero voltage applied to the gate, two back-to-back diodes exist in series between drain and source. One diode is formed by the pn junction between the n+ drain region and the p-type substrate, and the other diode is formed by the pn junction between the p-type substrate and the n+ source region. These back-to-back diodes prevent current conduction from drain to source when a voltage vDS is applied. In fact, the path between drain and source has a very high resistance (of the order of 1012 ). 5.1.3 Creating a Channel for Current Flow Consider next the situation depicted in Fig. 5.2. Here we have grounded the source and the drain and applied a positive voltage to the gate. Since the source is grounded, the gate voltage appears in effect between gate and source and thus is denoted vGS. The positive voltage on the gate causes, in the first instance, the free holes (which are positively charged) to be repelled from the region of the substrate under the gate (the channel region). These holes are pushed downward into the substrate, leaving behind a carrier-depletion region. The depletion region is populated by the bound negative charge associated with the acceptor atoms. These charges are “uncovered” because the neutralizing holes have been pushed downward into the substrate. As well, the positive gate voltage attracts electrons from the n+ source and drain regions (where they are in abundance) into the channel region. When a sufficient number of electrons accumulate near the surface of the substrate under the gate, an n region is in effect created, connecting the source and drain regions, as indicated in Fig. 5.2. Now if a voltage is applied between drain and source, current flows through this induced n region, carried by the mobile ϩ vGS Ϫ S Oxide (SiO2) 5.1 Device Structure and Physical Operation 251 Gate electrode Induced G n-type D channel nϩ L nϩ Depletion region p-type substrate B Figure 5.2 The enhancement-type NMOS transistor with a positive voltage applied to the gate. An n channel is induced at the top of the substrate beneath the gate. electrons. The induced n region thus forms a channel for current flow from drain to source and is aptly called so. Correspondingly, the MOSFET of Fig. 5.2 is called an n-channel MOSFET or, alternatively, an NMOS transistor. Note that an n-channel MOSFET is formed in a p-type substrate: The channel is created by inverting the substrate surface from p type to n type. Hence the induced channel is also called an inversion layer. The value of vGS at which a sufficient number of mobile electrons accumulate in the channel region to form a conducting channel is called the threshold voltage and is denoted Vt.4 Obviously, Vt for an n-channel FET is positive. The value of Vt is controlled during device fabrication and typically lies in the range of 0.3 V to 1.0 V. The gate and the channel region of the MOSFET form a parallel-plate capacitor, with the oxide layer acting as the capacitor dielectric. The positive gate voltage causes positive charge to accumulate on the top plate of the capacitor (the gate electrode). The corresponding negative charge on the bottom plate is formed by the electrons in the induced channel. An electric field thus develops in the vertical direction. It is this field that controls the amount of charge in the channel, and thus it determines the channel conductivity and, in turn, the current that will flow through the channel when a voltage vDS is applied. This is the origin of the name “field-effect transistor” (FET). The voltage across this parallel-plate capacitor, that is, the voltage across the oxide, must exceed Vt for a channel to form. When vDS = 0, as in Fig. 5.2, the voltage at every point along the channel is zero, and the voltage across the oxide (i.e., between the gate and the points along the channel) is uniform and equal to vGS. The excess of vGS over Vt is termed the effective voltage or the overdrive voltage and is the quantity that determines the charge in the channel. In this book, we shall denote (vGS − Vt) by vOV , vGS − Vt ≡ vOV (5.1) 4Some texts use VT to denote the threshold voltage. We use Vt to avoid confusion with the thermal voltage VT . 252 Chapter 5 MOS Field-Effect Transistors (MOSFETs) We can express the magnitude of the electron charge in the channel by |Q| = Cox(WL)vOV (5.2) where Cox, called the oxide capacitance, is the capacitance of the parallel-plate capacitor per unit gate area (in units of F/m2), W is the width of the channel, and L is the length of the channel. The oxide capacitance Cox is given by Cox = eox tox (5.3) where eox is the permittivity of the silicon dioxide, eox = 3.9e0 = 3.9 × 8.854 × 10−12 = 3.45 × 10−11 F/m The oxide thickness tox is determined by the process technology used to fabricate the MOSFET. As an example, for a process with tox = 4 nm, Cox = 3.45 × 10−11 4 × 10−9 = 8.6 × 10−3 F/m2 It is much more convenient to express Cox per micron squared. For our example, this yields 8.6 fF/μm2, where fF denotes femtofarad (10−15 F). For a MOSFET fabricated in this technology with a channel length L = 0.18 μm and a channel width W = 0.72 μm, the total capacitance between gate and channel is C = CoxWL = 8.6 × 0.18 × 0.72 = 1.1 fF Finally, note from Eq. (5.2) that as vOV is increased, the magnitude of the channel charge increases proportionately. Sometimes this is depicted as an increase in the depth of the channel; that is, the larger the overdrive voltage, the deeper the channel. 5.1.4 Applying a Small vDS Having induced a channel, we now apply a positive voltage vDS between drain and source, as shown in Fig. 5.3. We first consider the case where vDS is small (i.e., 50 mV or so). The voltage vDS causes a current iD to flow through the induced n channel. Current is carried by free electrons traveling from source to drain (hence the names source and drain). By convention, the direction of current flow is opposite to that of the flow of negative charge. Thus the current in the channel, iD, will be from drain to source, as indicated in Fig. 5.3. We now wish to calculate the value of iD. Toward that end, we first note that because vDS is small, we can continue to assume that the voltage between the gate and various points along the channel remains approximately constant and equal to the value at the source end, vGS. Thus, the effective voltage between the gate and the various points along the channel remains equal to vOV , and the channel charge Q is still given by Eq. (5.2). Of particular interest 5.1 Device Structure and Physical Operation 253 n+ iD n+ Figure 5.3 An NMOS transistor with vGS > Vt and with a small vDS applied. The device acts as a resistance whose value is determined by vGS. Specifically, the channel conductance is proportional to vGS – Vt, and thus iD is proportional to (vGS – Vt)vDS. Note that the depletion region is not shown (for simplicity). in calculating the current iD is the charge per unit channel length, which can be found from Eq. (5.2) as unit |Q| channel length = CoxW vOV (5.4) The voltage vDS establishes an electric field E across the length of the channel, |E| = vDS L (5.5) This electric field in turn causes the channel electrons to drift toward the drain with a velocity given by Electron drift velocity = μn |E| = μn v DS L (5.6) where μn is the mobility of the electrons at the surface of the channel. It is a physical parameter whose value depends on the fabrication process technology. The value of iD can now be found by multiplying the charge per unit channel length (Eq. 5.4) by the electron drift velocity (Eq. 5.6), W iD = (μnCox) L vOV vDS (5.7) 254 Chapter 5 MOS Field-Effect Transistors (MOSFETs) Thus, for small vDS, the channel behaves as a linear resistance whose value is controlled by the overdrive voltage vOV , which in turn is determined by vGS: W iD = (μnCox) L (vGS − Vt) vDS (5.8) The conductance gDS of the channel can be found from Eq. (5.7) or (5.8) as W gDS = (μnCox) L vOV (5.9) or gDS = (μnCox) W L (vGS − Vt) (5.10) Observe that the conductance is determined by the product of three factors: (μnCox), (W/L), and vOV (or equivalently, vGS − Vt). To gain insight into MOSFET operation, we consider each of the three factors in turn. The first factor, (μnCox), is determined by the process technology used to fabricate the MOSFET. It is the product of the electron mobility, μn, and the oxide capacitance, Cox. It makes physical sense for the channel conductance to be proportional to each of μn and Cox (why?) and hence to their product, which is termed the process transconductance parameter5 and given the symbol kn, where the subscript n denotes n channel, kn = μnCox (5.11) It can be shown that with μn having the dimensions of meters squared per volt-second (m2/V · s) and Cox having the dimensions of farads per meter squared (F/m2), the dimensions of kn are amperes per volt squared (A/V2). The second factor in the expression for the conductance gDS in Eqs. (5.9) and (5.10) is the transistor aspect ratio (W/L). That the channel conductance is proportional to the channel width W and inversely proportional to the channel length L should make perfect physical sense. The (W/L) ratio is obviously a dimensionless quantity that is determined by the device designer. Indeed, the values of W and L can be selected by the device designer to give the device the i−v characteristics desired. For a given fabrication process, however, there is a minimum channel length, Lmin. In fact, the minimum channel length that is possible with a given fabrication process is used to characterize the process and is being continually reduced as technology advances. For instance, in 2014 the state-of-the-art in commercially available MOS technology was a 32-nm process, meaning that for this process the minimum channel length possible was 32 nm. Finally, we should note that the oxide thickness tox scales down with Lmin. Thus, for a 0.13-μm technology, tox is 2.7 nm, but for the currently popular 65-nm technology, tox is about 2.2 nm. 5This name arises from the fact that μnCox determines the transconductance of the MOSFET, as will be seen shortly. 5.1 Device Structure and Physical Operation 255 The product of the process transconductance parameter kn and the transistor aspect ratio (W/L) is the MOSFET transconductance parameter kn, kn = kn(W/L) (5.12a) or kn = (μnCox)(W/L) (5.12b) The MOSFET parameter kn has the dimensions of A/V2. The third term in the expression of the channel conductance gDS is the overdrive voltage vOV . This is hardly surprising, since vOV directly determines the magnitude of electron charge in the channel. As will be seen, vOV is a very important circuit-design parameter. In this book, we will use vOV and vGS−Vt interchangeably. We conclude this subsection by noting that with vDS kept small, the MOSFET behaves as a linear resistance rDS whose value is controlled by the gate voltage vGS, 1 rDS = gDS 1 rDS = (μnCox)(W/L)vOV (5.13a) rDS = 1 (μn Cox )(W/L)(v GS − Vt) (5.13b) The operation of the MOSFET as a voltage-controlled resistance is further illustrated in Fig. 5.4, which is a sketch of iD versus vDS for various values of vGS. Observe that the resistance is infinite for vGS ≤ Vt and decreases as vGS is increased above Vt. It is interesting to note that although vGS is used as the parameter for the set of graphs in Fig. 5.4, the graphs in fact depend only on vOV (and, of course, kn). The description above indicates that for the MOSFET to conduct, a channel has to be induced. Then, increasing vGS above the threshold voltage Vt enhances the channel, hence the names enhancement-mode operation and enhancement-type MOSFET. Finally, we note that the current that leaves the source terminal (iS) is equal to the current that enters the drain terminal (iD), and the gate current iG = 0. iD vGS 0 Slope ϵ gDS = knVOV vGS ϭ Vt + VOV3 vGS ϭ Vt + VOV2 vGS ϭ Vt + VOV1 vGS Յ Vt vDS Figure 5.4 The iD–vDS characteristics of the MOSFET in Fig. 5.3 when the voltage applied between drain and source, vDS, is kept small. The device operates as a linear resistance whose value is controlled by vGS. 256 Chapter 5 MOS Field-Effect Transistors (MOSFETs) EXERCISE 5.1 A 0.18-μm fabrication process is specified to have tox = 4 nm, μn = 450 cm2/V · s, and Vt = 0.5 V. Find the value of the process transconductance parameter kn. For a MOSFET with minimum length fabricated in this process, find the required value of W so that the device exhibits a channel resistance rDS of 1 k at vGS = 1 V. Ans. 388 μA/V2; 0.93 μm 5.1.5 Operation as vDS Is Increased We next consider the situation as vDS is increased. For this purpose, let vGS be held constant at a value greater than Vt; that is, let the MOSFET be operated at a constant overdrive voltage VOV . Refer to Fig. 5.5, and note that vDS appears as a voltage drop across the length of the channel. That is, as we travel along the channel from source to drain, the voltage (measured relative to the source) increases from zero to vDS. Thus the voltage between the gate and points along the channel decreases from vGS = Vt + VOV at the source end to vGD = vGS − vDS = Vt + VOV − vDS at the drain end. Since the channel depth depends on this voltage, and specifically on the amount by which this voltage exceeds Vt, we find that the channel is no longer of uniform depth; rather, the channel will take the tapered shape shown in Fig. 5.5, being deepest at the source end (where the depth is proportional to VOV ) and shallowest at the drain end6 (where the depth is proportional to VOV − vDS). This point is further illustrated in Fig. 5.6. As vDS is increased, the channel becomes more tapered and its resistance increases correspondingly. Thus, the iD−vDS curve does not continue as a straight line but bends as shown in Fig. 5.7. The equation describing this portion of the iD−vDS curve can be easily derived by utilizing the information in Fig. 5.6. Specifically, note that the charge in the tapered channel is proportional to the channel cross-sectional area shown in Fig. 5.6(b). This area in turn can be easily seen as proportional to 1 2 [VOV +(VOV −v DS )] or VOV − 1 2 v DS . Thus, the relationship between iD and vDS can be found by replacing VOV in Eq. (5.7) by VOV − 1 2 v DS , iD = kn W L VOV − 1 2 v DS v DS (5.14) This relationship describes the semiparabolic portion of the iD−vDS curve in Fig. 5.7. It applies to the entire segment down to vDS = 0. Specifically, note that as vDS is reduced, we can neglect 1 2 v DS relative to VOV in the factor in parentheses, and the expression reduces to that in Eq. (5.7). The latter of course is an approximation and applies only for small vDS (i.e., near the origin). There is another useful interpretation of the expression in Eq. (5.14). From Fig. 5.6(a) we see that the average voltage along the channel is 1 2 v DS . Thus, the average voltage that gives rise to channel charge and hence to iD is no longer VOV but VOV − 1 2 v DS , which is indeed the factor that appears in Eq. (5.14). Finally, we note that Eq. (5.14) is frequently written in the 6For simplicity, we do not show in Fig. 5.5 the depletion region. Physically speaking, it is the widening of the depletion region as a result of the increased vDS that makes the channel shallower near the drain. 5.1 Device Structure and Physical Operation 257 nϩ nϩ Figure 5.5 Operation of the enhancement NMOS transistor as vDS is increased. The induced channel acquires a tapered shape, and its resistance increases as vDS is increased. Here, vGS is kept constant at a value > Vt; vGS = Vt + VOV . Voltage VGS Vt (VOV Ϫ 1 2 vDS) VOV Source 0 L Voltage drop 2 along the channel (a) Average = 1 2 vDS vGD vDS L Drain x ␣VOV Source Channel (b) ␣(VOV Ϫ vDS) Drain Figure 5.6 (a) For a MOSFET with vGS = Vt + VOV , application of vDS causes the voltage drop along the channel to vary linearly, with an average value of 1 2 v DS at the midpoint. Since v GD > Vt , the channel still exists at the drain end. (b) The channel shape corresponding to the situation in (a). While the depth of the channel at the source end is still proportional to VOV , that at the drain end is proportional to (VOV −vDS). 258 Chapter 5 MOS Field-Effect Transistors (MOSFETs) iD Curve bends because the channel resistance increases with vDS Almost a straight line with slope proportional to VOV Triode (vDS ≤ VOV) Saturation (vDS ≥ VOV) Current saturates because the channel is pinched off at the drain end, and vDS no longer affects the channel. vGS = Vt + VOV 0 VDS sat = VOV vDS Figure 5.7 The drain current iD versus the drain-to-source voltage vDS for an enhancement-type NMOS transistor operated with vGS = Vt + VOV . alternate form W iD = kn L VOV vDS − 1 2 v 2 DS (5.15) Furthermore, for an arbitrary value of VOV , we can replace VOV by (vGS−Vt) and rewrite Eq. (5.15) as W iD = kn L (v GS − Vt )v DS − 1 2 v 2 DS (5.16) 5.1.6 Operation for vDS ≥ VOV: Channel Pinch-Off and Current Saturation The above description of operation assumed that even though the channel became tapered, it still had a finite (nonzero) depth at the drain end. This in turn is achieved by keeping vDS sufficiently small that the voltage between the gate and the drain, vGD, exceeds Vt. This is indeed the situation shown in Fig. 5.6(a). Note that for this situation to obtain, vDS must not exceed VOV , for as vDS = VOV , vGD = Vt, and the channel depth at the drain end reduces to zero. Figure 5.8 shows vDS reaching VOV and vGD correspondingly reaching Vt. The zero depth of the channel at the drain end gives rise to the term channel pinch-off. Increasing vDS beyond this value (i.e., vDS > VOV ) has no effect on the channel shape and charge, and the current through the channel remains constant at the value reached for vDS = VOV . The drain current thus saturates at the value found by substituting vDS = VOV in Eq. (5.14), iD = 1 2 kn W L VO2V (5.17) Voltage 5.1 Device Structure and Physical Operation 259 VGS Vt VOV 1 2 VOV Source 0 L Voltage drop 2 along the channel (a) Average = 1 2 VOV vGD = Vt vDS = VOV L Drain x Source Channel Drain (b) Figure 5.8 Operation of MOSFET with vGS = Vt + VOV , as vDS is increased to VOV . At the drain end, vGD decreases to Vt and the channel depth at the drain end reduces to zero (pinch-off). At this point, the MOSFET enters the saturation mode of operation. Further increasing vDS (beyond VDSsat = VOV ) has no effect on the channel shape and iD remains constant. The MOSFET is then said to have entered the saturation region (or, equivalently, the saturation mode of operation). The voltage vDS at which saturation occurs is denoted VDSsat, VDSsat = VOV = VGS − Vt (5.18) It should be noted that channel pinch-off does not mean channel blockage: Current continues to flow through the pinched-off channel, and the electrons that reach the drain end of the channel are accelerated through the depletion region that exists there (not shown in Fig. 5.5) and into the drain terminal. Any increase in vDS above VDSsat appears as a voltage drop across the depletion region. Thus, both the current through the channel and the voltage drop across it remain constant in saturation. The saturation portion of the iD − vDS curve is, as expected, a horizontal straight line, as indicated in Fig. 5.7. Also indicated in Fig. 5.7 is the name of the region of operation obtained with a continuous (non-pinched-off) channel, the triode region. This name is a carryover from the days of vacuum-tube devices, whose operation a FET resembles. Finally, we note that the expression for iD in saturation can be generalized by replacing the constant overdrive voltage VOV by a variable one, vOV : iD = 1 2 kn W L v 2 OV (5.19) 260 Chapter 5 MOS Field-Effect Transistors (MOSFETs) Also, vOV can be replaced by (vGS−Vt) to obtain the alternate expression for saturationmode iD, iD = 1 2 kn W L (vGS − Vt)2 (5.20) Example 5.1 Consider a process technology for which Lmin = 0.4 μm, tox = 8 nm, μn = 450 cm2/V · s, and Vt = 0.7 V. (a) Find Cox and kn. (b) For a MOSFET with W/L = 8 μm/0.8 μm, calculate the values of VOV , VGS, and VDSmin needed to operate the transistor in the saturation region with a dc current ID = 100 μA. (c) For the device in (b), find the values of VOV and VGS required to cause the device to operate as a 1000- resistor for very small vDS. Solution (a) Cox = eox tox = 3.45 × 10−11 8 × 10−9 = 4.32 × 10−3 F/m2 = 4.32 fF/μm2 kn = μnCox = 450 (cm2/V · s) × 4.32 (fF/μm2) = 450 × 108 (μm2/V · s) × 4.32 × 10−15 (F/μm2) = 194 × 10−6 (F/V · s) = 194 μA/V2 (b) For operation in the saturation region, Thus, which results in iD = 1 2 kn W L VO2V 100 = 1 2 × 194 × 8 0.8 VO2V Thus, VOV = 0.32 V VGS = Vt + VOV = 1.02 V and VDSmin = VOV = 0.32 V 5.1 Device Structure and Physical Operation 261 (c) For the MOSFET in the triode region with vDS very small, rDS = 1 kn W L VOV Thus 1 1000 = 194 × 10−6 × 10 × VOV which yields Thus, VOV = 0.52 V VGS = 1.22 V EXERCISES 5.2 For a 0.18-μm process technology for which tox = 4 nm and μn = 450 cm2/V · s, find Cox, kn, and the overdrive voltage VOV required to operate a transistor having W/L = 20 in saturation with ID = 0.3 mA. What is the minimum value of VDS needed? Ans. 8.6 fF/μm2; 387 μA/V2; 0.28 V; 0.28 V D5.3 A circuit designer intending to operate a MOSFET in saturation is considering the effect of changing the device dimensions and operating voltages on the drain current ID. Specifically, by what factor does ID change in each of the following cases? (a) The channel length is doubled. (b) The channel width is doubled. (c) The overdrive voltage is doubled. (d) The drain-to-source voltage is doubled. (e) Changes (a), (b), (c), and (d) are made simultaneously. Which of these cases might cause the MOSFET to leave the saturation region? Ans. 0.5; 2; 4; no change; 4; case (c) if vDS is smaller than 2VOV 5.1.7 The p-Channel MOSFET Figure 5.9(a) shows a cross-sectional view of a p-channel enhancement-type MOSFET. The structure is similar to that of the NMOS device except that here the substrate is n type and the source and the drain regions are p+ type; that is, all semiconductor regions are reversed in polarity relative to their counterparts in the NMOS case. The PMOS and NMOS transistors are said to be complementary devices. 262 Chapter 5 MOS Field-Effect Transistors (MOSFETs) S G D p+ p+ n-type substrate S iD p+ B (a) ϩ vGS Ϫ G iG = 0 iD induced p channel n-type substrate ϩ vDS Ϫ iD D p+ B (b) Figure 5.9 (a) Physical structure of the PMOS transistor. Note that it is similar to the NMOS transistor shown in Fig. 5.1(b) except that all semiconductor regions are reversed in polarity. (b) A negative voltage vGS of magnitude greater than Vtp induces a p channel, and a negative vDS causes a current iD to flow from source to drain. To induce a channel for current flow between source and drain, a negative voltage is applied to the gate, that is, between gate and source, as indicated in Fig. 5.9(b). By increasing the magnitude of the negative vGS beyond the magnitude of the threshold voltage Vtp, which by convention is negative, a p channel is established as shown in Fig. 5.9(b). This condition can be described as vGS ≤ Vtp or, to avoid dealing with negative signs, |vGS| ≥ Vtp 5.1 Device Structure and Physical Operation 263 Now, to cause a current iD to flow in the p channel, a negative voltage vDS is applied to the drain.7 The current iD is carried by holes and flows through the channel from source to drain. As we have done for the NMOS transistor, we define the process transconductance parameter for the PMOS device as kp = μpCox where μp is the mobility of the holes in the induced p channel. Typically, μp = 0.25 μn to 0.5 μn and is process-technology dependent. The transistor transconductance parameter kp is obtained by multiplying kp by the aspect ratio W/L, kp = kp(W/L) The remainder of the description of the physical operation of the p-channel MOSFET follows that for the NMOS device, except of course for the sign reversals of all voltages. We will present the complete current–voltage characteristics of both NMOS and PMOS transistors in the next section. PMOS technology originally dominated MOS integrated-circuit manufacturing, and the original microprocessors utilized PMOS transistors. As the technological difficulties of fabricating NMOS transistors were solved, NMOS completely supplanted PMOS. The main reason for this change is that electron mobility μn is higher by a factor of 2 to 4 than the hole mobility μp, resulting in NMOS transistors having greater gains and speeds of operation than PMOS devices. Subsequently, a technology was developed that permits the fabrication of both NMOS and PMOS transistors on the same chip. Appropriately called complementary MOS, or CMOS, this technology is currently the dominant electronics technology. 5.1.8 Complementary MOS or CMOS As the name implies, complementary MOS technology employs MOS transistors of both polarities. Although CMOS circuits are somewhat more difficult to fabricate than NMOS, the availability of complementary devices makes possible many powerful circuit configurations. Indeed, at the present time CMOS is the most widely used of all the IC technologies. This statement applies to both analog and digital circuits. CMOS technology has virtually replaced designs based on NMOS transistors alone. Furthermore, by 2014 CMOS technology had taken over many applications that just a few years earlier were possible only with bipolar devices. Throughout this book, we will study many CMOS circuit techniques. Figure 5.10 shows a cross section of a CMOS chip illustrating how the PMOS and NMOS transistors are fabricated. Observe that while the NMOS transistor is implemented directly in the p-type substrate, the PMOS transistor is fabricated in a specially created n region, known as an n well. The two devices are isolated from each other by a thick region of oxide that functions as an insulator. Not shown on the diagram are the connections made to the p-type body and to the n well. The latter connection serves as the body terminal for the PMOS transistor. 7If a positive voltage is applied to the drain, the pn junction between the drain region and the substrate will become forward biased, and the device will no longer operate as a MOSFET. Proper MOSFET operation is predicated on the pn junctions between the source and drain regions and the substrate being always reverse biased. 264 Chapter 5 MOS Field-Effect Transistors (MOSFETs) NMOS S G D D Gate oxide Polysilicon SiO2 Thick SiO2 (isolation) nϩ nϩ pϩ p-type body PMOS G n well S SiO2 pϩ Figure 5.10 Cross section of a CMOS integrated circuit. Note that the PMOS transistor is formed in a separate n-type region, known as an n well. Another arrangement is also possible in which an n-type substrate (body) is used and the n device is formed in a p well. Not shown are the connections made to the p-type body and to the n well; the latter functions as the body terminal for the p-channel device. 5.1.9 Operating the MOS Transistor in the Subthreshold Region The above description of the n-channel MOSFET operation implies that for vGS < Vt, no current flows and the device is cut off. This is not entirely true, for it has been found that for values of vGS smaller than but close to Vt, a small drain current flows. In this subthreshold region of operation, the drain current is exponentially related to vGS, much like the iC–vBE relationship of a BJT, as will be shown in the next chapter. Although in most applications the MOS transistor is operated with vGS > Vt, there are special, but a growing number of, applications that make use of subthreshold operation. In Chapter 14, we will briefly consider subthreshold operation. 5.2 Current–Voltage Characteristics Building on the physical foundation established in the previous section for the operation of the enhancement MOS transistor, in this section we present its complete current–voltage characteristics. These characteristics can be measured at dc or at low frequencies and thus are called static characteristics. The dynamic effects that limit the operation of the MOSFET at high frequencies and high switching speeds will be discussed in Chapter 10. 5.2.1 Circuit Symbol Figure 5.11(a) shows the circuit symbol for the n-channel enhancement-type MOSFET. Observe that the spacing between the two vertical lines that represent the gate and the channel indicates the fact that the gate electrode is insulated from the body of the device. The polarity of the p-type substrate (body) and the n channel is indicated by the arrowhead on the line representing the body (B). This arrowhead also indicates the polarity of the transistor, namely, that it is an n-channel device. Although the MOSFET is a symmetrical device, it is often useful in circuit design to designate one terminal as the source and the other as the drain (without having to write S and 5.2 Current–Voltage Characteristics 265 D D D G BG BG S (a) S (b) S (c) Figure 5.11 (a) Circuit symbol for the n-channel enhancement-type MOSFET. (b) Modified circuit symbol with an arrowhead on the source terminal to distinguish it from the drain and to indicate device polarity (i.e., n channel). (c) Simplified circuit symbol to be used when the source is connected to the body or when the effect of the body on device operation is unimportant. D beside the terminals). This objective is achieved in the modified circuit symbol shown in Fig. 5.11(b). Here an arrowhead is placed on the source terminal, thus distinguishing it from the drain terminal. The arrowhead points in the normal direction of current flow and thus indicates the polarity of the device (i.e., n channel). Observe that in the modified symbol, there is no need to show the arrowhead on the body line. Although the circuit symbol of Fig. 5.11(b) clearly distinguishes the source from the drain, in practice it is the polarity of the voltage impressed across the device that determines source and drain; the drain is always positive relative to the source in an n-channel FET. In applications where the source is connected to the body of the device, a further simplification of the circuit symbol is possible, as indicated in Fig. 5.11(c). This symbol is also used in applications when the effect of the body on circuit operation is not important, as will be seen later. 5.2.2 The iD–vDS Characteristics Table 5.1 provides a compilation of the conditions and the formulas for the operation of the NMOS transistor in each of the three possible regions: the cutoff region, the triode region, and the saturation region. The first two are useful if the MOSFET is to be utilized as a switch. On the other hand, if the MOSFET is to be used to design an amplifier, it must be operated in the saturation region. The rationale for these choices will be addressed in Chapter 7. At the top of Table 5.1 we show a circuit consisting of an NMOS transistor and two dc supplies providing vGS and vDS. This conceptual circuit can be used to measure the iD–vDS characteristic curves of the NMOS transistor. Each curve is measured by setting vGS to a desired constant value, varying vDS, and measuring the corresponding iD. Two of these characteristic curves are shown in the accompanying diagram: one for vGS < Vtn and the other for vGS = Vtn + vOV . (Note that we now use Vtn to denote the threshold voltage of the NMOS transistor, to distinguish it from that of the PMOS transistor, denoted Vtp.) As Table 5.1 shows, the boundary between the triode region and the saturation region is determined by whether vDS is less or greater than the overdrive voltage vOV at which the transistor is operating. An equivalent way to check for the region of operation is to examine the relative values of the drain and gate voltages. To operate in the triode region, the gate voltage must exceed the drain voltage by at least Vtn volts, which ensures that the channel remains continuous (not pinched off). On the other hand, to operate in saturation, the channel must be 266 Chapter 5 MOS Field-Effect Transistors (MOSFETs) Table 5.1 Regions of Operation of the Enhancement NMOS Transistor pinched off at the drain end; pinch-off is achieved here by keeping vD higher than vG−Vtn, that is, not allowing vD to fall below vG by more than Vtn volts. The graphical construction of Fig. 5.12 should serve to remind the reader of these conditions. A set of iD−vDS characteristics for the NMOS transistor is shown in Fig. 5.13. Observe that each graph is obtained by setting vGS above Vtn by a specific value of overdrive voltage, denoted VOV1, VOV2, VOV3, and VOV4. This in turn is the value of vDS at which the corresponding graph saturates, and the value of the resulting saturation current is directly determined by the value of vOV , namely, 1 2 kn VO2V 1 , 1 2 kn VO2V 2, . . . The reader is advised to commit to memory both the structure of these graphs and the coordinates of the saturation points. Finally, observe that the boundary between the triode and the saturation regions, that is, the locus of the saturation points, is a parabolic curve described by 1 iD = 2 kn W L v 2 DS 5.2 Current–Voltage Characteristics 267 G ϩ Vtn Saturation Vtn ϩ VOV Ϫ VOV D Triode Figure 5.12 The relative levels of the terminal voltages of the enhancement NMOS transistor for operation in the triode region and in the S saturation region. Figure 5.13 The iD−vDS characteristics for an enhancement-type NMOS transistor. 5.2.3 The iD–vGS Characteristic When the MOSFET is used to design an amplifier, it is operated in the saturation region. As Fig. 5.13 indicates, in saturation the drain current is constant determined by vGS (or vOV ) and is independent of vDS. That is, the MOSFET operates as a constant-current source where the value of the current is determined by vGS. In effect, then, the MOSFET operates as a voltage-controlled current source with the control relationship described by or in terms of vOV , iD = 1 2 kn W L (vGS − Vtn)2 iD = 1 2 kn W L v 2 OV (5.21) (5.22) 268 Chapter 5 MOS Field-Effect Transistors (MOSFETs) This is the relationship that underlies the application of the MOSFET as an amplifier. That it is nonlinear should be of concern to those interested in designing linear amplifiers. Nevertheless, in Chapter 7, we will see how one can obtain linear amplification from this nonlinear control or transfer characteristic. Figure 5.14 shows the iD–vGS characteristic of an NMOS transistor operating in saturation. Note that if we are interested in a plot of iD versus vOV , we simply shift the origin to the point vGS = Vtn. The view of the MOSFET in the saturation region as a voltage-controlled current source is illustrated by the equivalent-circuit representation shown in Fig. 5.15. For reasons that will become apparent shortly, the circuit in Fig. 5.15 is known as a large-signal equivalent circuit. Note that the current source is ideal, with an infinite output resistance representing the independence, in saturation, of iD from vDS. This, of course, has been assumed in the idealized model of device operation utilized thus far. We are about to rectify an important shortcoming of this model. First, however, we present an example. iD vDS ≥ vGS – Vtn 0 Vtn 0 vGS Figure 5.14 The iD–vGS characteristic of an NMOS transistor operating in the satura- tion region. The iD–vOV characteristic can be obtained by simply relabeling the horizontal vOV axis, that is, shifting the origin to the point vGS = Vtn. kЈ Vtn)2 tn tn Figure 5.15 Large-signal, equivalent-circuit model of an n-channel MOSFET operating in the saturation region. 5.2 Current–Voltage Characteristics 269 Example 5.2 Consider an NMOS transistor fabricated in a 0.18-μm process with L = 0.18 μm and W = 2 μm. The process technology is specified to have Cox = 8.6 fF/μm2, μn = 450 cm2/V · s, and Vtn = 0.5 V. (a) Find VGS and VDS that result in the MOSFET operating at the edge of saturation with ID = 100 μA. (b) If VGS is kept constant, find VDS that results in ID = 50 μA. (c) To investigate the use of the MOSFET as a linear amplifier, let it be operating in saturation with VDS = 0.3 V. Find the change in iD resulting from vGS changing from 0.7 V by +0.01 V and by −0.01 V. Solution First we determine the process transconductance parameter kn, kn = μnCox = 450 × 10−4 × 8.6 × 10−15 × 1012 A/V2 = 387 μA/V2 and the transistor transconductance parameter kn, kn = kn W L = 387 2 0.18 = 4.3 mA/V2 (a) With the transistor operating in saturation, Thus, which results in ID = 1 2 kn VO2V 100 = 1 2 × 4.3 × 103 × VO2V Thus, VOV = 0.22 V VGS = Vtn + VOV = 0.5 + 0.22 = 0.72 V and since operation is at the edge of saturation, VDS = VOV = 0.22 V 270 Chapter 5 MOS Field-Effect Transistors (MOSFETs) Example 5.2 continued (b) With VGS kept constant at 0.72 V and ID reduced from the value obtained at the edge of saturation, the MOSFET will now be operating in the triode region, thus ID = kn VOV VDS − 1 2 VD2S 50 = 4.3 × 103 0.22VDS − 1 2 VD2S which can be rearranged to the form VD2S − 0.44VDS + 0.023 = 0 This quadratic equation has two solutions VDS = 0.06 V and VDS = 0.39 V The second answer is greater than VOV and thus is physically meaningless, since we know that the transistor is operating in the triode region. Thus we have VDS = 0.06 V (c) For vGS = 0.7 V, VOV = 0.2 V, and since VDS = 0.3 V, the transistor is operating in saturation and ID = 1 2 kn VO2V = 1 × 4300 × 0.04 2 = 86 μA Now for vGS = 0.710 V, vOV = 0.21 V and iD = 1 2 × 4300 × 0.212 = 94.8 μA and for vGS = 0.690 V, vOV = 0.19 V, and iD = 1 2 × 4300 × 0.192 = 77.6 μA Thus, with vGS = +0.01 V, iD = 8.8 μA; and for vGS = −0.01 V, iD = −8.4 μA. We conclude that the two changes are almost equal, an indication of almost-linear operation when the changes in vGS are kept small. This is just a preview of the “small-signal operation” of the MOSFET studied in Chapter 7. 5.2 Current–Voltage Characteristics 271 EXERCISES 5.4 An NMOS transistor is operating at the edge of saturation with an overdrive voltage VOV and a drain current ID. If VOV is doubled, and we must maintain operation at the edge of saturation, what should VDS be changed to? What value of drain current results? Ans. 2VOV ; 4ID 5.5 An n-channel MOSFET operating with VOV = 0.5 V exhibits a linear resistance rDS = 1 k when vDS is very small. What is the value of the device transconductance parameter kn? What is the value of the current ID obtained when vDS is increased to 0.5 V? and to 1 V? Ans. 2 mA/V2; 0.25 mA; 0.25 mA 5.2.4 Finite Output Resistance in Saturation Equation (5.21) and the corresponding large-signal equivalent circuit in Fig. 5.15, as well as the graphs in Fig. 5.13, indicate that in saturation, iD is independent of vDS. Thus, a change vDS in the drain-to-source voltage causes a zero change in iD, which implies that the incremental resistance looking into the drain of a saturated MOSFET is infinite. This, however, is an idealization based on the premise that once the channel is pinched off at the drain end, further increases in vDS have no effect on the channel’s shape. But, in practice, increasing vDS beyond vOV does affect the channel somewhat. Specifically, as vDS is increased, the channel pinch-off point is moved slightly away from the drain, toward the source. This is illustrated in Fig. 5.16, from which we note that the voltage across the channel remains constant at vOV , and the additional voltage applied to the drain appears as a voltage drop across the narrow depletion region between the end of the channel and the drain region. This voltage accelerates the electrons that reach the drain end of the channel and sweeps them across the depletion region into the drain. Note, however, that (with depletion-layer widening) the channel length is in effect reduced, from L to L − L, a phenomenon known as channel-length modulation. Now, since iD is inversely proportional to the channel length (Eq. 5.21), iD increases with vDS. Source Channel Drain Ϫ vOV LϪ ϩ Ϫ ϩ vDS Ϫ vOV L Figure 5.16 Increasing vDS beyond vDSsat causes the channel pinch-off point to move slightly away from the drain, thus reducing the effective channel length (by L). 272 Chapter 5 MOS Field-Effect Transistors (MOSFETs) This effect can be accounted for in the expression for iD by including a factor 1 + λ(vDS − vOV ) or, for simplicity, (1 + λvDS), iD = 1 2 kn W L (vGS − Vtn)2(1 + λvDS) (5.23) Here λ is a device parameter having the units of reciprocal volts V−1 . The value of λ depends both on the process technology used to fabricate the device and on the channel length L that the circuit designer selects. Specifically, the value of λ is much larger for newer submicron technologies than for older technologies. This makes intuitive sense: Newer technologies have very short channels, and are thus much more greatly impacted by the channel-length modulation effect. Also, for a given process technology, λ is inversely proportional to L. A typical set of iD–vDS characteristics showing the effect of channel-length modulation is displayed in Fig. 5.17. The observed linear dependence of iD on vDS in the saturation region is represented in Eq. (5.23) by the factor (1 + λvDS). From Fig. 5.17 we observe that when the straight-line iD–vDS characteristics are extrapolated, they intercept the vDS axis at the point, vDS = −VA, where VA is a positive voltage. Equation (5.23), however, indicates that iD = 0 at vDS = −1/λ. It follows that VA = 1 λ and thus VA is a device parameter with the dimensions of V. For a given process, VA is proportional to the channel length L that the designer selects for a MOSFET. We can isolate the dependence of VA on L by expressing it as VA = VAL where VA is entirely process-technology dependent, with the dimensions of volts per micron. Typically, VA falls in the range of 5 V/μm to 50 V/μm. The voltage VA is usually referred to as the Early voltage, after J. M. Early, who discovered a similar phenomenon for the BJT (Chapter 6). iD Triode Saturation VOV Slope = 1 ro ϪVA ϭ Ϫ1/ ␭ 0 vDS Figure 5.17 Effect of vDS on iD in the saturation region. The MOSFET parameter VA depends on the process technology and, for a given process, is proportional to the channel length L. 5.2 Current–Voltage Characteristics 273 Equation (5.23) indicates that when channel-length modulation is taken into account, the saturation values of iD depend on vDS. Thus, for a given vGS, a change vDS yields a corresponding change iD in the drain current iD. It follows that the output resistance of the current source representing iD in saturation is no longer infinite. Defining the output resistance ro as8 ro ≡ ∂ iD ∂ v DS −1 vGS constant (5.24) and using Eq. (5.23) results in ro = λ kn 2 W L (VGS − Vtn)2 −1 (5.25) which can be written as or, equivalently, ro = 1 λID (5.26) ro = VA ID (5.27) where ID is the drain current without channel-length modulation taken into account; that is, ID = 1 2 kn W L (VGS −Vtn )2 (5.27 ) Thus the output resistance is inversely proportional to the drain current.9 Finally, we show in Fig. 5.18 the large-signal, equivalent-circuit model incorporating ro. iG = 0 iD ϩ vGS kЈ Vtn)2 Ϫ Figure 5.18 Large-signal, equivalent-circuit model of the n-channel MOSFET in saturation, incorporating the output resistance ro. The output resistance models the linear dependence of iD on vDS and is given by Eq. (5.27). 8In this book we use ro to denote the output resistance in saturation, and rDS to denote the drain-to-source resistance in the triode region, for small vDS. 9In applying Eq. (5.27) we will usually drop the prime on ID and simply use ro = VA/ID where ID is the drain current without channel-length modulation. 274 Chapter 5 MOS Field-Effect Transistors (MOSFETs) EXERCISE 5.6 An NMOS transistor is fabricated in a 0.4-μm process having μnCox = 200 μA/V2 and VA = 50 V/μm of channel length. If L = 0.8 μm and W = 16 μm, find VA and λ. Find the value of ID that results when the device is operated with an overdrive voltage VOV = 0.5 V and VDS = 1 V. Also, find the value of ro at this operating point. If VDS is increased by 2 V, what is the corresponding change in ID? Ans. 40 V; 0.025 V−1; 0.51 mA; 80 k ; 0.025 mA 5.2.5 Characteristics of the p-Channel MOSFET The circuit symbol for the p-channel enhancement-type MOSFET is shown in Fig. 5.19(a). Figure 5.19(b) shows a modified circuit symbol in which an arrowhead pointing in the normal direction of current flow is included on the source terminal. For the case where the source is connected to the substrate, the simplified symbol of Fig. 5.19(c) is usually used. S S G BG B D D (a) (b) (c) Figure 5.19 (a) Circuit symbol for the p-channel enhancement-type MOSFET. (b) Modified symbol with an arrowhead on the source lead. (c) Simplified circuit symbol for the case where the source is connected to the body. The regions of operation of the PMOS transistor and the corresponding conditions and expression for iD are shown in Table 5.2. Observe that the equations are written in a way that emphasizes physical intuition and avoids the confusion of negative signs. Thus while Vtp is by convention negative, we use Vtp , and the voltages vSG and vSD are positive. Also, in all of our circuit diagrams we will always draw p-channel devices with their sources on top so that current flows from top to bottom. Finally, we note that PMOS devices also suffer from the channel-length modulation effect. This can be taken into account by including a factor (1 + |λ|vSD) in the saturation-region expression for iD as follows iD = 1 2 kp W L vSG − Vtp 2(1 + |λ|vSD) (5.28) 5.2 Current–Voltage Characteristics 275 Table 5.2 Regions of Operation of the Enhancement PMOS Transistor or equivalently iD = 1 2 kp W L vSG − Vtp 2 1 + vSD |VA| (5.29) where λ and VA (the Early voltage for the PMOS transistor) are by convention negative quantities, hence we use |λ| and |VA|. Finally, we should note that for a given CMOS fabrication process λn and λp are generally not equal, and similarly for VAn and VAp . To recap, to turn a PMOS transistor on, the gate voltage has to be made lower than that of the source by at least Vtp . To operate in the triode region, the drain voltage has to exceed that of the gate by at least Vtp ; otherwise, the PMOS operates in saturation. Finally, Fig. 5.20 provides a pictorial representation of these operating conditions. 276 Chapter 5 MOS Field-Effect Transistors (MOSFETs) ϩ ԽVtpԽ ϩ ԽVOVԽ Ϫ G S ԽVOVԽ Triode D ԽVtpԽ Saturation Figure 5.20 The relative levels of the terminal voltages of the enhancement-type PMOS transistor for operation in the triode region and in the saturation region. EXERCISE 5.7 The PMOS transistor shown in Fig. E5.7 has Vtp = −1 V, kp = 60 μA/V2, and W/L = 10. (a) Find the range of VG for which the transistor conducts. (b) In terms of VG, find the range of VD for which the transistor operates in the triode region. (c) In terms of VG, find the range of VD for which the transistor operates in saturation. (d) Neglecting channel-length modulation (i.e., assuming λ = 0), find the values of VOV and VG and the corresponding range of VD to operate the transistor in the saturation mode with ID = 75 μA. (e) If λ = – 0.02 V−1, find the value of ro corresponding to the overdrive voltage determined in (d). (f) For λ = – 0.02 V−1 and for the value of VOV determined in (d), find ID at VD = +3 V and at VD = 0 V; hence, calculate the value of the apparent output resistance in saturation. Compare to the value found in (e). ϩ5 V VG ID VD Figure E5.7 Ans. (a) VG ≤ + 4 V; (b) VD ≥ VG + 1; (c) VD ≤ VG + 1; (d) 0.5 V, 3.5 V, ≤4.5 V; (e) 0.67 M ; (f) 78 μA, 82.5 μA, 0.67 M (same) 5.3 MOSFET Circuits at DC Having studied the current–voltage characteristics of MOSFETs, we now consider circuits in which only dc voltages and currents are of concern. Specifically, we shall present a series of design and analysis examples of MOSFET circuits at dc. The objective is to instill in the 5.3 MOSFET Circuits at DC 277 reader a familiarity with the device and the ability to perform MOSFET circuit analysis both rapidly and effectively. In the following examples, to keep matters simple and thus focus attention on the essence of MOSFET circuit operation, we will generally neglect channel-length modulation; that is, we will assume λ = 0. We will find it convenient to work in terms of the overdrive voltage; VOV = VGS – Vtn for NMOS and |VOV | = VSG − Vtp for PMOS. Example 5.3 Design the circuit of Fig. 5.21: that is, determine the values of RD and RS so that the transistor operates at ID = 0.4 mA and VD = +0.5 V. The NMOS transistor has Vt = 0.7 V, μnCox = 100 μA/V2, L = 1 μm, and W = 32 μm. Neglect the channel-length modulation effect (i.e., assume that λ = 0). VDD = ϩ2.5 V VSS = Ϫ2.5 V Figure 5.21 Circuit for Example 5.3. Solution To establish a dc voltage of +0.5 V at the drain, we must select RD as follows: RD = VDD − ID VD = 2.5 − 0.5 = 5 k 0.4 To determine the value required for RS, we need to know the voltage at the source, which can be easily found if we know VGS. This in turn can be determined from VOV . Toward that end, we note that since VD = 0.5 V is greater than VG, the NMOS transistor is operating in the saturation region, and we can use the saturation-region expression of iD to determine the required value of VOV , ID = 1 2 μnCox W L VO2V Then substituting ID = 0.4 mA = 400 μA, μnCox = 100 μA/V2, and W/L = 32/1 gives 400 = 1 2 × 100 × 32 1 VO2V 278 Chapter 5 MOS Field-Effect Transistors (MOSFETs) Example 5.3 continued which results in Thus, VOV = 0.5 V VGS = Vt + VOV = 0.7 + 0.5 = 1.2 V Referring to Fig. 5.21, we note that the gate is at ground potential. Thus, the source must be at –1.2 V, and the required value of RS can be determined from RS = VS − VSS ID = −1.2 − (−2.5) = 3.25 k 0.4 EXERCISE D5.8 Redesign the circuit of Fig. 5.21 for the following case: VDD = –VSS = 2.5 V, Vt = 1 V, μnCox = 60 μA/V2, W/L = 120 μm/3 μm, ID = 0.3 mA, and VD = +0.4 V. Ans. RD = 7 k ; RS = 3.3 k Example 5.4 Figure 5.22 shows an NMOS transistor with its drain and gate terminals connected together. Find the i−v relationship of the resulting two-terminal device in terms of the MOSFET parameters kn = kn(W/L) and Vtn. Neglect channel-length modulation (i.e., λ = 0). Note that this two-terminal device is known as a diode-connected transistor. iϩ v Ϫ Figure 5.22 5.3 MOSFET Circuits at DC 279 Solution Since vD = vG implies operation in the saturation mode, 1W iD = 2 kn L vGS − Vtn 2 Now, i = iD and v = vGS, thus i = 1 2 kn W L v − Vtn 2 Replacing kn W L by kn results in i= 1 2 kn v − Vtn 2 EXERCISES D5.9 For the circuit in Fig. E5.9, find the value of R that results in VD = 0.7 V. The MOSFET has Vtn = 0.5 V, μn Cox = 0.4 mA/V2, W/L = 0.72 0.18 μm μm , and λ = 0. Ans. 34.4 k ϩ1.8 V R VD Q1 Figure E5.9 D5.10 Figure E5.10 shows a circuit obtained by augmenting the circuit of Fig. E5.9 considered in Exercise 5.9 with a transistor Q2 identical to Q1 and a resistance R2. Find the value of R2 that results in Q2 operating at the edge of the saturation region. Use your solution to Exercise 5.9. Ans. 50 k 280 Chapter 5 MOS Field-Effect Transistors (MOSFETs) VDD = 1.8 V R2 R Q2 Q1 Figure E5.10 Example 5.5 Design the circuit in Fig. 5.23 to establish a drain voltage of 0.1 V. What is the effective resistance between drain and source at this operating point? Let Vtn = 1 V and kn(W/L) = 1 mA/ V2. VDD = ϩ5 V ID RD VD = ϩ0.1 V Figure 5.23 Circuit for Example 5.5. Solution Since the drain voltage is lower than the gate voltage by 4.9 V and Vtn = 1 V, the MOSFET is operating in the triode region. Thus the current ID is given by W ID = kn L VGS − Vtn VDS − 1 2 VD2S 1 ID = 1 × (5 − 1) × 0.1 − × 0.01 2 = 0.395 mA 5.3 MOSFET Circuits at DC 281 The required value for RD can be found as follows: RD = VDD − VD ID = 5 − 0.1 = 12.4 k 0.395 In a practical discrete-circuit design problem, one selects the closest standard value available for, say, 5% resistors—in this case, 12 k ; see Appendix J. Since the transistor is operating in the triode region with a small VDS, the effective drain-to-source resistance can be determined as follows: rDS = VDS ID = 0.1 = 253 0.395 Alternatively, we can determine rDS by using the formula 1 rDS = knVOV to obtain rDS = 1× 1 (5 − 1) = 0.25 k = 250 which is close to the value found above. EXERCISE 5.11 If in the circuit of Example 5.5 the value of RD is doubled, find approximate values for ID and VD. Ans. 0.2 mA; 0.05 V Example 5.6 Analyze the circuit shown in Fig. 5.24(a) to determine the voltages at all nodes and the currents through all branches. Let Vtn = 1 V and kn(W/L) = 1 mA/V2. Neglect the channel-length modulation effect (i.e., assume λ = 0). 282 Chapter 5 MOS Field-Effect Transistors (MOSFETs) Example 5.6 continued VDD = ϩ10 V RG1 = 10 M⍀ RD = 6 k⍀ RG2 = 10 M⍀ RS = 6 k⍀ 10 M⍀ ϩ5 V ϩ10 V 0.5 μA ID 0 10 M⍀ ID 6 k⍀ 10 Ϫ 6 ID 6 ID 6 k⍀ (a) (b) Figure 5.24 (a) Circuit for Example 5.6. (b) The circuit with some of the analysis details shown. Solution Since the gate current is zero, the voltage at the gate is simply determined by the voltage divider formed by the two 10-M resistors, VG = VDD RG2 RG2 + RG1 = 10 × 10 10 + 10 = +5 V With this positive voltage at the gate, the NMOS transistor will be turned on. We do not know, however, whether the transistor will be operating in the saturation region or in the triode region. We shall assume saturation-region operation, solve the problem, and then check the validity of our assumption. Obviously, if our assumption turns out not to be valid, we will have to solve the problem again for triode-region operation. Refer to Fig. 5.24(b). Since the voltage at the gate is 5 V and the voltage at the source is ID ( mA) × 6 (k ) = 6ID (V), we have VGS = 5 − 6ID Thus ID is given by ID = 1 2 kn W L VGS − Vtn 2 = 1 ×1× 2 5 − 6ID − 1 2 5.3 MOSFET Circuits at DC 283 which results in the following quadratic equation in ID: 18ID2 − 25ID + 8 = 0 This equation yields two values for ID: 0.89 mA and 0.5 mA. The first value results in a source voltage of 6 × 0.89 = 5.34 V, which is greater than the gate voltage and does not make physical sense as it would imply that the NMOS transistor is cut off. Thus, ID = 0.5 mA VS = 0.5 × 6 = +3 V VGS = 5 − 3 = 2 V VD = 10 − 6 × 0.5 = +7 V Since VD > VG − Vtn, the transistor is operating in saturation, as initially assumed. EXERCISES 5.12 For the circuit of Fig. 5.24, what is the largest value that RD can have while the transistor remains in the saturation mode? Ans. 12 k D5.13 Redesign the circuit of Fig. 5.24 for the following requirements: VDD = +5 V, ID = 0.32 mA, VS = 1.6 V, VD = 3.4 V, with a 1-μA current through the voltage divider RG1, RG2. Assume the same MOSFET as in Example 5.6. Ans. RG1 = 1.6 M ; RG2 = 3.4 M , RS = RD = 5 k Example 5.7 Design the circuit of Fig. 5.25 so that the transistor operates in saturation with ID = 0.5 mA and VD = +3 V. Let the PMOS transistor have Vtp = −1 V and kp(W/L) = 1 mA/V2. Assume λ = 0. What is the largest value that RD can have while maintaining saturation-region operation? 284 Chapter 5 MOS Field-Effect Transistors (MOSFETs) Example 5.7 continued VDD = ϩ5 V RG1 VD = ϩ3 V RG2 RD ID = 0.5 mA Figure 5.25 Circuit for Example 5.7. Solution Since the MOSFET is to be in saturation, we can write 1W ID = 2 kp L VOV 2 Substituting ID = 0.5 mA and kpW/L = 1 mA/V2, we obtain VOV = 1 V and VSG = Vtp + VOV = 1 + 1 = 2 V Since the source is at +5 V, the gate voltage must be set to +3 V. This can be achieved by the appropriate selection of the values of RG1 and RG2. A possible selection is RG1 = 2 M and RG2 = 3 M . The value of RD can be found from RD = VD ID = 3 0.5 = 6 k Saturation-mode operation will be maintained up to the point that VD exceeds VG by Vtp ; that is, until VDmax = 3 + 1 = 4 V This value of drain voltage is obtained with RD given by 4 RD = 0.5 = 8 k 5.3 MOSFET Circuits at DC 285 EXERCISE D5.14 For the circuit in Fig. E5.14, find the value of R that results in the PMOS transistor operating with an overdrive voltage VOV = 0.6 V. The threshold voltage is Vtp = − 0.4 V, the process transconductance parameter kp = 0.1 mA/V2, and W/L = 10 μm/0.18 μm. Ans. 800 ϩ1.8 V R Figure E5.14 Example 5.8 The NMOS and PMOS transistors in the circuit of Fig. 5.26(a) are matched, with kn Wn/Ln = kp Wp/Lp = 1 mA/V2 and Vtn = − Vtp = 1 V. Assuming λ = 0 for both devices, find the drain currents iDN and iDP, as well as the voltage vO, for vI = 0 V, +2.5 V, and −2.5 V. ϩ2.5 V ϩ2.5 V QP QP iDP IDP vI vO 0 V vO iDN IDN QN 10 k⍀ QN 10 k⍀ Ϫ2.5 V (a) Figure 5.26 Circuits for Example 5.8. Ϫ2.5 V (b) 286 Chapter 5 MOS Field-Effect Transistors (MOSFETs) Example 5.8 continued ϩ2.5 V IDN QN vO IDN 10 k⍀ Ϫ2.5 V ϩ2.5 V QP IDP vO IDP 10 k⍀ Ϫ2.5 V (c) (d) Figure 5.26 continued Solution Figure 5.26(b) shows the circuit for the case vI = 0 V. We note that since QN and QP are perfectly matched and are operating at equal values of VGS = 2.5 V, the circuit is symmetrical, which dictates that vO = 0 V. Thus both QN and QP are operating with VDG = 0 and, hence, in saturation. The drain currents can now be found from IDP = IDN = 1 2 × 1 × (2.5 − 1)2 = 1.125 mA Next, we consider the circuit with vI = +2.5 V. Transistor QP will have a VSG of zero and thus will be cut off, reducing the circuit to that shown in Fig. 5.26(c). We note that vO will be negative, and thus vGD will be greater than Vtn, causing QN to operate in the triode region. For simplicity we shall assume that vDS is small and thus use IDN kn Wn/Ln VGS − Vtn VDS = 1[2.5 − (−2.5) − 1][vO − (−2.5)] From the circuit diagram shown in Fig. 5.26(c), we can also write IDN (mA) = 0 − vO 10(k ) 5.3 MOSFET Circuits at DC 287 These two equations can be solved simultaneously to yield IDN = 0.244 mA vO = −2.44 V Note that VDS = −2.44 − (−2.5) = 0.06 V, which is small as assumed. Finally, the situation for the case vI = −2.5 V [Fig. 5.26(d)] will be the exact complement of the case vI = +2.5 V : Transistor QN will be off. Thus IDN = 0, QP will be operating in the triode region with IDP = 0.244 mA and vO = +2.44 V. EXERCISE 5.15 The NMOS and PMOS transistors in the circuit of Fig. E5.15 are matched with kn Wn/Ln = kp Wp/Lp = 1 mA/V2 and Vtn = − Vtp = 1 V. Assuming λ = 0 for both devices, find the drain currents iDN and iDP and the voltage vO for vI = 0 V, +2.5 V, and –2.5 V. Ans. vI = 0 V: 0 mA, 0 mA, 0 V; vI = + 2.5 V: 0.104 mA, 0 mA, 1.04 V; vI = − 2.5 V: 0 mA, 0.104 mA, –1.04 V ϩ2.5 V QN iDN vI vO iDP QP 10 k⍀ Ϫ2.5 V Figure E5.15 Concluding Remark If a MOSFET is conducting but its mode of operation (saturation or triode) is not known, we assume operation in the saturation region, solve the problem, and check whether the conditions for saturation-mode operation are satisfied. If not, then the MOSFET is operating in the triode region and the analysis is done accordingly. 288 Chapter 5 MOS Field-Effect Transistors (MOSFETs) GORDON MOORE— HIS LAW: A half-century ago, Gordon Moore, who would go on to become a cofounder first of Fairchild Semiconductor and then of Intel, presented a startling idea in the issue of Electronics Magazine dated April 19, 1965. Moore, who had a doctorate in chemistry, had projected the potential growth of the integrated-circuit industry based on five points spanning a seven-year period from 1958 to 1965. The conclusion he reached—that the number of transistors per chip had been increasing and would continue to increase by a factor of 2 every two years or so—was destined to propel progress in integrated circuits over the succeeding decades into the twenty-first century. Doubling of the number of transistors was predicted on the basis of another prediction: the continuing shrinkage of transistor dimensions. In early recognition of the importance of this prediction, Carver Mead, a pioneer in very large scale integration (VLSI), soon began to refer to this prediction as “Moore’s law.” (See Chapter 15, Section 15.1, for the implications of Moore’s law). 5.4 The Body Effect and Other Topics In this section we briefly consider a number of important though secondary issues. 5.4.1 The Role of the Substrate—The Body Effect In many applications the source terminal is connected to the substrate (or body) terminal B, which results in the pn junction between the substrate and the induced channel (review Fig. 5.5) having a constant zero (cutoff) bias. In such a case the substrate does not play any role in circuit operation and its existence can be ignored altogether. In integrated circuits, however, the substrate is usually common to many MOS transistors. In order to maintain the cutoff condition for all the substrate-to-channel junctions, the substrate is usually connected to the most negative power supply in an NMOS circuit (the most positive in a PMOS circuit). The resulting reverse-bias voltage between source and body (VSB in an n-channel device) will have an effect on device operation. To appreciate this fact, consider an NMOS transistor and let its substrate be made negative relative to the source. The reverse-bias voltage will widen the depletion region (refer to Fig. 5.2). This in turn reduces the channel depth. To return the channel to its former state, vGS has to be increased. The effect of VSB on the channel can be most conveniently represented as a change in the threshold voltage Vt. Specifically, it has been shown that increasing the reverse substrate bias voltage VSB results in an increase in Vt according to the relationship Vt = Vt0 + γ 2φf + VSB − 2φf (5.30) where Vt0 is the threshold voltage for VSB = 0; φf is a physical parameter with (2φf ) typically 0.6 V; γ is a fabrication-process parameter given by γ = 2qNAes Cox (5.31) where q is the magnitude of the electron charge (1.6 × 10−19 C), NA is the doping concentration of the p-type substrate, and es is the permittivity 1.04 × 10−12 F/cm). The parameter γ has the of silicon (11.7√e0 = 11.7 × 8.854 × dimension of V and is typically 10−14 = 0.4 V1/2. Finally, note that Eq. (5.30) applies equally well for p-channel devices with VSB replaced by 5.4 The Body Effect and Other Topics 289 the reverse bias of the substrate, VBS (or, alternatively, replace VSB by |VSB|) and note that γ is negative. Also, in evaluating γ , NA must be replaced with ND, the doping concentration of the n well in which the PMOS is formed. For p-channel devices, 2φf is typically 0.75 V, and γ is typically –0.5 V1/2. EXERCISE 5.16 An NMOS transistor has Vt0 = 0.8 V, 2φf = 0.7 V, and γ = 0.4 V1/2. Find Vt when VSB = 3 V. Ans. 1.23 V Equation (5.30) indicates that an incremental change in VSB gives rise to an incremental change in Vt, which in turn results in an incremental change in iD even though vGS might have been kept constant. It follows that the body voltage controls iD; thus the body acts as another gate for the MOSFET, a phenomenon known as the body effect. Here we note that the parameter γ is known as the body-effect parameter. 5.4.2 Temperature Effects Both Vt and k are temperature sensitive. The magnitude of Vt decreases by about 2 mV for every 1°C rise in temperature. This decrease in |Vt| gives rise to a corresponding increase in drain current as temperature is increased. However, because k decreases with temperature and its effect is a dominant one, the overall observed effect of a temperature increase is a decrease in drain current. This very interesting result is put to use in applying the MOSFET in power circuits (Chapter 12). 5.4.3 Breakdown and Input Protection As the voltage on the drain is increased, a value is reached at which the pn junction between the drain region and substrate suffers avalanche breakdown (see Section 3.5.3). This breakdown usually occurs at voltages of 20 V to 150 V and results in a somewhat rapid increase in current (known as a weak avalanche). Another breakdown effect that occurs at lower voltages (about 20 V) in modern devices is called punch-through. It occurs in devices with relatively short channels when the drain voltage is increased to the point that the depletion region surrounding the drain region extends through the channel to the source. The drain current then increases rapidly. Normally, punch-through does not result in permanent damage to the device. Yet another kind of breakdown occurs when the gate-to-source voltage exceeds about 30 V. This is the breakdown of the gate oxide and results in permanent damage to the device. Although 30 V may seem high, it must be remembered that the MOSFET has a very high input resistance and a very small input capacitance, and thus small amounts of static charge accumulating on the gate capacitor can cause its breakdown voltage to be exceeded. To prevent the accumulation of static charge on the gate capacitor of a MOSFET, gate-protection devices are usually included at the input terminals of MOS integrated circuits. The protection mechanism invariably makes use of clamping diodes. 290 Chapter 5 MOS Field-Effect Transistors (MOSFETs) 5.4.4 Velocity Saturation At high longitudinal electric fields, the drift velocity of charge carriers in the channel reaches an upper limit (approximately 107 cm/s for electrons and holes in silicon). This effect, which in modern very-short-channel devices can occur for vDS lower than 1 V, is called velocity saturation. It can be shown that when velocity saturation occurs, the current iD will no longer be related to vGS by the square-law relationship. Rather, iD becomes linearly dependent on vGS and the transconductance gm becomes constant and independent of vGS. In Chapter 15, we shall consider velocity saturation in our study of deep-submicron (i.e., L < 0.25 μm) CMOS digital circuits. 5.4.5 The Depletion-Type MOSFET We conclude this section with a brief discussion of another type of MOSFET, the depletion-type MOSFET. Its structure is similar to that of the enhancement-type MOSFET with one important difference: The depletion MOSFET has a physically implanted channel. Thus an n-channel depletion-type MOSFET has an n-type silicon region connecting the n+ source and the n+ drain regions at the top of the p-type substrate. Thus if a voltage vDS is applied between drain and source, a current iD flows for vGS = 0. In other words, there is no need to induce a channel, unlike the case of the enhancement MOSFET. The channel depth and hence its conductivity can be controlled by vGS in exactly the same manner as in the enhancement-type device. Applying a positive vGS enhances the channel by attracting more electrons into it. Here, however, we also can apply a negative vGS, which causes electrons to be repelled from the channel, and thus the channel becomes shallower and its conductivity decreases. The negative vGS is said to deplete the channel of its charge carriers, iD Depletion mode Enhancement mode vDS ≥ vGS Ϫ Vt iD ϩ IDSS D iG = 0 G vDS ϩ vGS S Ϫ 0 vGS Ϫ Vt (a) (b) Figure 5.27 The circuit symbol (a) and the iD–vGS characteristic in saturation (b) for an n-channel depletion-type MOSFET. Summary 291 and this mode of operation (negative vGS) is called depletion mode. As the magnitude of vGS is increased in the negative direction, a value is reached at which the channel is completely depleted of charge carriers and iD is reduced to zero even though vDS may be still applied. This negative value of vGS is the threshold voltage of the n-channel depletion-type MOSFET. The description above suggests (correctly) that a depletion-type MOSFET can be operated in the enhancement mode by applying a positive vGS and in the depletion mode by applying a negative vGS. This is illustrated in Fig. 5.27, which shows both the circuit symbol for the depletion NMOS transistor (Fig. 5.27a) and its iD–vGS characteristic. Observe that here the threshold voltage Vtn is negative. The iD–vDS characteristics (not shown) are similar to those for the enhancement-type MOSFET except for the negative Vtn. Finally, note that the device symbol denotes the existing channel via the shaded area next to the vertical line. Depletion-type MOSFETs can be fabricated on the same IC chip as enhancement-type devices, resulting in circuits with improved characteristics, as will be shown in a later chapter. The depletion-type MOSFET, however, is a specialty device and is not commonly used. EXERCISE 5.17 For a depletion-type NMOS transistor with Vt = −2 V and kn(W/L) = 2 mA/V2, find the minimum vDS required to operate in the saturation region when vGS = +1 V. What is the corresponding value of iD? Ans. 3 V; 9 mA Summary The enhancement-type MOSFET is currently the most widely used semiconductor device. It is the basis of CMOS technology, which is the most popular IC fabrication technology at this time. CMOS provides both n-channel (NMOS) and p-channel (PMOS) transistors, which increases design flexibility. The minimum MOSFET channel length achievable with a given CMOS process is used to characterize the process. This figure has been continually reduced and is currently 32 nm. The overdrive voltage, vOV ≡ vGS − Vt , is the key quantity that governs the operation of the MOSFET. For the MOSFET to operate in the saturation region, which is the region for amplifier application, vDS ≥ vOV , and the resulting iD = 1 2 μn Cox (W/L)v 2 OV (for NMOS; replace μn with μp for PMOS). If vDS < vOV , the MOSFET operates in the triode region, which together with cutoff is used for operating the MOSFET as a switch. Tables 5.1 and 5.2 provide summaries of the conditions and relationships that describe the operation of NMOS and PMOS transistors, respectively. In saturation, iD shows some linear dependence on vDS as a result of the change in channel length. This channel-length modulation phenomenon becomes more pronounced as L decreases. It is modeled by ascribing an output resistance ro = VA /ID to the MOSFET model. Here, the Early voltage VA = VA L, where VA is a process-dependent parameter. In the analysis of dc MOSFET circuits, if a MOSFET is conducting, but its region of operation (saturation or triode) is not known, one assumes saturation-mode operation. Then, one solves the problem and checks to determine whether the assumption was justified. If not, then the transistor is operating in the triode region, and the analysis is done accordingly. The depletion-type MOSFET has an implanted channel and thus can be operated in either the depletion or enhancement mode. It is characterized by the same equations used for the enhancement device except for having a negative Vtn (positive Vtp for depletion PMOS transistors). PROBLEMS Computer Simulations Problems Problems identified by the Multisim/PSpice icon are intended to demonstrate the value of using SPICE simulation to verify hand analysis and design, and to investigate important issues such as allowable signal swing and amplifier nonlinear distortion. Instructions to assist in setting up PSPice and Multisim simulations for all the indicated problems can be found in the corresponding files on the website. Note that if a particular parameter value is not specified in the problem statement, you are to make a reasonable assumption. Section 5.1: Device Structure and Physical Operation 5.1 MOS technology is used to fabricate a capacitor, utilizing the gate metallization and the substrate as the capacitor electrodes. Find the area required per 1-pF capacitance for oxide thickness ranging from 2 nm to 10 nm. For a square plate capacitor of 10 pF, what dimensions are needed? 5.2 Calculate the total charge stored in the channel of an NMOS transistor having Cox = 9 fF/μm2, L = 0.36 μm, and W = 3.6 μm, and operated at VOV = 0.2 V and VDS = 0 V. 5.3 Use dimensional analysis to show that the units of the process transconductance parameter kn are A/V2. What are the dimensions of the MOSFET transconductance parameter kn? 5.4 An NMOS transistor that is operated with a small vDS is found to exhibit a resistance rDS. By what factor will rDS change in each of the following situations? (a) VOV is doubled. (b) The device is replaced with another fabricated in the same technology but with double the width. (c) The device is replaced with another fabricated in the same technology but with both the width and length doubled. (d) The device is replaced with another fabricated in a more advanced technology for which the oxide thickness is halved and similarly for W and L (assume μn remains unchanged). D 5.5 An NMOS transistor fabricated in a technology for which kn = 400 μA/V2 and Vt = 0.5 V is required to operate with a small vDS as a variable resistor ranging in value from 250 to 1 k . Specify the range required for the control voltage VGS and the required transistor width W. It is required to use the smallest possible device, as limited by the minimum channel length of this technology (Lmin = 0.18 μm) and the maximum allowed voltage of 1.8 V. 5.6 Sketch a set of iD−vDS characteristic curves for an NMOS transistor operating with a small vDS (in the manner shown in Fig. 5.4). Let the MOSFET have kn = 5 mA/V2 and Vtn = 0.5 V. Sketch and clearly label the graphs for VGS = 0.5, 1.0, 1.5, 2.0, and 2.5 V. Let VDS be in the range 0 to 50 mV. Give the value of rDS obtained for each of the five values of VGS. Although only a sketch, your diagram should be drawn to scale as much as possible. D 5.7 An n-channel MOS device in a technology for which oxide thickness is 4 nm, minimum channel length is 0.18 μm, kn = 400 μA/V2, and Vt = 0.5 V operates in the triode region, with small vDS and with the gate–source voltage in the range 0 V to +1.8 V. What device width is needed to ensure that the minimum available resistance is 1 k ? 5.8 Consider an NMOS transistor operating in the triode region with an overdrive voltage VOV . Find an expression for the incremental resistance rds ≡ 1 ∂ iD v∂ DS vDS =VDS = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 5 PROBLEMS Problems 293 Give the values of rds in terms of kn and VOV for VDS = 0, 0.2VOV , 0.5VOV , 0.8VOV , and VOV . 5.9 An NMOS transistor with kn = 4 mA/V2 and Vt = 0.5 V is operated with VGS = 1.0 V. At what value of VDS does the transistor enter the saturation region? What value of ID is obtained in saturation? 5.10 Consider a CMOS process for which Lmin = 0.25 μm, tox = 6 nm, μn = 460 cm2/V · s, and Vt = 0.5 V. (a) Find Cox and kn. (b) For an NMOS transistor with W/L = 20 μm/0.25 μm, calculate the values of VOV , VGS, and VDSmin needed to operate the transistor in the saturation region with a dc current ID = 0.5 mA. (c) For the device in (b), find the values of VOV and VGS required to cause the device to operate as a 100- resistor for very small vDS. 5.11 A p-channel MOSFET with a threshold voltage Vtp = −0.7 V has its source connected to ground. (a) What should the gate voltage be for the device to operate with an overdrive voltage of VOV = 0.4 V? (b) With the gate voltage as in (a), what is the highest voltage allowed at the drain while the device operates in the saturation region? (c) If the drain current obtained in (b) is 0.5 mA, what would the current be for VD = −20 mV and for VD = −2V? 5.12 With the knowledge that μp = 0.4 μn, what must be the relative width of n-channel and p-channel devices having equal channel lengths if they are to have equal drain currents when operated in the saturation mode with overdrive voltages of the same magnitude? 5.13 An n-channel device has kn = 100 μA/V2, Vt = 0.7 V, and W/L = 20. The device is to operate as a switch for small vDS, utilizing a control voltage vGS in the range 0 V to 5 V. Find the switch closure resistance, rDS, and closure voltage, VDS, obtained when vGS = 5 V and iD = 1 mA. If μp 0.4 μn, what must W/L be for a p-channel device that provides the same performance as the n-channel device in this application? 5.14 Consider an n-channel MOSFET with tox = 6 nm, μn = 460 cm2/V · s, Vt = 0.5 V, and W/L = 10. Find the drain current in the following cases: (a) vGS = 2.5 V and vDS = 1 V (b) vGS = 2 V and vDS = 1.5 V (c) vGS = 2.5 V and vDS = 0.2 V (d) vGS = vDS = 2.5 V *5.15 This problem illustrates the central point in the electronics revolution that has been in effect for the past four decades: By continually reducing the MOSFET size, we are able to pack more devices on an IC chip. Gordon Moore, co-founder of Intel Corporation, predicted this exponential growth of chip-packing density very early in the history of the development of the integrated circuit in the formulation that has become known as Moore’s law. The table on the next page shows four technology generations, each characterized by the minimum possible MOSFET channel length (row 1). In going from one generation to another, both L and tox are scaled by the same factor. The power supply utilized VDD is also scaled by the same factor, to keep the magnitudes of all electrical fields within the device unchanged. Unfortunately, but for good reasons, Vt cannot be scaled similarly. Complete the table entries, noting that row 5 asks for the transconductance parameter of an NMOS transistor with W/L = 10; row 9 asks for the value of ID obtained with VGS = VDS = VDD; row 10 asks for the power P = VDDID dissipated in the circuit. An important quantity is the power density, P/A, asked for in row 11. Finally, you are asked to find the number of transistors that can be placed on an IC chip fabricated in each of the technologies in terms of the number obtained with the 0.5-μm technology (n). = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 5 PROBLEMS 294 Chapter 5 MOS Field-Effect Transistors (MOSFETs) 1 L (μm) 0.5 0.25 0.18 0.13 2 tox (nm) 10 3 Cox (fF/μm2) 4 kn (μA/V2) (μn = 500 cm2/ V · s) 5 kn (mA/V2) For W/L = 10 6 Device area, A (μm2) 7 VDD (V) 5 8 Vt (V) 0.7 0.5 0.4 0.4 9 ID (mA) For VGS = VDS = VDD 10 P (mW) 11 P/A (mW/μm2) 12 Devices per chip n Section 5.2: Current–Voltage Characteristics In the following problems, when λ is not specified, assume it is zero. 5.16 Show that when channel-length modulation is neglected (i.e., λ = 0), plotting iD/kn versus vDS for various values of vOV , and plotting iD/kn versus vOV for vDS ≥ vOV , results in universal representation of the iD−vDS and iD−vGS characteristics of the NMOS transistor. That is, the resulting graphs are both technology and device independent. Furthermore, these graphs apply equally well to the PMOS transistor by a simple relabeling of variables. (How?) What is the slope at vDS = 0 of each of the iD/kn versus vDS graphs? For the iD/kn versus vOV graph, find the slope at a point vOV = VOV . 5.17 An NMOS transistor having Vt = 0.8 V is operated in the triode region with vDS small. With VGS = 1.2 V, it is found to have a resistance rDS of 1 k . What value of VGS is required to obtain rDS = 200 ? Find the corresponding resistance values obtained with a device having twice the value of W. 5.18 A particular MOSFET for which Vtn = 0.5 V and kn(W/L) = 1.6 mA/V2 is to be operated in the saturation region. If iD is to be 50 μA, find the required vGS and the minimum required vDS. Repeat for iD = 200 μA. = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 5 PROBLEMS Problems 295 5.19 A particular n-channel MOSFET is measured to have a drain current of 0.4 mA at VGS = VDS = 1 V and of 0.1 mA at VGS = VDS = 0.8 V. What are the values of kn and Vt for this device? iϩ v iϩ v D 5.20 For a particular IC-fabrication process, the transconductance parameter kn = 400 μA/V2, and Vt = 0.5 V. In an application in which vGS = vDS = Vsupply = 1.8 V, a drain current of 2 mA is required of a device of minimum length of 0.18 μm. What value of channel width must the design use? Ϫ (a) Figure P5.24 Ϫ (b) 5.21 An NMOS transistor, operating in the linear-resistance region with vDS = 50 mV, is found to conduct 25 μA for vGS = 1 V and 50 μA for vGS = 1.5 V. What is the apparent value of threshold voltage Vt? If kn = 50 μA/V2, what is the device W/L ratio? What current would you expect to flow with vGS = 2 V and vDS = 0.1 V? If the device is operated at vGS = 2 V, at what value of vDS will the drain end of the MOSFET channel just reach pinch-off, and what is the corresponding drain current? 5.25 For the circuit in Fig. P5.25, sketch iD versus vS for vS varying from 0 to VDD. Clearly label your sketch. VDD 5.22 For an NMOS transistor, for which Vt = 0.4 V, operating with vGS in the range of 1.0 V to 1.8 V, what is the largest value of vDS for which the channel remains continuous? 5.23 An NMOS transistor, fabricated with W = 20 μm and L = 1 μm in a technology for which kn = 100 μA/V2 and Vt = 0.8 V, is to be operated at very low values of vDS as a linear resistor. For vGS varying from 1.0 V to 4.8 V, what range of resistor values can be obtained? What is the available range if (a) the device width is halved? (b) the device length is halved? (c) both the width and length are halved? iD vS ϩ Ϫ Figure P5.25 5.26 For the circuit in Fig. P5.26, find an expression for vDS in terms of iD. Sketch and clearly label a graph for vDS versus iD. 5.24 When the drain and gate of a MOSFET are connected together, a two-terminal device known as a “diode-connected transistor” results. Figure P5.24 shows such devices obtained iD from MOS transistors of both polarities. Show that (a) the i–v relationship is given by i= 1k W 2L v − Vt 2 (b) the incremental resistance r for a device biased to operate at v = Vt + VOV is given by ∂i W r ≡ 1 ∂v = 1 k L VOV ϩ vDS Ϫ Figure P5.26 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 5 PROBLEMS 296 Chapter 5 MOS Field-Effect Transistors (MOSFETs) Voltage (V) Case VS VG VD VGS VOV VDS a +1.0 +1.0 +2.0 b +1.0 +2.5 +2.0 c +1.0 +2.5 +1.5 d +1.0 +1.5 0 e 0 +2.5 +1.0 f +1.0 +1.0 +1.0 g −1.0 0 0 h −1.5 0 0 i −1.0 0 +1.0 j +0.5 +2.0 +0.5 Region of operation *5.27 The table above lists 10 different cases labeled (a) to (j) for operating an NMOS transistor with Vt = 1 V. In each case the voltages at the source, gate, and drain (relative to the circuit ground) are specified. You are required to complete the table entries. Note that if you encounter a case for which vDS is negative, you should exchange the drain and source before solving the problem. You can do this because the MOSFET is a symmetric device. 5.28 The NMOS transistor in Fig. P5.28 has Vt = 0.4 V and kn(W/L) = 1 mA/V2. Sketch and clearly label iD versus vG with vG varying in the range 0 to +1.8 V. Give equations for the various portions of the resulting graph. 5.29 Figure P5.29 shows two NMOS transistors operating in saturation at equal VGS and VDS. (a) If the two devices are matched except for a maximum possible mismatch in their W/L ratios of 3%, what is the maximum resulting mismatch in the drain currents? (b) If the two devices are matched except for a maximum possible mismatch in their Vt values of 10 mV, what is the maximum resulting mismatch in the drain currents? Assume that the nominal value of Vt is 0.6 V. ϩ1V ϩ2.5 V iD ID1 ID2 ϩ1.0 V Q1 Q2 vG ϩ Ϫ Figure P5.28 Figure P5.29 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 5 PROBLEMS Problems 297 5.30 For a particular MOSFET operating in the saturation region at a constant vGS, iD is found to be 0.5 mA for vDS = 1 V and 0.52 mA for vDS = 2 V. What values of ro, VA, and λ correspond? 5.31 A particular MOSFET has VA = 20 V. For operation at 0.1 mA and 1 mA, what are the expected output resistances? In each case, for a change in vDS of 1 V, what percentage change in drain current would you expect? D 5.32 In a particular IC design in which the standard channel length is 1 μm, an NMOS device with W/L of 10 operating at 200 μA is found to have an output resistance of 100 k , about 1 of that needed. What dimensional change 5 can be made to solve the problem? What is the new device length? The new device width? The new W/L ratio? What is VA for the standard device in this IC? The new device? D 5.33 For a particular n-channel MOS technology, in which the minimum channel length is 0.5 μm, the associated value of λ is 0.03 V−1. If a particular device for which L is 1.5 μm operates in saturation at vDS = 1 V with a drain current of 100 μA, what does the drain current become if vDS is raised to 5 V? What percentage change does this represent? What can be done to reduce the percentage by a factor of 2? 5.34 An NMOS transistor is fabricated in a 0.5-μm process having kn = 200 μA/V2 and VA = 20 V/μm of channel length. If L = 1.5 μm and W = 15 μm, find VA and λ. Find the value of ID that results when the device is operated with an overdrive voltage of 0.5 V and VDS = 2 V. Also, find the value of ro at this operating point. If VDS is increased by 1 V, what is the corresponding change in ID? 5.35 If in an NMOS transistor, both W and L are quadrupled and VOV is halved, by what factor does ro change? D 5.36 Consider the circuit in Fig. P5.29 with both transistors perfectly matched but with the dc voltage at the drain of Q1 lowered to +2 V. If the two drain currents are to be matched within 1% (i.e., the maximum difference allowed between the two currents is 1%), what is the minimum required value of VA? If the technology is specified to have VA = 100 V/μm, what is the minimum channel length the designer must use? 5.37 Complete the missing entries in the following table, which describes characteristics of suitably biased NMOS transistors: MOS 1 2 3 4 λ (V−1) 0.02 VA (V) 20 ID (mA) 0.5 ro (k ) 25 100 0.1 100 500 5.38 A PMOS transistor has kp(W/L) = 100 μA/V2, Vt = −1.0 V, and λ = –0.02 V−1. The gate is connected to ground and the source to +5 V. Find the drain current for vD = +4 V, +2 V, +1 V, 0 V, and –5 V. 5.39 A p-channel transistor for which Vt = 0.8 V and VA = 40 V operates in saturation with vGS = 3 V, vDS = 4 V, and iD = 3 mA. Find corresponding signed values for vGS, vSG, vDS, vSD, Vt, VA, λ, and kp(W/L). 5.40 The table below lists the terminal voltages of a PMOS transistor in six cases, labeled a, b, c, d, e, and f. The transistor has Vtp = −1 V. Complete the table entries. VS VG VD VSG |VOV | VSD Region of operation a +2 +2 0 b +2 +1 0 c +2 0 0 d +2 0 +1 e +2 0 +1.5 f +2 0 +2 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 5 PROBLEMS 298 Chapter 5 MOS Field-Effect Transistors (MOSFETs) 5.41 The PMOS transistor in Fig. P5.41 has Vtp = −0.5 V. As the gate voltage vG is varied from +3 V to 0 V, the transistor moves through all of its three possible modes of operation. Specify the values of vG at which the device changes modes of operation. saturated-mode operation of each transistor at ID = I? In the latter limiting situation, what do V1, V2, V3, and V4 become? ϩ2.5 V ϩ1 V ϩ1 V vG ϩ Ϫ ϩ3 V ϩ1 V I V1 Q1 Q2 V2 I Figure P5.41 Ϫ1.5 V (a) (b) *5.42 Various NMOS and PMOS transistors, numbered 1 ϩ2.5 V to 4, are measured in operation, as shown in the table at the bottom of the page. For each transistor, find the values of μCoxW/L and Vt that apply and complete the table, with V in I volts, I in μA, and μCoxW/L in μA/V2. Assume λ = 0. *5.43 All the transistors in the circuits shown in Fig. P5.43 V3 have the same values of Vt , k , W/L, and λ. Moreover, λ is negligibly small. All operate in saturation at ID = I and VGS = Q3 VDS = 1 V. Find the voltages V1, V2, V3, and V4. If Vt = 0.5 V and I = 0.1 mA, how large a resistor can be inserted in series with each drain while maintaining saturation? If the current source I requires at least 0.5 V between its terminals (c) to operate properly, what is the largest resistor that can be placed in series with each MOSFET source while ensuring Figure P5.43 ϩ1.25 V Q4 V4 I Ϫ1.25 V (d) Case Transistor VS VG VD ID Type Mode μCoxW/L Vt a 1 0 1 2.5 100 1 0 1.5 2.5 400 b 2 5 3 −4.5 50 2 5 2 −0.5 450 c 3 5 3 4 200 3 5 2 0 800 d 4 −2 0 0 72 4 −4 0 −3 270 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 5 PROBLEMS Problems 299 Section 5.3: MOSFET Circuits at DC Note: If λ is not specified, assume it is zero. D 5.44 Design the circuit of Fig. P5.44 to establish a drain current of 0.1 mA and a drain voltage of +0.3 V. The MOSFET has Vt = 0.5 V, μnCox = 400 μA/V2, L = 0.4 μm, and W = 5 μm. Specify the required values for RS and RD. edge of saturation is obtained when the following condition is satisfied: W L RD 2.5 k ϩ1.3 V ϩ1 V RD RD Figure P5.47 RS Ϫ1 V Figure P5.44 5.45 The NMOS transistor in the circuit of Fig. P5.44 has Vt = 0.4 V and kn = 4 mA/V2. The voltages at the source and the drain are measured and found to be −0.6 V and +0.2 V, respectively. What current ID is flowing, and what must the values of RD and RS be? What is the largest value for RD for which ID remains unchanged from the value found? D 5.48 It is required to operate the transistor in the circuit of Fig. P5.47 at the edge of saturation with ID = 0.1 mA. If Vt = 0.4 V, find the required value of RD. D 5.49 The PMOS transistor in the circuit of Fig. P5.49 has Vt = −0.5 V, μpCox = 100 μA/V2, L = 0.18 μm, and λ = 0. Find the values required for W and R in order to establish a drain current of 180 μA and a voltage VD of 1 V. 1.8 V D 5.46 For the circuit in Fig. E5.10, assume that Q1 and Q2 are matched except for having different widths, W1 and W2. Let Vt = 0.5 V, kn = 0.4 mA/V2, L1 = L2 = 0.36 μm, W1 = 1.44 μm, and λ = 0. (a) Find the value of R required to establish a current of 50 μA in Q1. (b) Find W2 and R2 so that Q2 operates at the edge of saturation with a current of 0.5 mA. 5.47 The transistor in the circuit of Fig. P5.47 has kn = 0.4 mA/V2, Vt = 0.4 V, and λ = 0. Show that operation at the Figure P5.49 D 5.50 The NMOS transistors in the circuit of Fig. P5.50 have Vt = 0.5 V, μnCox = 250 μA/V2, λ = 0, and L1 = L2 = 0.25 μm. Find the required values of gate width for each of Q1 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 5 PROBLEMS 300 Chapter 5 MOS Field-Effect Transistors (MOSFETs) and Q2, and the value of R, to obtain the voltage and current values indicated. ϩ2.5 V the drain current is 0.5 mA and the drain voltage is +7 V. If the transistor is replaced with another having Vt = 1.5 V with kn(W/L) = 1.5 mA/V2, find the new values of ID and VD. Comment on how tolerant (or intolerant) the circuit is to changes in device parameters. 0.5 mA ϩ1.8 V ϩ1.0 V D 5.53 Using a PMOS transistor with Vt = −1.5 V, kp (W/L) = 4 mA/V2, and λ = 0, design a circuit that resembles that in Fig. 5.24(a). Using a 10-V supply, design for a gate voltage of +6 V, a drain current of 0.5 mA, and a drain voltage of +5 V. Find the values of RS and RD. Also, find the values of the resistances in the voltage divider feeding the gate, assuming a 1-μA current in the divider. Figure P5.50 5.54 The MOSFET in Fig. P5.54 has Vt = 0.4 V, kn = 500 μA/V2, and λ = 0. Find the required values of W/L and of R so that when vI = VDD = +1.3 V, rDS = 50 and vO = 50 mV. D 5.51 The NMOS transistors in the circuit of Fig. P5.51 have Vt = 0.5 V, μnCox = 90 μA/V2, λ = 0, and L1 = L2 = L3 = 0.5 μm. Find the required values of gate width for each of Q1, Q2, and Q3 to obtain the voltage and current values indicated. VDD R ϩ2.5 V 90 ␮A vO vI ϩ1.5 V ϩ0.8 V Figure P5.54 Figure P5.51 5.52 Consider the circuit of Fig. 5.24(a). In Example 5.5 it was found that when Vt = 1 V and kn(W/L) = 1 mA/V2, 5.55 In the circuits shown in Fig. P5.55, transistors are characterized by Vt = 1 V, k W/L = 4 mA/V2, and λ = 0. (a) Find the labeled voltages V1 through V7. (b) In each of the circuits, replace the current source with a resistor. Select the resistor value to yield a current as close to that of the current source as possible, while using resistors specified in the 1% table provided in Appendix J. = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 5 PROBLEMS ϩ5 V 2 k⍀ V2 V1 2 mA Ϫ5 V (a) ϩ5 V 2 mA V4 ϩ5 V 2 mA V3 (b) ϩ5 V V6 ϩ5 V V1 10 ␮A (a) ϩ5 V V3 1 mA Problems 301 ϩ5 V V2 100 ␮A (b) 10 ␮A V4 V5 1.5 k⍀ Ϫ5 V (c) Figure P5.55 V7 2 mA (d) (c) 1 mA V5 (d) ϩ5 V 400 k⍀ V6 5.56 For each of the circuits in Fig. P5.56, find the labeled node voltages. For all transistors, kn(W/L) = 0.5 mA/V2, (e) (f ) Vt = 0.8 V, and λ = 0. Figure P5.56 continued = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem 302 Chapter 5 MOS Field-Effect Transistors (MOSFETs) CHAPTER 5 PROBLEMS ϩ5 V 2.2 k⍀ V7 ϩ5 V V8 400 k⍀ ϩ10 V ϩ VSG Ϫ ϩ VSD Ϫ R Ϫ5 V (g) (h) I Figure P5.56 continued 5.57 For each of the circuits shown in Fig. P5.57, find the labeled node voltages. The NMOS transistors have Vt = 0.9 V and kn(W/L) = 1.5 mA/V2. 5V Figure P5.58 5.59 For the circuits in Fig. P5.59, μnCox = 3 μpCox = 270 μA/V2, Vt = 0.5 V, λ = 0, L = 1 μm, and W = 3 μm, unless otherwise specified. Find the labeled currents and voltages. 5V 2.5 V 1k 3V 3V Q1 V1 Q2 V2 1k V3 Q1 V4 Q2 V5 1k 2.5 V (a) (b) Figure P5.57 *5.58 For the circuit in Fig. P5.58: (a) Show that for the PMOS transistor to operate in saturation, the following condition must be satisfied: IR ≤ | Vtp | (b) If the transistor is specified to have |Vtp| = 1 V and kp = 0.2 mA/V2, and for I = 0.1 mA, find the voltages VSD and VSG for R = 0, 10 k , 30 k , and 100 k . I1 V2 I3 V4 (a) (b) 3V W = 9 μm I6 V5 (c) Figure P5.59 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 5 PROBLEMS Problems 303 *5.60 For the devices in the circuit of Fig. P5.60, Vt = 1 V, λ = 0, μnCox = 50 μA/V2, L = 1 μm, and W = 10 μm. Find V2 and I2. How do these values change if Q3 and Q4 are made to have W = 100 μm? ϩ5 V Q4 I2 V2 Q3 Q2 Q1 Figure P5.60 5.61 In the circuit of Fig. P5.61, transistors Q1 and Q2 have Vt = 0.7 V, and the process transconductance parameter kn = 125 μA/V2. Find V1, V2, and V3 for each of the following cases: (a) (W/L)1 = (W/L)2 = 20 (b) (W/L)1 = 1.5(W/L)2 = 20 ϩ2.5 V 20 k⍀ V1 Q1 20 k⍀ V2 Q2 Section 5.4: The Body Effect and Other Topics 5.62 In a particular application, an n-channel MOSFET operates with VSB in the range 0 V to 4 V. If Vt0 is nominally 1.0 V, find the range of Vt that results if γ = 0.5 V1/2 and 2φf = 0.6 V. If the gate oxide thickness is increased by a factor of 4, what does the threshold voltage become? 5.63 A p-channel transistor operates in saturation with its source voltage 3 V lower than its substrate. For γ = 0.5 V1/2, 2φf = 0.75 V, and Vt0 = −0.7 V, find Vt. *5.64 (a) Using the expression for iD in saturation and neglecting the channel-length modulation effect (i.e., let λ = 0), derive an expression for the per unit change in iD per °C ∂iD/iD /∂T in terms of the per unit change in kn per °C ∂kn/kn /∂T , the temperature coefficient of Vt in V/°C ∂Vt/∂T , and VGS and Vt. (b) If Vt decreases by 2 mV for every °C rise in temperature, find the temperature coefficient of kn that results in iD decreasing by 0.2%/°C when the NMOS transistor with Vt = 1 V is operated at VGS = 5 V. 5.65 A depletion-type n-channel MOSFET with knW/L = 2 mA/V2 and Vt = −3 V has its source and gate grounded. Find the region of operation and the drain current for vD = 0.1 V, 1 V, 3 V, and 5 V. Neglect the channel-length-modulation effect. 5.66 For a particular depletion-mode NMOS device, Vt = −2 V, knW/L = 200 μA/V2, and λ = 0.02 V−1. When operated at vGS = 0, what is the drain current that flows for vDS = 1 V, 2 V, 3 V, and 10 V? What does each of these currents become if the device width is doubled with L the same? With L also doubled? *5.67 Neglecting the channel-length-modulation effect, show that for the depletion-type NMOS transistor of Fig. P5.67, the i−v relationship is given by i = 1 2 kn (W/L) v2 − 2Vtv i = − 1 2 kn (W/L)Vt2 for v ≥ Vt for v ≤ Vt (Recall that Vt is negative.) Sketch the i−v relationship for the case: Vt = −2 V and kn(W/L) = 2 mA/V2. V3 iϩ 200 ␮A v Figure P5.61 Ϫ Figure P5.67 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 6 Bipolar Junction Transistors (BJTs) Introduction 305 6.1 Device Structure and Physical Operation 306 6.2 Current–Voltage Characteristics 320 6.3 BJT Circuits at DC 333 6.4 Transistor Breakdown and Temperature Effects 351 Summary 354 Problems 355 IN THIS CHAPTER YOU WILL LEARN 1. The physical structure of the bipolar transistor and how it works. 2. How the voltage between two terminals of the transistor controls the current that flows through the third terminal, and the equations that describe these current–voltage characteristics. 3. How to analyze and design circuits that contain bipolar transistors, resistors, and dc sources. Introduction In this chapter, we study the other major three-terminal device: the bipolar junction transistor (BJT). The presentation of the material in this chapter parallels but does not rely on that for the MOSFET in Chapter 5; thus, if desired, the BJT can be studied before the MOSFET. Three-terminal devices are far more useful than two-terminal ones, such as the diodes studied in Chapter 4, because they can be used in a multitude of applications, ranging from signal amplification to the design of digital logic and memory circuits. The basic principle involved is the use of the voltage between two terminals to control the current flowing in the third terminal. In this way, a three-terminal device can be used to realize a controlled source, which as we learned in Chapter 1 is the basis for amplifier design. Also, in the extreme, the control signal can be used to cause the current in the third terminal to change from zero to a large value, thus allowing the device to act as a switch. The switch is the basis for the realization of the logic inverter, the basic element of digital circuits. The invention of the BJT in 1948 at the Bell Telephone Laboratories ushered in the era of solid-state circuits. The result was not just the replacement of vacuum tubes by transistors in radios and television sets but the eruption of an electronics revolution that led to major changes in the way we work, play, and indeed, live. The invention of the transistor also eventually led to the dominance of information technology and the emergence of the knowledge-based economy. The bipolar transistor enjoyed nearly three decades as the device of choice in the design of both discrete and integrated circuits. Although the MOSFET had been known very early on, it was not until the 1970s and 1980s that it became a serious competitor to the BJT. By 2014, the MOSFET was undoubtedly the most widely used electronic device, and CMOS technology the technology of choice in the design of integrated circuits. Nevertheless, the BJT remains a significant device that excels in certain applications. The BJT remains popular in discrete-circuit design, where it is used together with other discrete components such as resistors and capacitors to implement circuits that are assembled 305 306 Chapter 6 Bipolar Junction Transistors (BJTs) on printed-circuit boards (PCBs). Here we note the availability of a very wide selection of BJT types that fit nearly every conceivable application. As well, the BJT is still the preferred device in some very demanding analog and digital integrated-circuit applications. This is especially true in very-high-frequency and high-speed circuits. In particular, a very-high-speed digital logic-circuit family based on bipolar transistors, namely, emitter-coupled logic, is still in use (Chapter 15). Finally, bipolar transistors can be combined with MOSFETs to create innovative circuits that take advantage of the high-input-impedance and low-power operation of MOSFETs and the very-high-frequency operation and high-current-driving capability of bipolar transistors. The resulting technology is known as BiCMOS, and it is finding increasingly larger areas of application (see Chapters 8, 9, 13, and 15). In this chapter, we shall start with a description of the physical operation of the BJT. Though simple, this physical description provides considerable insight regarding the performance of the transistor as a circuit element. We then quickly move from describing current flow in terms of electrons and holes to a study of the transistor terminal characteristics. Circuit models for transistor operation in different modes will be developed and utilized in the analysis and design of transistor circuits. The main objective of this chapter is to develop in the reader a high degree of familiarity with the BJT. Thus, it lays the foundation for the use of the BJT in amplifier design (Chapter 7). 6.1 Device Structure and Physical Operation 6.1.1 Simplified Structure and Modes of Operation Figure 6.1 shows a simplified structure for the BJT. A practical transistor structure will be shown later (see also Appendix A, which deals with fabrication technology). As shown in Fig. 6.1, the BJT consists of three semiconductor regions: the emitter region (n type), the base region ( p type), and the collector region (n type). Such a transistor is called an npn transistor. Another transistor, a dual of the npn as shown in Fig. 6.2, has a p-type emitter, an n-type base, and a p-type collector, and is appropriately called a pnp transistor. A terminal is connected to each of the three semiconductor regions of the transistor, with the terminals labeled emitter (E), base (B), and collector (C). The transistor consists of two pn junctions, the emitter–base junction (EBJ) and the collector–base junction (CBJ). Depending on the bias condition (forward or reverse) of each of these junctions, different modes of operation of the BJT are obtained, as shown in Table 6.1. The active mode is the one used if the transistor is to operate as an amplifier. Switching applications (e.g., logic circuits) utilize both the cutoff mode and the saturation mode. As the name implies, in the cutoff mode no current flows because both junctions are reverse biased. As we will see shortly, charge carriers of both polarities—that is, electrons and holes—participate in the current-conduction process in a bipolar transistor, which is the reason for the name bipolar.1 1This should be contrasted with the situation in the MOSFET, where current is conducted by charge carriers of one type only: electrons in n-channel devices or holes in p-channel devices. In earlier days, some referred to FETs as unipolar devices. Emitter (E) n-type Emitter region p-type Base region 6.1 Device Structure and Physical Operation 307 n-type Collector region Metal contact Collector (C) Emitter–base junction (EBJ) Base (B) Figure 6.1 A simplified structure of the npn transistor. Collector–base junction (CBJ) Metal contact p n p E Emitter Base Collector C region region region B Figure 6.2 A simplified structure of the pnp transistor. Table 6.1 BJT Modes of Operation Mode Cutoff Active Saturation EBJ Reverse Forward Forward CBJ Reverse Reverse Forward 6.1.2 Operation of the npn Transistor in the Active Mode Of the three modes of operation of the BJT, the active mode is the most important. Therefore, we begin our study of the BJT by considering its physical operation in the active mode.2 This situation is illustrated in Fig. 6.3 for the npn transistor. Two external voltage sources (shown as batteries) are used to establish the required bias conditions for active-mode operation. The voltage VBE causes the p-type base to be higher in potential than the n-type emitter, thus forward biasing the emitter–base junction. The collector–base voltage VCB causes the n-type collector to be at a higher potential than the p-type base, thus reverse biasing the collector–base junction. 2The material in this section assumes that the reader is familiar with the operation of the pn junction under forward-bias conditions (Section 3.5). 308 Chapter 6 Bipolar Junction Transistors (BJTs) Forward-biased Reverse-biased E iE n p Injected electrons Diffusing electrons iE iB2 Injected holes (iB1) iB iB –+ vBE iE iE B n Collected electrons iC C iC Recombined electrons (iB2) –+ vCB iC iC VBE VCB Figure 6.3 Current flow in an npn transistor biased to operate in the active mode. (Reverse current components due to drift of thermally generated minority carriers are not shown.) Current Flow The forward bias on the emitter–base junction will cause current to flow across this junction. Current will consist of two components: electrons injected from the emitter into the base, and holes injected from the base into the emitter. As will become apparent shortly, it is highly desirable to have the first component (electrons from emitter to base) be much larger than the second component (holes from base to emitter). This can be accomplished by fabricating the device with a heavily doped emitter and a lightly doped base; that is, the device is designed to have a high density of electrons in the emitter and a low density of holes in the base. The current that flows across the emitter–base junction will constitute the emitter current iE, as indicated in Fig. 6.3. The direction of iE is “out of ” the emitter lead, which, following the usual conventions, is in the direction of the positive-charge flow (hole current) and opposite to the direction of the negative-charge flow (electron current), with the emitter current iE being equal to the sum of these two components. However, since the electron component is much larger than the hole component, the emitter current will be dominated by the electron component. From our study in Section 3.5 of the current flow across a forward-biased pn junction, we know that the magnitude of both the electron component and the hole component of iE will be proportional to evBE /VT , where vBE is the forward voltage across the base–emitter junction and VT is the thermal voltage (approximately 25 mV at room temperature). Let’s now focus our attention on the first current component, namely, that carried by electrons injected from the emitter into the base. These electrons will be minority carriers in the p-type base region. Because their concentration will be highest at the emitter side of the base, the injected electrons will diffuse through the base region toward the collector. In their journey across the base, some of the electrons will combine with holes, which are majority carriers in the base. However, since the base is usually very thin and, as mentioned earlier, lightly doped, the proportion of electrons that are “lost” through this recombination process will be quite small. Thus, most of the diffusing electrons will reach the boundary of the collector–base depletion region. Because the collector is more positive than the base (by the 6.1 Device Structure and Physical Operation 309 reverse-bias voltage vCB), these successful electrons will be swept across the CBJ depletion region into the collector. They will thus get collected and constitute the collector current iC. The Collector Current From the foregoing statements, we see that the collector current is carried by the electrons that reach the collector region. Its direction will be opposite to that of the flow of electrons, and thus into the collector terminal. Its magnitude will be proportional to evBE /VT , thus iC = I evBE /VT S (6.1) where the constant of proportionality IS, as in the case of the diode, is called the saturation current and is a transistor parameter. We will have more to say about IS shortly. An important observation to make here is that iC is independent of the value of vCB. That is, as long as the collector is positive with respect to the base, the electrons that reach the collector side of the base region will be swept into the collector and will register as collector current. The Base Current Reference to Fig. 6.3 shows that the base current iB is composed of two components. The first component iB1 is due to the holes injected from the base region into the emitter region. This current component is proportional to evBE /VT . The second component of base current, iB2, is due to holes that have to be supplied by the external circuit in order to replace the holes lost from the base through the recombination process. Because iB2 is proportional to the number of electrons injected into the base, it also will be proportional to evBE /VT . Thus the total base current, iB = iB1 + iB2, will be proportional to evBE /VT , and can be expressed as a fraction of the collector current iC as follows: iB = iC β (6.2) That is, iB = I e S vBE /VT β (6.3) where β is a transistor parameter. For modern npn transistors, β is in the range 50 to 200, but it can be as high as 1000 for special devices. For reasons that will become clear later, the parameter β is called the common-emitter current gain. The above description indicates that the value of β is highly influenced by two factors: the width of the base region, W, and the relative dopings of the base region and the emitter region, NA/ND. To obtain a high β (which is highly desirable since β represents a gain parameter) the base should be thin (W small) and lightly doped and the emitter heavily doped (making NA/ND small). For modern integrated circuit fabrication technologies, W is in the nanometer range. The Emitter Current Since the current that enters a transistor must leave it, it can be seen from Fig. 6.3 that the emitter current iE is equal to the sum of the collector current iC and the base current iB; that is, iE = iC + iB (6.4) 310 Chapter 6 Bipolar Junction Transistors (BJTs) Use of Eqs. (6.2) and (6.4) gives β+1 iE = β iC That is, iE = β + β 1 IS evBE /VT Alternatively, we can express Eq. (6.5) in the form iC = αiE where the constant α is related to β by α = β β + 1 Thus the emitter current in Eq. (6.6) can be written iE = (IS/α)evBE /VT Finally, we can use Eq. (6.8) to express β in terms of α, that is, β = 1 α − α (6.5) (6.6) (6.7) (6.8) (6.9) (6.10) It can be seen from Eq. (6.8) that α is a constant (for a particular transistor) that is less than but very close to unity. For instance, if β = 100, then α 0.99. Equation (6.10) reveals an important fact: Small changes in α correspond to very large changes in β. This mathematical observation manifests itself physically, with the result that transistors of the same type may have widely different values of β. For reasons that will become apparent later, α is called the common-base current gain. Minority-Carrier Distribution Our understanding of the physical operation of the BJT can be enhanced by considering the distribution of minority charge carriers in the base and the emitter. Figure 6.4 shows the profiles of the concentration of electrons in the base and holes in the emitter of an npn transistor operating in the active mode. Observe that since the doping concentration in the emitter, ND, is much higher than the doping concentration in the base, NA, the concentration of electrons injected from emitter to base, np(0), is much higher than the concentration of holes injected from the base to the emitter, pn(0). Both quantities are proportional to evBE /VT , thus np(0) = np0 evBE /VT (6.11) where np0 is the thermal-equilibrium value of the minority-carrier (electron) concentration in the base region. Next, observe that because the base is very thin, the concentration of excess electrons decays almost linearly (as opposed to the usual exponential decay, as observed for the excess holes in the emitter region). Furthermore, the reverse bias on the collector–base junction causes the concentration of excess electrons at the collector side of the base to be zero. (Recall that electrons that reach that point are swept into the collector.) The tapered minority-carrier concentration profile (Fig. 6.4) causes the electrons injected into the base to diffuse through the base region toward the collector. This electron diffusion 6.1 Device Structure and Physical Operation 311 Carrier concentration Emitter (n) EBJ depletion region Hole np (0) concentration Base (p) Electron concentration np (ideal) CBJ depletion region Collector (n) pn (0) np (with pn0 recombination) Effective base width W Distance (x) Figure 6.4 Profiles of minority-carrier concentrations in the base and in the emitter of an npn transistor operating in the active mode: vBE > 0 and vCB ≥ 0. current In is directly proportional to the slope of the straight-line concentration profile, In = AE qDn dnp(x) dx = AEqDn − np(0) W (6.12) where AE is the cross-sectional area of the base–emitter junction (in the direction perpendicular to the page), q is the magnitude of the electron charge, Dn is the electron diffusivity in the base, and W is the effective width of the base. Observe that the negative slope of the minority-carrier concentration results in a negative current In across the base; that is, In flows from right to left (in the negative direction of x), which corresponds to the usual convention, namely, opposite to the direction of electron flow. The recombination in the base region, though slight, causes the excess minority-carrier concentration profile to deviate from a straight line and take the slightly concave shape indicated by the broken line in Fig. 6.4. The slope of the concentration profile at the EBJ is slightly higher than that at the CBJ, with the difference accounting for the small number of electrons lost in the base region through recombination. Finally, we have the collector current iC = In, which will yield a negative value for iC, indicating that iC flows in the negative direction of the x axis (i.e., from right to left). Since we will take this to be the positive direction of iC, we can drop the negative sign in Eq. (6.12). Doing this and substituting for np (0) from Eq. (6.11), we can thus express the collector current iC as iC = I evBE /VT S where the saturation current IS is given by IS = AEqDnnp0/W 312 Chapter 6 Bipolar Junction Transistors (BJTs) Substituting np0 = ni2/NA, where ni is the intrinsic carrier density and NA is the doping concentration in the base, we can express IS as IS = AE qDnni2 NAW (6.13) The saturation current IS is inversely proportional to the base width W and is directly proportional to the area of the EBJ. Typically IS is in the range of 10−12 A to 10−18 A (depending on the size of the device). Because IS is proportional to ni2, it is a strong function of temperature, approximately doubling for every 5°C rise in temperature. (For the dependence of ni2 on temperature, refer to Eq. 3.2.) Since IS is directly proportional to the junction area (i.e., the device size), it will also be referred to as the scale current. Two transistors that are identical except that one has an EBJ area, say, twice that of the other will have saturation currents with that same ratio (i.e., 2). Thus for the same value of vBE the larger device will have a collector current twice that in the smaller device. This concept is frequently employed in integrated-circuit design. Recapitulation and Equivalent-Circuit Models We have presented a first-order model for the operation of the npn transistor in the active mode. Basically, the forward-bias voltage vBE causes an exponentially related current iC to flow in the collector terminal. The collector current iC is independent of the value of the collector voltage as long as the collector–base junction remains reverse biased; that is, vCB ≥ 0. Thus in the active mode the collector terminal behaves as an ideal constant-current source where the value of the current is determined by vBE. The base current iB is a factor 1/β of the collector current, and the emitter current is equal to the sum of the collector and base currents. Since iB is much smaller than iC (i.e., β 1), iE iC. More precisely, the collector current is a fraction α of the emitter current, with α smaller than, but close to, unity. This first-order model of transistor operation in the active mode can be represented by the equivalent circuit shown in Fig. 6.5(a). Here, diode DE has a scale current ISE equal to (IS/α) and thus provides a current iE related to vBE according to Eq. (6.9). The current of the controlled source, which is equal to the collector current, is controlled by vBE according to the exponential relationship indicated, a restatement of Eq. (6.1). This model is in essence a nonlinear voltage-controlled current source. It can be converted to the current-controlled current-source model shown in Fig. 6.5(b) by expressing the current of the controlled source as αiE. Note that this model is also nonlinear because of the exponential relationship of the current iE through diode DE and the voltage vBE. From this model we observe that if the transistor is used as a two-port network with the input port between E and B and the output port between C and B (i.e., with B as a common terminal), then the current gain observed is equal to α. Thus α is called the common-base current gain. Two other equivalent-circuit models, shown in Fig. 6.5(c) and (d), may be used to represent the operation of the BJT. The model of Fig. 6.5(c) is essentially a voltage-controlled current source. However, here diode DB conducts the base current and thus its current scale factor is IS/β, resulting in the iB–vBE relationship given in Eq. (6.3). By simply expressing the collector current as βiB we obtain the current-controlled current-source model shown in Fig. 6.5(d). From this latter model we observe that if the transistor is used as a two-port network with the input port between B and E and the output port between C and E (i.e., with E as the common terminal), then the current gain observed is equal to β. Thus β is called the common-emitter current gain. 6.1 Device Structure and Physical Operation 313 Figure 6.5 Large-signal equivalent-circuit models of the npn BJT operating in the forward active mode. Finally, we note that the models in Fig. 6.5 apply for any positive value of vBE. That is, unlike the models we will be discussing in Chapter 7, here there is no limitation on the size of vBE, and thus these models are referred to as large-signal models. Example 6.1 An npn transistor having IS = 10−15A and β = 100 is connected as follows: The emitter is grounded, the base is fed with a constant-current source supplying a dc current of 10 μA, and the collector is connected to a 5-V dc supply via a resistance RC of 3 k . Assuming that the transistor is operating in the active mode, find VBE and VCE. Use these values to verify active-mode operation. Replace the current source with a resistance connected from the base to the 5-V dc supply. What resistance value is needed to result in the same operating conditions? 314 Chapter 6 Bipolar Junction Transistors (BJTs) Example 6.1 continued Solution If the transistor is operating in the active mode, it can be represented by one of the four possible equivalent-circuit models shown in Fig. 6.5. Because the emitter is grounded, either the model in Fig. 6.5(c) or that in Fig. 6.5(d) would be suitable. Since we know the base current IB, the model of Fig. 6.5(d) is the most suitable. 10 A IB VCC 5V RC 3 k B C RB IC IB VCC 5V B RC C IC DB VBE bIB DB VCE VBE bIB VCE E E (a) (b) Figure 6.6 Circuits for Example 6.1. Figure 6.6(a) shows the circuit as described with the transistor represented by the model of Fig. 6.5(d). We can determine VBE from the exponential characteristic of DB as follows: VBE = VT ln IB IS /β = 25 ln 10 × 10−6 10−17 = 690 mV = 0.69 V Next we determine the value of VCE from VCE = VCC − RC IC where IC = βIB = 100 × 10 × 10−6 = 10−3 A = 1 mA 6.1 Device Structure and Physical Operation 315 Thus, VCE = 5 − 3 × 1 = +2 V Since VC at +2 V is higher than VB at 0.69 V, the transistor is indeed operating in the active mode. Now, replacing the 10-μA current source with a resistance RB connected from the base to the 5-V dc supply VCC, as in Fig. 6.6(b), the value of RB must be RB = VCC − VBE IB = 5 − 0.69 10 μA = 431 k EXERCISES 6.1 Consider an npn transistor with vBE = 0.7 V at iC = 1 mA. Find vBE at iC = 0.1 mA and 10 mA. Ans. 0.64 V; 0.76 V 6.2 Transistors of a certain type are specified to have β values in the range of 50 to 150. Find the range of their α values. Ans. 0.980 to 0.993 6.3 Measurement of an npn BJT in a particular circuit shows the base current to be 14.46 μA, the emitter current to be 1.460 mA, and the base–emitter voltage to be 0.7 V. For these conditions, calculate α, β, and IS. Ans. 0.99; 100; 10−15 A 6.4 Calculate β for two transistors for which α = 0.99 and 0.98. For collector currents of 10 mA, find the base current of each transistor. Ans. 99; 49; 0.1 mA; 0.2 mA 6.5 A transistor for which IS = 10−16 A and β = 100 is conducting a collector current of 1 mA. Find vBE. Also, find ISE and ISB for this transistor. Ans. 747.5 mV; 1.01 × 10−16 A; 10−18A 6.6 For the circuit in Fig. 6.6(a) analyzed in Example 6.1, find the maximum value of RC that will still result in active-mode operation. Ans. 4.31 k 6.1.3 Structure of Actual Transistors Figure 6.7 shows a more realistic (but still simplified) cross section of an npn BJT. Note that the collector virtually surrounds the emitter region, thus making it difficult for the electrons injected into the thin base to escape being collected. In this way, the resulting α is close to 316 Chapter 6 Bipolar Junction Transistors (BJTs) E B C p n n Figure 6.7 Cross section of an npn BJT. unity and β is large. Also, observe that the device is not symmetrical, and thus the emitter and collector cannot be interchanged.3 For more detail on the physical structure of actual devices, the reader is referred to Appendix A. The structure in Fig. 6.7 indicates also that the CBJ has a much larger area than the EBJ. Thus the CB diode DC has a saturation current ISC that is much larger than the saturation current of the EB diode DE. Typically, ISC is 10 to 100 times larger than ISE (recall that ISE = IS/α IS). EXERCISE 6.7 A particular transistor has IS = 10−15 A and α the collector scale current ISC. Ans. 10−13 A 1. If the CBJ area is 100 times the area of the EBJ, find 6.1.4 Operation in the Saturation Mode4 As mentioned above, for the BJT to operate in the active mode, the CBJ must be reverse biased. Thus far, we have stated this condition for the npn transistor as vCB ≥ 0. However, we know that a pn junction does not effectively become forward biased until the forward voltage across it exceeds approximately 0.4 V. It follows that one can maintain active-mode operation of an npn transistor for negative vCB down to approximately −0.4 V. This is illustrated in Fig. 6.8, which is a sketch of iC versus vCB for an npn transistor operated with a constant emitter current IE. As expected, iC is independent of vCB in the active mode, a situation that extends 3If the emitter and collector are reversed—that is, the CBJ is forward biased and the EBJ is reverse biased—the device operates in a mode called the “reverse-active mode.” The resulting values of α and β, denoted αR and βR (with R denoting reverse), are much lower than the values of α and β, respectively, obtained in the “forward”-active mode discussed above. Hence, the reverse-active mode has no practical application. The MOSFET, on the other hand, being a perfectly symmetrical device, can operate equally well with its drain and source terminals interchanged. 4Saturation means something completely different in a BJT and in a MOSFET. The saturation mode of operation of the BJT is analogous to the triode region of operation of the MOSFET. On the other hand, the saturation region of operation of the MOSFET corresponds to the active mode of BJT operation. Saturation mode iC aIE Active mode 6.1 Device Structure and Physical Operation 317 iE IE 0.4 V 0 vCB Expanded scale Figure 6.8 The iC−vCB characteristic of an npn transistor fed with a constant emitter current IE. The transistor enters the saturation mode of operation for vCB < – 0.4 V, and the collector current diminishes. iB / ISC evBC VT iC B DC vBE DB E C / ISevBE VT Figure 6.9 Modeling the operation of an npn transistor in saturation by augmenting the model of Fig. 6.5(c) with a forward-conducting diode DC. Note that the current through DC increases iB and reduces iC. for vCB going negative to approximately −0.4 V. Below this value of vCB, the CBJ begins to conduct sufficiently that the transistor leaves the active mode and enters the saturation mode of operation, where iC decreases. To see why iC decreases in saturation, we can construct a model for the saturated npn transistor as follows. We augment the model of Fig. 6.5(c) with the forward-conducting CBJ diode DC, as shown in Fig. 6.9. Observe that the current iBC will subtract from the controlled-source current, resulting in the reduced collector current iC given by iC = IS evBE /VT − I evBC /VT SC (6.14) where ISC is the saturation current for DC and is related to IS by the ratio of the areas of the CBJ and the EBJ. The second term in Eq. (6.14) will play an increasing role as vBC exceeds 0.4 V or so, causing iC to decrease and eventually reach zero. Figure 6.9 also indicates that in saturation the base current will increase to the value iB = (IS/β)evBE /VT + I evBC /VT SC (6.15) Equations (6.14) and (6.15) can be combined to obtain the ratio iC/iB for a saturated transistor. We observe that this ratio will be lower than the value of β. Furthermore, the ratio will decrease as vBC is increased and the transistor is driven deeper into saturation. Because iC/iB 318 Chapter 6 Bipolar Junction Transistors (BJTs) of a saturated transistor can be set to any desired value lower than β by adjusting vBC, this ratio is known as forced β and denoted βforced, β forced = iC iB ≤β saturation (6.16) As will be shown later, in analyzing a circuit we can determine whether the BJT is in the saturation mode by either of the following two tests: 1. Is the CBJ forward biased by more than 0.4 V? 2. Is the ratio iC/iB lower than β? The collector-to-emitter voltage vCE of a saturated transistor can be found from Fig. 6.9 as the difference between the forward-bias voltages of the EBJ and the CBJ, VCEsat = VBE − VBC (6.17) Recalling that the CBJ has a much larger area than the EBJ, VBC will be smaller than VBE by 0.1 to 0.3 V. Thus, VCEsat 0.1 to 0.3 V Typically we will assume that a transistor at the edge of saturation has VCEsat = 0.3 V, while a transistor deep in saturation has VCEsat = 0.2 V. EXERCISES 6.8 Use Eq. (6.14) to show that iC reaches zero at VCE = VT ln ISC /IS Calculate VCE for a transistor whose CBJ has 100 times the area of the EBJ. Ans. 115 mV 6.9 Use Eqs. (6.14), (6.15), and (6.16) to show that a BJT operating in saturation with VCE = VCEsat has a forced β given by e − I VCEsat /VT /I β forced =β eVCEsat /VT SC S + βISC/IS Find βforced for β = 100, ISC/IS = 100, and VCEsat = 0.2 V. Ans. 22.2 6.1.5 The pnp Transistor The pnp transistor operates in a manner similar to that of the npn device described above. Figure 6.10 shows a pnp transistor biased to operate in the active mode. Here the voltage VEB causes the p-type emitter to be higher in potential than the n-type base, thus forward biasing the emitter–base junction. The collector–base junction is reverse biased by the voltage VBC, which keeps the p-type collector lower in potential than the n-type base. 6.1 Device Structure and Physical Operation 319 Forward biased Reverse biased p n p iE Injected holes Diffusing holes Collected holes iC E iE Injected iB2 electrons iB1 Recombined holes iB +– vEB iB +– vBC iE iE B iC VEB VBC Figure 6.10 Current flow in a pnp transistor biased to operate in the active mode. iC C iC Unlike the npn transistor, current in the pnp device is mainly conducted by holes injected from the emitter into the base as a result of the forward-bias voltage VEB. Since the component of emitter current contributed by electrons injected from base to emitter is kept small by using a lightly doped base, most of the emitter current will be due to holes. The electrons injected from base to emitter give rise to the first component of base current, iB1. Also, a number of the holes injected into the base will recombine with the majority carriers in the base (electrons) and will thus be lost. The disappearing base electrons will have to be replaced from the external circuit, giving rise to the second component of base current, iB2. The holes that succeed in reaching the boundary of the depletion region of the collector–base junction will be attracted by the negative voltage on the collector. Thus these holes will be swept across the depletion region into the collector and appear as collector current. It can easily be seen from the above description that the current–voltage relationship of the pnp transistor will be identical to that of the npn transistor except that vBE has to be replaced by vEB. Also, the large-signal, active-mode operation of the pnp transistor can be modeled by any of four equivalent circuits similar to those for the npn transistor in Fig. 6.5. Two of these four circuits are shown in Fig. 6.11. Finally, we note that the pnp transistor can operate in the saturation mode in a manner analogous to that described for the npn device. EXERCISES 6.10 Consider the model in Fig. 6.11(a) applied in the case of a pnp transistor whose base is grounded, the emitter is fed by a constant-current source that supplies a 2-mA current into the emitter terminal, and the collector is connected to a –10-V dc supply. Find the emitter voltage, the base current, and the collector current if for this transistor β = 50 and IS = 10−14 A. Ans. 0.650 V; 39.2 μA; 1.96 mA 6.11 For a pnp transistor having IS = 10−11 A and β = 100, calculate vEB for iC = 1.5 A. Ans. 0.643 V 320 Chapter 6 Bipolar Junction Transistors (BJTs) iE D iB (IS a) iE E IS evEB VT iC vEB B iB DB (IS b) IS e vEB VT C iC (a) (b) Figure 6.11 Two large-signal models for the pnp transistor operating in the active mode. THE INVENTION OF THE BJT: The first working transistor was demonstrated at the Bell Labs in late 1947 by John Bardeen and Walter Brattain, who were part of a team led by William Shockley. Made of germanium, the device became known as a point-contact transistor and operated on the field-effect principle. Within a few weeks, however, Shockley wrote a complete description of the bipolar junction transistor (BJT) and filed for a U.S. patent with the title “Circuit Element Utilizing Semiconductor Material.” BJTs dominated the electronics world from the early 1950s to the mid-1970s, when MOSFETs took over the leading position. In 1956, Shockley, Bardeen, and Brattain shared the Nobel Prize in Physics for the discovery of the transistor effect. 6.2 Current–Voltage Characteristics 6.2.1 Circuit Symbols and Conventions The physical structure used thus far to explain transistor operation is rather cumbersome to employ in drawing the schematic of a multitransistor circuit. Fortunately, a very descriptive and convenient circuit symbol exists for the BJT. Figure 6.12(a) shows the symbol for the npn transistor; the pnp symbol is given in Fig. 6.12(b). In both symbols the emitter is distinguished by an arrowhead. This distinction is important because, as we have seen in the last section, practical BJTs are not symmetric devices. The polarity of the device—npn or pnp—is indicated by the direction of the arrowhead on the emitter. This arrowhead points in the direction of normal current flow in the emitter, which is also the forward direction of the base–emitter junction. Since we have adopted a drawing convention by which currents flow from top to bottom, we will always draw pnp transistors in the manner shown in Fig. 6.12(b) (i.e., with their emitters on top). Figure 6.13 shows npn and pnp transistors connected to dc sources so as to operate in the active mode. Figure 6.13 also indicates the reference and actual directions of current flow throughout the transistor. Our convention will be to take the reference direction to coincide 6.2 Current–Voltage Characteristics 321 npn pnp (a) (b) Figure 6.12 Circuit symbols for BJTs. Figure 6.13 Voltage polarities and current flow in transistors operating in (a) (b) the active mode. with the normal direction of current flow. Hence, normally, we should not encounter a negative value for iE, iB, or iC. The convenience of the circuit-drawing convention that we have adopted should be obvious from Fig. 6.13. Note that currents flow from top to bottom and that voltages are higher at the top and lower at the bottom. The arrowhead on the emitter also implies the polarity of the emitter–base voltage that should be applied in order to forward bias the emitter–base junction. Just a glance at the circuit symbol of the pnp transistor, for example, indicates that we should make the emitter higher in voltage than the base (by vEB) in order to cause current to flow into the emitter (downward). Note that the symbol vEB means the voltage by which the emitter (E) is higher than the base (B). Thus for a pnp transistor operating in the active mode vEB is positive, while in an npn transistor vBE is positive. From the discussion of Section 6.1 it follows that an npn transistor whose EBJ is forward biased (usually, VBE 0.7 V) will operate in the active mode as long as the collector voltage does not fall below that of the base by more than approximately 0.4 V. Otherwise, the transistor leaves the active mode and enters the saturation region of operation.5 In a parallel manner, the pnp transistor will operate in the active mode if the EBJ is forward biased (usually, VEB 0.7 V) and the collector voltage is not allowed to rise above that of the base by more than 0.4 V or so. Otherwise, the CBJ becomes forward biased, and the pnp transistor enters the saturation region of operation. 5It is interesting to contrast the active-mode operation of the BJT with the corresponding mode of operation of the MOSFET: The BJT needs a minimum vCE of about 0.3 V, and the MOSFET needs a minimum vDS equal to VOV , which for modern technologies is in the range of 0.2 V to 0.3 V. Thus we see a great deal of similarity! Also note that reverse biasing the CBJ of the BJT corresponds to pinching off the channel of the MOSFET. This condition results in the collector current (drain current in the MOSFET) being independent of the collector voltage (the drain voltage in the MOSFET). 322 Chapter 6 Bipolar Junction Transistors (BJTs) E B ϩ 0.7 V Ϫ Active 0.4 V C 0.3 V Saturation ϩ 0.7 V Ϫ B 0.3 V Saturation C 0.4 V Active E (a) npn (b) pnp Figure 6.14 Graphical representation of the conditions for operating the BJT in the active mode and in the saturation mode. Table 6.2 Summary of the BJT Current–Voltage Relationships in the Active Mode iC = IS evBE /VT iB = iC β = IS β evBE /VT iE = iC α = IS α evBE /VT Note: For the pnp transistor, replace vBE with vEB. iC = αiE iB = (1 − α)iE = iE β +1 iC = βiB α β = 1−α iE = (β + 1)iB β α= β+1 VT = thermal voltage = kT q 25 mV at room temperature For greater emphasis, we show in Fig. 6.14 a graphical construction that illustrates the conditions for operating the BJT in the active mode and in the saturation mode. Also, for easy reference, we present in Table 6.2 a summary of the BJT current–voltage relationships in the active mode of operation. The Collector–Base Reverse Current (ICBO) In our discussion of current flow in transistors we ignored the small reverse currents carried by thermally generated minority carriers. Although such currents can be safely neglected in modern transistors, the reverse current across the collector–base junction deserves some mention. This current, denoted ICBO, is the reverse current flowing from collector to base with the emitter open-circuited (hence the subscript O). This current is usually in the nanoampere range, a value that is many times higher than its theoretically predicted value. As with the diode reverse current, ICBO contains a substantial leakage component, and its value is dependent on vCB. ICBO depends strongly on temperature, approximately doubling for every 10°C rise.6 6 The temperature coefficient of ICBO is different from that of IS because ICBO contains a substantial leakage component. 6.2 Current–Voltage Characteristics 323 Example 6.2 The transistor in the circuit of Fig. 6.15(a) has β = 100 and exhibits a vBE of 0.7 V at iC = 1 mA. Design the circuit so that a current of 2 mA flows through the collector and a voltage of +5 V appears at the collector. 15 V 15 V RC IC 2 mA RC 2 IB 0.02 mA VC 5V VBE IE IC IB VE VBE RE 2.02 mA RE 15 V (a) 15 V (b) Figure 6.15 Circuit for Example 6.2. Solution Refer to Fig. 6.15(b). We note at the outset that since we are required to design for VC = +5 V, the CBJ will be reverse biased and the BJT will be operating in the active mode. To obtain a voltage VC = +5 V, the voltage drop across RC must be 15 – 5 = 10 V. Now, since IC = 2 mA, the value of RC should be selected according to RC = 10 V 2 mA = 5 k Since vBE = 0.7 V at iC = 1 mA, the value of vBE at iC = 2 mA is 2 VBE = 0.7 + VT ln 1 = 0.717 V Since the base is at 0 V, the emitter voltage should be VE = −0.717 V For β = 100, α = 100/101 = 0.99. Thus the emitter current should be IE = IC α = 2 0.99 = 2.02 mA 324 Chapter 6 Bipolar Junction Transistors (BJTs) Example 6.2 continued Now the value required for RE can be determined from RE = VE −(−15) IE = −0.717 + 15 = 7.07 k 2.02 This completes the design. We should note, however, that the calculations above were made with a degree of precision that is usually neither necessary nor justified in practice in view, for instance, of the expected tolerances of component values. Nevertheless, we chose to do the design precisely in order to illustrate the various steps involved. EXERCISES D6.12 Repeat Example 6.2 for a transistor fabricated in a modern integrated-circuit process. Such a process yields devices that exhibit larger vBE at the same iC because they have much smaller junction areas. The dc power supplies utilized in modern IC technologies fall in the range of 1 V to 3 V. Design a circuit similar to that shown in Fig. 6.15 except that now the power supplies are ±1.5 V and the BJT has β = 100 and exhibits vBE of 0.8 V at iC = 1 mA. Design the circuit so that a current of 2 mA flows through the collector and a voltage of +0.5 V appears at the collector. Ans. RC = 500 ; RE = 338 6.13 In the circuit shown in Fig. E6.13, the voltage at the emitter was measured and found to be –0.7 V. If β = 50, find IE, IB, IC, and VC. 10 V 5k IB IC VC IE VE 10 k 10 V Figure E6.13 Ans. 0.93 mA; 18.2 μA; 0.91 mA; +5.45 V 6.2 Current–Voltage Characteristics 325 6.14 In the circuit shown in Fig. E6.14, measurement indicates VB to be +1.0 V and VE to be +1.7 V. What are α and β for this transistor? What voltage VC do you expect at the collector? 10 V 5k VE VB 100 k VC 5k 10 V Ans. 0.994; 165; –1.75 V Figure E6.14 6.2.2 Graphical Representation of Transistor Characteristics It is sometimes useful to describe the transistor i–v characteristics graphically. Figure 6.16 shows the iC–vBE characteristic, which is the exponential relationship iC = I evBE /VT S which is identical to the diode i–v relationship. The iE –vBE and iB–vBE characteristics are also exponential but with different scale currents: IS/α for iE, and IS/β for iB. Since the constant of the exponential characteristic, 1/VT , is quite high ( 40), the curve rises very sharply. For vBE smaller than about 0.5 V, the current is negligibly small.7 Also, over most of the normal current range vBE lies in the range of 0.6 V to 0.8 V. In performing rapid first-order dc calculations, we normally will assume that VBE 0.7 V, which is similar to the approach used in the analysis of diode circuits (Chapter 4). For a pnp transistor, the iC –vEB characteristic will look identical to that of Fig. 6.16 with vBE replaced with vEB. 7The iC−vBE characteristic is the BJT’s counterpart of the iD–vGS characteristic of the MOSFET. They share an important attribute: In both cases the voltage has to exceed a “threshold” for the device to conduct appreciably. In the case of the MOSFET, there is a formal threshold voltage, Vt, which lies typically in the range of 0.4 V to 0.8 V. For the BJT, there is an “apparent threshold” of approximately 0.5 V. The iD–vGS characteristic of the MOSFET is parabolic, and thus is less steep than the iC–vBE characteristic of the BJT. As will be seen in Chapter 7, this difference has a direct and significant implication for the value of transconductance gm realized with each device. 326 Chapter 6 Bipolar Junction Transistors (BJTs) Figure 6.16 The iC–vBE characteristic for an npn transistor. Figure 6.17 Effect of temperature on the iC–vBE characteristic. At a constant emitter current (broken line), vBE changes by −2 mV/°C. As in silicon diodes, the voltage across the emitter–base junction decreases by about 2 mV for each rise of 1°C in temperature, provided the junction is operating at a constant current. Figure 6.17 illustrates this temperature dependence by depicting iC–vBE curves for an npn transistor at three different temperatures. EXERCISE 6.15 Consider a pnp transistor with vEB = 0.7 V at iE = 1 mA. Let the base be grounded, the emitter be fed by a 2-mA constant-current source, and the collector be connected to a –5-V supply through a 1-k resistance. If the temperature increases by 30°C, find the changes in emitter and collector voltages. Neglect the effect of ICBO. Ans. –60 mV; 0 V 6.2.3 Dependence of iC on the Collector Voltage—The Early Effect When operated in the active region, practical BJTs show some dependence of the collector current on the collector voltage, with the result that, unlike the graph shown in Fig. 6.8, their iC–vCB characteristics are not perfectly horizontal straight lines. To see this dependence more 6.2 Current–Voltage Characteristics 327 Figure 6.18 (a) Conceptual circuit for measuring the iC–vCE characteristics of the BJT. (b) The iC–vCE characteristics of a practical BJT. clearly, consider the conceptual circuit shown in Fig. 6.18(a). The transistor is connected in the common-emitter configuration; that is, here the emitter serves as a common terminal between the input and output ports. The voltage VBE can be set to any desired value by adjusting the dc source connected between base and emitter. At each value of VBE, the corresponding iC–vCE characteristic curve can be measured point by point by varying the dc source connected between collector and emitter and measuring the corresponding collector current. The result is the family of iC–vCE characteristic curves shown in Fig. 6.18(b) and known as common-emitter characteristics. At low values of vCE (lower than about 0.3 V), as the collector voltage goes below that of the base by more than 0.4 V, the collector–base junction becomes forward biased and the transistor leaves the active mode and enters the saturation mode. Shortly, we shall look at the details of the iC–vCE curves in the saturation region. At this time, however, we wish to examine the characteristic curves in the active region in detail. We observe that the characteristic curves, though still straight lines, have finite slope. In fact, when extrapolated, the characteristic lines meet at a point on the negative vCE axis, at vCE = –VA. The voltage VA, a positive number, is a parameter for the particular BJT, with typical values in the range of 10 V to 100 V. As noted earlier, it is called the Early voltage, after J. M. Early, the engineering scientist who first studied this phenomenon. At a given value of vBE, increasing vCE increases the reverse-bias voltage on the collector–base junction, and thus increases the width of the depletion region of this junction (refer to Fig. 6.4). This in turn results in a decrease in the effective base width W. Recalling that IS is inversely proportional to W (Eq. 6.13), we see that IS will increase and that iC increases proportionally. This is the Early effect. For obvious reasons, it is also known as the base-width modulation effect.8 8Recall that the MOSFET’s counterpart is the channel-length modulation effect. These two effects are remarkably similar and have been assigned the same name, Early effect. 328 Chapter 6 Bipolar Junction Transistors (BJTs) The linear dependence of iC on vCE can be explicitly accounted for by assuming that IS remains constant and including the factor (1 + vCE/VA)in the equation for iC as follows: iC = I evBE /VT S 1 + vCE VA (6.18) The nonzero slope of the iC–vCE straight lines indicates that the output resistance looking into the collector is not infinite. Rather, it is finite and defined by ro ≡ Using Eq. (6.18) we can show that −1 ∂ iC ∂ v CE vBE = constant (6.19) ro = VA + VCE IC (6.20) where IC and VCE are the coordinates of the point at which the BJT is operating on the particular iC–vCE curve (i.e., the curve obtained for vBE equal to constant value VBE at which Eq. (6.19) is evaluated). Alternatively, we can write ro = VA IC (6.21) where IC is the value of the collector current with the Early effect neglected; that is, IC = I eVBE /VT S (6.22) It is rarely necessary to include the dependence of iC on vCE in dc bias design and analysis that is performed by hand. Such an effect, however, can be easily included in the SPICE simulation of circuit operation, which is frequently used to “fine-tune” pencil-and-paper analysis or design. The finite output resistance ro can have a significant effect on the gain of transistor amplifiers. This is particularly the case in integrated-circuit amplifiers, as will be shown in Chapter 8. Fortunately, there are many situations in which ro can be included relatively easily in pencil-and-paper analysis. The output resistance ro can be included in the circuit model of the transistor.9 This is illustrated in Fig. 6.19, where we show the two large-signal circuit models of a Figure 6.19 Large-signal, equivalent-circuit models of an npn BJT operating in the active mode in the common-emitter configuration with the output resistance ro included. 9In applying Eq. (6.21) to determine ro we will usually drop the prime and simply use ro = VA/IC where IC is the collector current without the Early effect. 6.2 Current–Voltage Characteristics 329 common-emitter npn transistor operating in the active mode, those in Fig 6.5(c) and (d), with the resistance ro connected between the collector and the emitter terminals. EXERCISES 6.16 Use the circuit model in Fig. 6.19(a) to express iC in terms of evBE /VT and vCE and thus show that this circuit is a direct representation of Eq. (6.18). 6.17 Find the output resistance of a BJT for which VA = 100 V at IC = 0.1, 1, and 10 mA. Ans. 1 M ; 100 k ; 10 k 6.18 Consider the circuit in Fig. 6.18(a). At VCE = 1 V, VBE is adjusted to yield a collector current of 1 mA. Then, while VBE is kept constant, VCE is raised to 11 V. Find the new value of IC. For this transistor, VA = 100 V. Ans. 1.1 mA 6.2.4 An Alternative Form of the Common-Emitter Characteristics An alternative way of expressing the transistor common-emitter characteristics is illustrated in Fig. 6.20. Here the base current iB rather than the base–emitter voltage vBE is used as a parameter. That is, each iC–vCE curve is measured with the base fed with a constant current IB. The resulting characteristics, shown in Fig. 6.20(b), look similar to those in Fig. 6.18. Figure 6.20(c) shows an expanded view of the characteristics in the saturation region. The Common-Emitter Current Gain β In the active region of the characteristics shown in Fig. 6.20(b) we have identified a particular point Q. Note that this operating point for the transistor is characterized by a base current IB, a collector current IC, and a collector–emitter voltage VCE. The ratio IC/IB is the transistor β. However, there is another way to measure β: change the base current by an increment iB and measure the resulting increment iC, while keeping VCE constant. This is illustrated in Fig. 6.20(b). The ratio iC/ iB should, according to our study thus far, yield an identical value for β. It turns out, however, that the latter value of β (called incremental, or ac, β) is a little different from the dc β (i.e., IC/IB). Such a distinction, however, is too subtle for our needs in this book. We shall use β to denote both dc and incremental values.10 The Saturation Voltage VCEsat and Saturation Resistance RCEsat Refer next to the expanded view of the common-emitter characteristics in the saturation region shown in Fig. 6.20(c). The “bunching together” of the curves in the saturation region implies that the incremental β is lower there than in the active region. A possible operating point in the saturation region is that labeled X. It is characterized by a base current IB, a collector current ICsat, and a collector–emitter voltage VCEsat. From our previous discussion of saturation, recall that ICsat = βforced IB, where βforced < β. 10Manufacturers of bipolar transistors use hFE to denote the dc value of β and hfe to denote the incremental β. These symbols come from the h-parameter description of two-port networks (see Appendix C), with the subscript F(f) denoting forward and E(e) denoting common emitter. 330 Chapter 6 Bipolar Junction Transistors (BJTs) IB iB iC vCE iC Saturation region Active region iB = . . . iB = . . . ΔiC iB = IB + ΔiB IC Q iB = IB iB = . . . 0 VCE iB = 0 vCE (a) (b) iB IB bIB Slope 1 RCEsat ICsat X VCEsat (c) Figure 6.20 Common-emitter characteristics. (a) Basic CE circuit; note that in (b) the horizontal scale is expanded around the origin to show the saturation region in some detail. A much greater expansion of the saturation region is shown in (c) . The iC–vCE curves in saturation are rather steep, indicating that the saturated transistor exhibits a low collector-to-emitter resistance RCEsat, RCEsat ≡ ∂ vCE ∂ iC iB = IB iC = ICsat (6.23) Typically, RCEsat ranges from a few ohms to a few tens of ohms. IB B VBE ICsat 0.7 V 0.2 V VCEsat 6.2 Current–Voltage Characteristics 331 E Figure 6.21 A simplified equivalent-circuit model of the saturated transistor. That the collector-to-emitter resistance of a saturated BJT is small should have been anticipated from the fact that between C and E we now have two forward-conducting diodes in series11 (see also Fig. 6.9). A simple model for the saturated BJT is shown in Fig. 6.21. Here VBE is assumed constant (approximately 0.7 V) and VCE also is assumed constant, VCEsat 0.2 V. That is, we have neglected the small saturation resistance RCEsat for the sake of making the model simple for hand calculations. Example 6.3 For the circuit in Fig. 6.22, it is required to determine the value of the voltage VBB that results in the transistor operating (a) in the active mode with VCE = 5 V (b) at the edge of saturation (c) deep in saturation with βforced = 10 For simplicity, assume that VBE remains constant at 0.7 V. The transistor β is specified to be 50. VCC 10V VBB RB 10 k IC IB RC 1 k VCE VBE Figure 6.22 Circuit for Example 6.3. 11In the corresponding mode of operation for the MOSFET, the triode region, the resistance between drain and source is small because it is the resistance of the continuous (non-pinched-off) channel. 332 Chapter 6 Bipolar Junction Transistors (BJTs) Example 6.3 continued Solution (a) To operate in the active mode with VCE = 5 V, IC = VCC − VCE RC = 10 − 5 = 5 mA 1k IB = IC β = 5 50 = 0.1 mA Now the required value of VBB can be found as follows: VBB = IBRB + VBE = 0.1 × 10 + 0.7 = 1.7 V (b) Operation at the edge of saturation is obtained with VCE = 0.3 V. Thus 10 − 0.3 IC = 1 = 9.7 mA Since, at the edge of saturation, IC and IB are still related by β, 9.7 IB = 50 = 0.194 mA The required value of VBB can be determined as VBB = 0.194 × 10 + 0.7 = 2.64 V (c) To operate deep in saturation, VCE = VCEsat 0.2 V Thus, IC = 10 − 0.2 1 = 9.8 mA We then use the value of forced β to determine the required value of IB as IB = IC β forced = 9.8 10 = 0.98 mA and the required VBB can now be found as VBB = 0.98 × 10 + 0.7 = 10.5 V Observe that once the transistor is in saturation, increasing VBB and thus IB results in negligible change in IC since VCEsat will change only slightly. Thus IC is said to saturate, which is the origin of the name “saturation mode of operation.” 6.3 BJT Circuits at DC 333 EXERCISES 6.19 Repeat Example 6.3 for RC = 10 k . Ans. 0.8 V; 0.894 V; 1.68 V 6.20 For the circuit in Fig. 6.22, find VCE for VBB = 0 V. Ans. + 10 V 6.21 For the circuit in Fig. 6.22, let VBB be set to the value obtained in Example 6.3, part (a), namely, VBB = 1.7 V. Verify that the transistor is indeed operating in the active mode. Now, while keeping VBB constant, find the value to which RC should be increased in order to obtain (a) operation at the edge of saturation and (b) operation deep in saturation with βforced = 10. Ans. (a) 1.94 k ; (b) 9.8 k 6.3 BJT Circuits at DC We are now ready to consider the analysis of BJT circuits to which only dc voltages are applied. In the following examples we will use the simple model in which |VBE| of a conducting transistor is 0.7 V and |VCE| of a saturated transistor is 0.2 V, and we will neglect the Early effect. These models are shown in Table 6.3. Better models can, of course, be used to obtain more accurate results. This, however, is usually achieved at the expense of speed of analysis; more importantly, the attendant complexity could impede the circuit designer’s ability to gain insight regarding circuit behavior. Accurate results using elaborate models can be obtained using circuit simulation with SPICE. This is almost always done in the final stages of a design and certainly before circuit fabrication. Computer simulation, however, is not a substitute for quick pencil-and-paper circuit analysis, an essential ability that aspiring circuit designers must master. The following series of examples is a step in that direction. As will be seen, in analyzing a circuit the first question that one must answer is: In which mode is the transistor operating? In some cases, the answer will be obvious. For instance, a quick check of the terminal voltages will indicate whether the transistor is cut off or conducting. If it is conducting, we have to determine whether it is operating in the active mode or in saturation. In some cases, however, this may not be obvious. Needless to say, as the reader gains practice and experience in transistor circuit analysis and design, the answer will be apparent in a much larger proportion of problems. The answer, however, can always be determined by utilizing the following procedure. Assume that the transistor is operating in the active mode and, using the active-mode model in Table 6.3, proceed to determine the various voltages and currents that correspond. Then check for consistency of the results with the assumption of active-mode operation; that is, is VCB of an npn transistor greater than −0.4 V (or VCB of a pnp transistor lower than 0.4 V)? If the answer is yes, then our task is complete. If the answer is no, assume saturation-mode operation and, using the saturation-mode model in Table 6.3, proceed to determine currents and voltages 334 Chapter 6 Bipolar Junction Transistors (BJTs) Table 6.3 Simplified Models for the Operation of the BJT in DC Circuits npn Active EBJ: Forward Biased CBJ: Reverse Biased IB B VBE 0.7 V bIB E C VCE > 0.3 V VEB 0.7 V B IB Saturation EBJ: Forward IB B Biased CBJ: VBE 0.7 V Forward Biased IC bforced IB C VCEsat 0.2 V VEB 0.7 V B E IB pnp E bIB VEC > 0.3 V C E VECsat 0.2 V C IC = bforcedIB and then check for consistency of the results with the assumption of saturation-mode operation. Here the test is usually to compute the ratio IC/IB and to verify that it is lower than the transistor β (i.e., βforced < β). Since β for a given transistor type varies over a wide range,12 one must use the lowest specified β for this test. Finally, note that the order of these two assumptions can be reversed. A Note on Units Except when otherwise specified, throughout this book we use a consistent set of units, namely, volts (V), milliamps (mA), and kilohms (k ). 12That is, if one buys BJTs of a certain part number, the manufacturer guarantees only that their values of β fall within a certain range, say 50 to 150. 6.3 BJT Circuits at DC 335 Example 6.4 Consider the circuit shown in Fig. 6.23(a), which is redrawn in Fig. 6.23(b) to remind the reader of the convention employed throughout this book for indicating connections to dc sources. We wish to analyze this circuit to determine all node voltages and branch currents. We will assume that β is specified to be 100. 10 V IC 4V IB RE 3.3 k RC 4.7 k VC VE IE RC 4.7 k 4V RE 3.3 k 10 V (a) (b) 10 V 3 0.99 1 0.99 mA 4V 4.7 k 10 0.99 4.7 5.3 V 4 5 1.00 0.99 0.01 mA 3.3 k 4 0.7 3.3 V 1 3.3 1 mA 2 3.3 (c) Figure 6.23 Analysis of the circuit for Example 6.4: (a) circuit; (b) circuit redrawn to remind the reader of the convention used in this book to show connections to the dc sources; (c) analysis with the steps numbered. 336 Chapter 6 Bipolar Junction Transistors (BJTs) Example 6.4 continued Solution Glancing at the circuit in Fig. 6.23(a), we note that the base is connected to +4 V and the emitter is connected to ground through a resistance RE. Therefore, it is reasonable to conclude that the base–emitter junction will be forward biased. Assuming that this is the case and assuming that VBE is approximately 0.7 V, it follows that the emitter voltage will be VE = 4 − VBE 4 − 0.7 = 3.3 V We are now in an opportune position; we know the voltages at the two ends of RE and thus can determine the current IE through it, IE = VE − RE 0 = 3.3 3.3 = 1 mA Since the collector is connected through RC to the +10-V power supply, it appears possible that the collector voltage will be higher than the base voltage, which implies active-mode operation. Assuming that this is the case, we can evaluate the collector current from The value of α is obtained from Thus IC will be given by IC = αIE α = β β +1 = 100 101 0.99 IC = 0.99 × 1 = 0.99 mA We are now in a position to use Ohm’s law to determine the collector voltage VC, VC = 10 − ICRC = 10 − 0.99 × 4.7 +5.3 V Since the base is at +4 V, the collector–base junction is reverse biased by 1.3 V, and the transistor is indeed in the active mode as assumed. It remains only to determine the base current IB, as follows: IB = β IE +1 = 1 101 0.01 mA Before leaving this example, we wish to emphasize strongly the value of carrying out the analysis directly on the circuit diagram. Only in this way will one be able to analyze complex circuits in a reasonable length of time. Figure 6.23(c) illustrates the above analysis on the circuit diagram, with the order of the analysis steps indicated by the circled numbers. 6.3 BJT Circuits at DC 337 Example 6.5 We wish to analyze the circuit of Fig. 6.24(a) to determine the voltages at all nodes and the currents through all branches. Note that this circuit is identical to that of Fig. 6.23 except that the voltage at the base is now +6 V. Assume that the transistor β is specified to be at least 50. 10 V 4.7 k 6V 3.3 k (a) 10 V 3 1.6 mA 4.7 k 6V 10 1.6 4.7 2.48 4 Impossible, not in active mode 3.3 k 6 0.7 5.3 V 1 5.3 1.6 mA 2 3.3 (b) (c) Figure 6.24 Analysis of the circuit for Example 6.5. Note that the circled numbers indicate the order of the analysis steps. 338 Chapter 6 Bipolar Junction Transistors (BJTs) Example 6.5 continued Solution With +6 V at the base, the base–emitter junction will be forward biased; thus, VE = +6 − VBE 6 − 0.7 = 5.3 V and 5.3 IE = 3.3 = 1.6 mA Now, assuming active-mode operation, IC = αIE IE; thus, VC = +10 − 4.7 × IC 10 − 7.52 = 2.48 V The details of the analysis performed above are illustrated in Fig. 6.24(b). Since the collector voltage calculated is less than the base voltage by 3.52 V, it follows that our original assumption of active-mode operation is incorrect. In fact, the transistor has to be in the saturation mode. Assuming this to be the case, the values of VE and IE will remain unchanged. The collector voltage, however, becomes VC = VE + VCEsat 5.3 + 0.2 = +5.5 V from which we can determine IC as 10 − 5.5 IC = 4.7 = 0.96 mA and IB can now be found as IB = IE − IC = 1.6 − 0.96 = 0.64 mA Thus the transistor is operating at a forced β of β forced = IC IB = 0.96 0.64 = 1.5 Since βforced is less than the minimum specified value of β, the transistor is indeed saturated. We should emphasize here that in testing for saturation the minimum value of β should be used. By the same token, if we are designing a circuit in which a transistor is to be saturated, the design should be based on the minimum specified β. Obviously, if a transistor with this minimum β is saturated, then transistors with higher values of β will also be saturated. The details of the analysis are shown in Fig. 6.24(c), where the order of the steps used is indicated by the circled numbers. 6.3 BJT Circuits at DC 339 Example 6.6 We wish to analyze the circuit in Fig. 6.25(a) to determine the voltages at all nodes and the currents through all branches. Note that this circuit is identical to that considered in Examples 6.4 and 6.5 except that now the base voltage is zero. 2 1 (a) (b) Figure 6.25 Example 6.6: (a) circuit; (b) analysis, with the order of the analysis steps indicated by circled numbers. Solution Since the base is at zero volts and the emitter is connected to ground through RE, the base–emitter junction cannot conduct and the emitter current is zero. Also, the collector–base junction cannot conduct, since the n-type collector is connected through RC to the positive power supply while the p-type base is at ground. It follows that the collector current will be zero. The base current will also have to be zero, and the transistor is in the cutoff mode of operation. The emitter voltage will be zero, while the collector voltage will be equal to +10 V, since the voltage drops across RE and RC are zero. Figure 6.25(b) shows the analysis details. EXERCISES D6.22 For the circuit in Fig. 6.23(a), find the highest voltage to which the base can be raised while the transistor remains in the active mode. Assume α 1. Ans. +4.7 V D6.23 Redesign the circuit of Fig. 6.23(a) (i.e., find new values for RE and RC) to establish a collector current of 0.5 mA and a reverse-bias voltage on the collector–base junction of 2 V. Assume α 1. Ans. RE = 6.6 k ; RC = 8 k 340 Chapter 6 Bipolar Junction Transistors (BJTs) D6.24 For the circuit in Fig. 6.24(a), find the value to which the base voltage should be changed so that the transistor operates in saturation with a forced β of 5. Ans. +5.18 V Example 6.7 We want to analyze the circuit of Fig. 6.26(a) to determine the voltages at all nodes and the currents through all branches. V 10 V RE 2 k 10 V 10 0.7 2 2k 4.65 mA 2 5 0.05 mA 0.7 V 1 RC 1 k 3 0.99 4.65 4.6 mA 10 4.6 1 5.4 V 4 1k V 10 V (a) 10 V (b) Figure 6.26 Example 6.7: (a) circuit; (b) analysis, with the steps indicated by circled numbers. Solution The base of this pnp transistor is grounded, while the emitter is connected to a positive supply (V + = +10 V) through RE. It follows that the emitter–base junction will be forward biased with VE = VEB 0.7 V Thus the emitter current will be given by IE = V + − VE RE = 10 − 0.7 2 = 4.65 mA 6.3 BJT Circuits at DC 341 Since the collector is connected to a negative supply (more negative than the base voltage) through RC, it is possible that this transistor is operating in the active mode. Assuming this to be the case, we obtain IC = αIE Since no value for β has been given, we shall assume β = 100, which results in α = 0.99. Since large variations in β result in small differences in α, this assumption will not be critical as far as determining the value of IC is concerned. Thus, IC = 0.99 × 4.65 = 4.6 mA The collector voltage will be VC = V − + ICRC = −10 + 4.6 × 1 = −5.4 V Thus the collector–base junction is reverse biased by 5.4 V, and the transistor is indeed in the active mode, which supports our original assumption. It remains only to calculate the base current, IB = β IE +1 = 4.65 101 0.05 mA Obviously, the value of β critically affects the base current. Note, however, that in this circuit the value of β will have no effect on the mode of operation of the transistor. Since β is generally an ill-specified parameter, this circuit represents a good design. As a rule, one should strive to design the circuit such that its performance is as insensitive to the value of β as possible. The analysis details are illustrated in Fig. 6.26(b). EXERCISES D6.25 For the circuit in Fig. 6.26(a), find the largest value to which RC can be raised while the transistor remains in the active mode. Ans. 2.26 k D6.26 Redesign the circuit of Fig. 6.26(a) (i.e., find new values for RE and RC) to establish a collector current of 1 mA and a reverse bias on the collector–base junction of 4 V. Assume α 1. Ans. RE = 9.3 k ; RC = 6 k 342 Chapter 6 Bipolar Junction Transistors (BJTs) Example 6.8 We want to analyze the circuit in Fig. 6.27(a) to determine the voltages at all nodes and the currents in all branches. Assume β = 100. (a) (b) Figure 6.27 Example 6.8: (a) circuit; (b) analysis, with the steps indicated by the circled numbers. Solution The base–emitter junction is clearly forward biased. Thus, IB = +5 − VBE RB 5 − 0.7 = 0.043 mA 100 Assume that the transistor is operating in the active mode. We now can write IC = βIB = 100 × 0.043 = 4.3 mA The collector voltage can now be determined as VC = 10 − ICRC = 10 − 4.3 × 2 = +1.4 V Since the base voltage VB is VB = VBE + 0.7 V it follows that the collector–base junction is reverse biased by 0.7 V and the transistor is indeed in the active mode. The emitter current will be given by IE = (β + 1)IB = 101 × 0.043 4.3 mA We note from this example that the collector and emitter currents depend critically on the value of β. In fact, if β were 10% higher, the transistor would leave the active mode and enter saturation. Therefore this clearly is a bad design. The analysis details are illustrated in Fig. 6.27(b). 6.3 BJT Circuits at DC 343 EXERCISE D6.27 The circuit of Fig. 6.27(a) is to be fabricated using a transistor type whose β is specified to be in the range of 50 to 150. That is, individual units of this same transistor type can have β values anywhere in this range. Redesign the circuit by selecting a new value for RC so that all fabricated circuits are guaranteed to be in the active mode. What is the range of collector voltages that the fabricated circuits may exhibit? Ans. RC = 1.5 k ; VC = 0.3 V to 6.8 V Example 6.9 We want to analyze the circuit of Fig. 6.28(a) to determine the voltages at all nodes and the currents through all branches. The minimum value of β is specified to be 30. 5V 1k 10 k 10 k 5V 4 IE 5 ( VB 1 0.7) 2 IB VB /10 10 k 1 VB 7 IC VB 0.5 ( 5) 10 1k VE VB 0.7 3 VEC sat 0.2 V 5 VC VB 0.5 6 10 k 5V (a) 5V (b) Figure 6.28 Example 6.9: (a) circuit; (b) analysis with steps numbered. Solution A quick glance at this circuit reveals that the transistor will be either active or saturated. Assuming active-mode operation and neglecting the base current, we see that the base voltage will be approximately zero volts, the emitter voltage will be approximately +0.7 V, and the emitter current will be approximately 4.3 mA. Since the maximum current that the collector can support while the transistor remains in the active mode is approximately 0.5 mA, it follows that the transistor is definitely saturated. 344 Chapter 6 Bipolar Junction Transistors (BJTs) Example 6.9 continued Assuming that the transistor is saturated and denoting the voltage at the base by VB (refer to Fig. 6.28b), it follows that VE = VB + VEB VB + 0.7 VC = VE − VECsat VB + 0.7 − 0.2 = VB + 0.5 IE = +5 − VE 1 = 5 − VB − 0.7 1 = 4.3 − VB mA IB = VB 10 = 0.1VB mA IC = VC − (−5) 10 = VB + 0.5 + 5 10 = 0.1VB + 0.55 mA Using the relationship IE = IB + IC, we obtain 4.3 − VB = 0.1VB + 0.1VB + 0.55 which results in 3.75 VB = 1.2 Substituting in the equations above, we obtain 3.13 V VE = 3.83 V VC = 3.63 V IE = 1.17 mA IC = 0.86 mA IB = 0.31 mA from which we see that the transistor is saturated, since the value of forced β is β forced = 0.86 0.31 2.8 which is much smaller than the specified minimum β. Example 6.10 We want to analyze the circuit of Fig. 6.29(a) to determine the voltages at all nodes and the currents through all branches. Assume β = 100. 6.3 BJT Circuits at DC 345 15 V RB 1 RC 5k 100 k RB 2 50 k RE 3k 15 V RC VBB 5V 5k RBB 33.3 k IE IB L RE 3k (a) (b) 15 V 1.28 mA 5V 33.3 k 5k 8.6 V 0.013 mA 4.57 V 1.29 mA 3.87 V 3k 15 V 100 k 50 k 0.103 mA 0.013 mA 4.57 V 0.09 mA (c) (d) Figure 6.29 Circuits for Example 6.10. Solution The first step in the analysis consists of simplifying the base circuit using The´venin’s theorem. The result is shown in Fig. 6.29(b), where VBB = +15 RB2 RB1 + RB2 = 50 15 100 + 50 = +5 V RBB = RB1 RB2 = 100 50 = 33.3 k 346 Chapter 6 Bipolar Junction Transistors (BJTs) Example 6.10 continued To evaluate the base or the emitter current, we have to write a loop equation around the loop labeled L in Fig. 6.29(b). Note, however, that the current through RBB is different from the current through RE. The loop equation will be VBB = IBRBB + VBE + IE RE Now, assuming active-mode operation, we replace IB with IB = β IE + 1 and rearrange the equation to obtain IE = RE VBB − VBE + [RBB/(β + 1)] For the numerical values given we have 5 − 0.7 IE = 3 + (33.3/101) = 1.29 mA The base current will be The base voltage is given by 1.29 IB = 101 = 0.0128 mA VB = VBE + IE RE = 0.7 + 1.29 × 3 = 4.57 V We can evaluate the collector current as IC = αIE = 0.99 × 1.29 = 1.28 mA The collector voltage can now be evaluated as VC = +15 − ICRC = 15 − 1.28 × 5 = 8.6 V It follows that the collector is higher in potential than the base by 4.03 V, which means that the transistor is in the active mode, as had been assumed. The results of the analysis are given in Fig. 6.29(c, d). EXERCISE 6.28 If the transistor in the circuit of Fig. 6.29(a) is replaced with another having half the value of β (i.e., β = 50), find the new value of IC, and express the change in IC as a percentage. Ans. IC = 1.15 mA; –10% 6.3 BJT Circuits at DC 347 Example 6.11 We wish to analyze the circuit in Fig. 6.30(a) to determine the voltages at all nodes and the currents through all branches. 15 V RB1 100 k Q1 RB2 50 k RC1 5 k Q2 IC1 IB2 RE 3 k RE2 2 k IE2 IC2 RC2 2.7 k (a) 15 V 0.103 mA 1.252 mA 100 k 2.78 mA 5k 4.57 V Q2 8.74 V Q1 0.0275 mA 1.28 mA 0.013 mA 3.87 V 50 k 0.09 mA 3k 1.29 mA 2k 9.44 V 2.7 k 7.43 V 2.75 mA (b) Figure 6.30 Circuits for Example 6.11. 348 Chapter 6 Bipolar Junction Transistors (BJTs) Example 6.11 continued Solution We first recognize that part of this circuit is identical to the circuit we analyzed in Example 6.10 —namely, the circuit of Fig. 6.29(a). The difference, of course, is that in the new circuit we have an additional transistor Q2 together with its associated resistors RE2 and RC2. Assume that Q1 is still in the active mode. The following values will be identical to those obtained in the previous example: VB1 = +4.57 V IB1 = 0.0128 mA IE1 = 1.29 mA IC1 = 1.28 mA However, the collector voltage will be different than previously calculated, since part of the collector current IC1 will flow in the base lead of Q2 (IB2). As a first approximation we may assume that IB2 is much smaller than IC1; that is, we may assume that the current through RC1 is almost equal to IC1. This will enable us to calculate VC1: VC1 + 15 − IC1RC1 = 15 − 1.28 × 5 = +8.6 V Thus Q1 is in the active mode, as had been assumed. As far as Q2 is concerned, we note that its emitter is connected to +15 V through RE2. It is therefore safe to assume that the emitter–base junction of Q2 will be forward biased. Thus the emitter of Q2 will be at a voltage VE2 given by VE2 = VC1 + VEB Q2 8.6 + 0.7 = +9.3 V The emitter current of Q2 may now be calculated as IE2 = +15 − VE2 RE2 = 15 − 9.3 2 = 2.85 mA Since the collector of Q2 is returned to ground via RC2, it is possible that Q2 is operating in the active mode. Assume this to be the case. We now find IC2 as IC2 = α2IE2 = 0.99 × 2.85 = 2.82 mA assuming β2 = 100 The collector voltage of Q2 will be VC2 = IC2RC2 = 2.82 × 2.7 = 7.62 V which is lower than VB2 by 0.98 V. Thus Q2 is in the active mode, as assumed. 6.3 BJT Circuits at DC 349 It is important at this stage to find the magnitude of the error incurred in our calculations by the assumption that IB2 is negligible. The value of IB2 is given by IB2 = IE2 β2 + 1 = 2.85 101 = 0.028 mA which is indeed much smaller than IC1 (1.28 mA). If desired, we can obtain more accurate results by iterating one more time, assuming IB2 to be 0.028 mA. The new values will be Current in RC1 = IC1 − IB2 = 1.28 − 0.028 = 1.252 mA VC1 = 15 − 5 × 1.252 = 8.74 V VE2 = 8.74 + 0.7 = 9.44 V IE2 = 15 − 9.44 2 = 2.78 mA IC2 = 0.99 × 2.78 = 2.75 mA VC2 = 2.75 × 2.7 = 7.43 V 2.78 IB2 = 101 = 0.0275 mA Note that the new value of IB2 is very close to the value used in our iteration, and no further iterations are warranted. The final results are indicated in Fig. 6.30(b). The reader justifiably might be wondering about the necessity for using an iterative scheme in solving a linear (or linearized) problem. Indeed, we can obtain the exact solution (if we can call anything we are doing with a first-order model exact!) by writing appropriate equations. The reader is encouraged to find this solution and then compare the results with those obtained above. It is important to emphasize, however, that in most such problems it is quite sufficient to obtain an approximate solution, provided we can obtain it quickly and, of course, correctly. In the above examples, we frequently used a precise value of α to calculate the collector current. Since α 1, the error in such calculations will be very small if one assumes α = 1 and IC = IE. Therefore, except in calculations that depend critically on the value of α (e.g., the calculation of base current), one usually assumes α 1. EXERCISES 6.29 For the circuit in Fig. 6.30, find the total current drawn from the power supply. Hence find the power dissipated in the circuit. Ans. 4.135 mA; 62 mW 6.30 The circuit in Fig. E6.30 is to be connected to the circuit in Fig. 6.30(a) as indicated; specifically, the base of Q3 is to be connected to the collector of Q2. If Q3 has β = 100, find the new value of VC2 and the values of VE3 and IC3. 350 Chapter 6 Bipolar Junction Transistors (BJTs) To collector of Q2 in Fig. 6.30 (a) Ans. +7.06 V; +6.36 V; 13.4 mA Figure E6.30 Example 6.12 We desire to evaluate the voltages at all nodes and the currents through all branches in the circuit of Fig. 6.31(a). Assume β = 100. On 5 – 0.7 = 10 ϩ 101 ϫ 1 0.039 mA 0 0 3.9 mA Off –5V (a) (b) Figure 6.31 Example 6.12: (a) circuit; (b) analysis with the steps numbered. 6.4 Transistor Breakdown and Temperature Effects 351 Solution By examining the circuit, we conclude that the two transistors Q1 and Q2 cannot be simultaneously conducting. Thus if Q1 is on, Q2 will be off, and vice versa. Assume that Q2 is on. It follows that current will flow from ground through the 1-k resistor into the emitter of Q2. Thus the base of Q2 will be at a negative voltage, and base current will be flowing out of the base through the 10-k resistor and into the +5-V supply. This is impossible, since if the base is negative, current in the 10-k resistor will have to flow into the base. Thus we conclude that our original assumption—that Q2 is on—is incorrect. It follows that Q2 will be off and Q1 will be on. The question now is whether Q1 is active or saturated. The answer in this case is obvious: Since the base is fed with a +5-V supply and since base current flows into the base of Q1, it follows that the base of Q1 will be at a voltage lower than +5 V. Thus the collector–base junction of Q1 is reverse biased and Q1 is in the active mode. It remains only to determine the currents and voltages using techniques already described in detail. The results are given in Fig. 6.31(b). EXERCISES 6.31 Solve the problem in Example 6.12 for the case of a voltage of –5 V feeding the bases. What voltage appears at the emitters? Ans. –3.9 V 6.32 Solve the problem in Example 6.12 with the voltage feeding the bases changed to +10 V. Assume that βmin = 30, and find VE, VB, IC1, and IC2. Ans. +4.8 V; +5.5 V; 4.35 mA; 0 6.4 Transistor Breakdown and Temperature Effects We conclude this chapter with a brief discussion of two important nonideal effects in the BJT: voltage breakdown, and the dependence of β on IC and temperature. 6.4.1 Transistor Breakdown The maximum voltages that can be applied to a BJT are limited by the EBJ and CBJ breakdown effects that follow the avalanche multiplication mechanism described in Section 3.5.3. Consider first the common-base configuration (Fig. 6.32(a)). The iC– vCB characteristics in Fig. 6.32(b) indicate that for iE = 0 (i.e., with the emitter open-circuited) the collector–base junction breaks down at a voltage denoted by BVCBO. For iE > 0, breakdown occurs at voltages smaller than BVCBO. Typically, for discrete BJTs, BVCBO is greater than 50 V. 352 Chapter 6 Bipolar Junction Transistors (BJTs) Saturation region iC Active region aIE1 aIE2 iC vCB iE IE1 iE IE2 0 iE 0.4 V Expanded scale (a) (b) iE 0 vCB BVCBO Figure 6.32 The BJT common-base characteristics including the transistor breakdown region. Figure 6.33 The BJT common-emitter characteristics including the breakdown region. Next consider the common-emitter characteristics of Fig. 6.33, which show breakdown occurring at a voltage BVCEO. Here, although breakdown is still of the avalanche type, the effects on the characteristics are more complex than in the common-base configuration. We will not explain these in detail; it is sufficient to point out that typically BVCEO is about half BVCBO. On transistor data sheets, BVCEO is sometimes referred to as the sustaining voltage LVCEO. 6.4 Transistor Breakdown and Temperature Effects 353 Breakdown of the CBJ in either the common-base or common-emitter configuration is not destructive as long as the power dissipation in the device is kept within safe limits. This, however, is not the case with the breakdown of the emitter–base junction. The EBJ breaks down in an avalanche manner at a voltage BVEBO much smaller than BVCBO. Typically, BVEBO is in the range of 6 V to 8 V, and the breakdown is destructive in the sense that the β of the transistor is permanently reduced. This does not prevent use of the EBJ as a zener diode to generate reference voltages in IC design. In such applications one is not concerned with the β-degradation effect. A circuit arrangement to prevent EBJ breakdown in IC amplifiers will be discussed in Chapter 13. Transistor breakdown and the maximum allowable power dissipation are important parameters in the design of power amplifiers (Chapter 12). EXERCISE 6.33 What is the output voltage of the circuit in Fig. E6.33 if the transistor BVBCO = 70 V? μA Ans. –60 V Figure E6.33 6.4.2 Dependence of β on IC and Temperature Throughout this chapter we have assumed that the transistor common-emitter dc current gain, β or hFE, is constant for a given transistor. In fact, β depends on the dc current at which the transistor is operating, as shown in Fig. 6.34. The physical processes that give rise to this dependence are beyond the scope of this book. Note, however, that there is a current range over which β is highest. Normally, one arranges to operate the transistor at a current within this range. Figure 6.34 also shows the dependence of β on temperature. The fact that β increases with temperature can lead to serious problems in transistors that operate at large power levels (see Chapter 12). 354 Chapter 6 Bipolar Junction Transistors (BJTs) μ Figure 6.34 Typical dependence of β on IC and on temperature in an integrated-circuit npn silicon transistor intended for operation around 1 mA. Summary Depending on the bias conditions on its two junctions, the BJT can operate in one of three possible modes: cutoff (both junctions reverse biased), active (the EBJ forward biased and the CBJ reverse biased), and saturation (both junctions forward biased). Refer to Table 6.1. For amplifier applications, the BJT is operated in the active mode. Switching applications make use of the cutoff and saturation modes. A BJT operating in the active mode provides a collector current iC = I e|vBE |/VT S . The base current iB = iC /β , and the emitter current iE = iC + iB. Also, iC = αiE, and thus β = α/(1 − α) and α = β/(β + 1). See Table 6.2. To ensure operation in the active mode, the collector voltage of an npn transistor must be kept higher than approximately 0.4 V below the base voltage. For a pnp transistor, the collector voltage must be lower than approximately 0.4 V above the base voltage. Otherwise, the CBJ becomes forward biased, and the transistor enters the saturation region. At a constant collector current, the magnitude of the base–emitter voltage decreases by about 2 mV for every 1°C rise in temperature. The BJT will be at the edge of saturation when vCE is reduced to about 0.3 V. In saturation, vCE 0.2 V, and the ratio of iC to iB is lower than β (i.e., βforced < β). In the active mode, iC shows a slight dependence on vCE. This phenomenon, known as the Early effect, is modeled by ascribing a finite (i.e., noninfinite) output resistance to the BJT: ro = VA /IC, where VA is the Early voltage and IC is the dc collector current without the Early effect taken into account. In discrete circuits, ro plays a minor role and can usually be neglected. This is not the case, however, in integrated-circuit design (Chapter 8). The dc analysis of transistor circuits is greatly simplified by assuming that VBE 0.7 V. Refer to Table 6.3. If the BJT is conducting, one assumes it is operating in the active mode and, using the active-mode model, proceeds to determine all currents and voltages. The validity of the initial assumption is then checked by determining whether the CBJ is reverse biased. If it is, the analysis is complete; otherwise, we assume the BJT is operating in saturation and redo the analysis, using the saturation-mode model and checking at the end that IC < βIB. PROBLEMS Computer Simulations Problems Problems identified by the Multisim/PSpice icon are intended to demonstrate the value of using SPICE simulation to verify hand analysis and design, and to investigate important issues such as allowable signal swing and amplifier nonlinear distortion. Instructions to assist in setting up PSPice and Multisim simulations for all the indicated problems can be found in the corresponding files on the website. Note that if a particular parameter value is not specified in the problem statement, you are to make a reasonable assumption. Section 6.1: Device Structure and Physical Operation 6.1 The terminal voltages of various npn transistors are measured during operation in their respective circuits with the following results: Case 1 2 3 4 5 6 E 0 0 −0.7 −0.7 1.3 0 B C 0.7 0.7 0.8 0.1 0 1.0 0 −0.6 2.0 5.0 0 5.0 Mode In this table, where the entries are in volts, 0 indicates the reference terminal to which the black (negative) probe of the voltmeter is connected. For each case, identify the mode of operation of the transistor. 6.2 Two transistors, fabricated with the same technology but having different junction areas, when operated at a base-emitter voltage of 0.75 V, have collector currents of 0.5 mA and 2 mA. Find IS for each device. What are the relative junction areas? transistors are operated in the active mode and conduct equal collector currents, what do you expect the difference in their vBE values to be? 6.5 Find the collector currents that you would expect for operation at vBE = 700 mV for transistors for which IS = 10−13 A and IS = 10−18 A. For the transistor with the larger EBJ, what is the vBE required to provide a collector current equal to that provided by the smaller transistor at vBE = 700 mV? Assume active-mode operation in all cases. 6.6 In this problem, we contrast two BJT integrated-circuit fabrication technologies: For the “old” technology, a typical npn transistor has IS = 2 × 10−15 A, and for the “new” technology, a typical npn transistor has IS = 2 × 10−18 A. These typical devices have vastly different junction areas and base width. For our purpose here we wish to determine the vBE required to establish a collector current of 1 mA in each of the two typical devices. Assume active-mode operation. 6.7 Consider an npn transistor whose base–emitter drop is 0.76 V at a collector current of 5 mA. What current will it conduct at vBE = 0.70 V? What is its base–emitter voltage for iC = 5 μA? 6.8 In a particular BJT, the base current is 10 μA, and the collector current is 800 μA. Find β and α for this device. 6.9 Find the values of β that correspond to α values of 0.5, 0.8, 0.9, 0.95, 0.98, 0.99, 0.995, and 0.999. 6.10 Find the values of α that correspond to β values of 1, 2, 10, 20, 50, 100, 200, 500, and 1000. *6.11 Show that for a transistor with α close to unity, if α changes by a small per-unit amount ( α/α), the corresponding per-unit change in β is given approximately by β β β α α 6.3 In a particular technology, a small BJT operating at vBE = 30VT conducts a collector current of 200 μA. What is the corresponding saturation current? For a transistor in the same technology but with an emitter junction that is 32 times larger, what is the saturation current? What current will this transistor conduct at vBE = 30VT ? What is the base–emitter voltage of the latter transistor at iC = 1 mA? Assume active-mode operation in all cases. 6.4 Two transistors have EBJ areas as follows: AE1 = 200 μm × 200 μm and AE2 = 0.4 μm × 0.4 μm. If the two Now, for a transistor whose nominal β is 100, find the percentage change in its α value corresponding to a drop in its β of 10%. 6.12 An npn transistor of a type whose β is specified to range from 50 to 300 is connected in a circuit with emitter grounded, collector at + 10 V, and a current of 10 μA injected into the base. Calculate the range of collector and emitter currents that can result. What is the maximum power dissipated in the transistor? (Note: Perhaps you can see why this is a bad way to establish the operating current in the collector of a BJT.) = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem 356 Chapter 6 Bipolar Junction Transistors (BJTs) CHAPTER 6 PROBLEMS 6.13 A BJT is specified to have IS = 5 × 10−15 A and β that falls in the range of 50 to 200. If the transistor is operated in the active mode with vBE set to 0.700 V, find the expected range of iC, iB, and iE. 6.14 Measurements made on a number of transistors operating in the active mode with iE = 1 mA indicate base currents of 10 μA, 20 μA, and 50 μA. For each device, find iC, β, and α. 6.15 Measurements of VBE and two terminal currents taken on a number of npn transistors operating in the active mode are tabulated below. For each, calculate the missing current value as well as α, β, and IS as indicated by the table. Transistor a b c d e VBE (mV) IC (mA) IB (μA) IE (mA) α β IS 700 1.000 10 690 1.000 1.020 580 5 0.235 780 10.10 120 820 1050 75.00 6.16 When operated in the active mode, a particular npn BJT conducts a collector current of 1 mA and has vBE = 0.70 V and iB = 10 μA. Use these data to create specific transistor models of the form shown in Fig. 6.5(a) to (d). 6.17 Using the npn transistor model of Fig. 6.5(b), consider the case of a transistor for which the base is connected to ground, the collector is connected to a 5-V dc source through a 2-k resistor, and a 2-mA current source is connected to the emitter with the polarity so that current is drawn out of the emitter terminal. If β = 100 and IS = 5 × 10−15 A, find the voltages at the emitter and the collector and calculate the base current. D 6.18 Consider an npn transistor operated in the active mode and represented by the model of Fig. 6.5(d). Let the transistor be connected as indicated by the equivalent circuit shown in Fig. 6.6(b). It is required to calculate the values of RB and RC that will establish a collector current IC of 0.5 mA and a collector-to-emitter voltage VCE of 1 V. The BJT is specified to have β = 50 and IS = 5 × 10−15 A. 6.19 An npn transistor has a CBJ with an area 100 times that of the EBJ. If IS = 10−15 A, find the voltage drop across EBJ and across CBJ when each is forward biased and conducting a current of 1 mA. Also find the forward current each junction would conduct when forward biased with 0.5 V. *6.20 We wish to investigate the operation of the npn transistor in saturation using the model of Fig. 6.9. Let IS = 10−15 A, vBE = 0.7 V, β = 100, and ISC/IS = 100. For each of three values of vCE (namely, 0.4 V, 0.3 V, and 0.2 V), find vBC, iBC, iBE, iB, iC, and iC/iB. Present your results in tabular form. Also find vCE that results in iC = 0. *6.21 Use Eqs. (6.14), (6.15), and (6.16) to show that an npn transistor operated in saturation exhibits a collector-to-emitter voltage, VCEsat, given by VCEsat = VT ln ISC 1 + βforced IS 1 − βforced/β Use this relationship to evaluate VCEsat for βforced = 50, 10, 5, and 1 for a transistor with β = 100 and with a CBJ area 100 times that of the EBJ. Present your results in a table. 6.22 Consider the pnp large-signal model of Fig. 6.11(b) applied to a transistor having IS = 10−14 A and β = 50. If the emitter is connected to ground, the base is connected to a current source that pulls 10 μA out of the base terminal, and the collector is connected to a negative supply of −5 V via a 8.2-k resistor, find the collector voltage, the emitter current, and the base voltage. 6.23 A pnp transistor has vEB = 0.7 V at a collector current of 1 mA. What do you expect vEB to become at iC = 10 mA? At iC = 100 mA? 6.24 A pnp transistor modeled with the circuit in Fig. 6.11 (b) is connected with its base at ground, collector at –2.0 V, and a 1-mA current is injected into its emitter. If the transistor is said to have β = 10, what are its base and collector currents? In which direction do they flow? If IS = 10−15 A, what voltage results at the emitter? What does the collector current become if a transistor with β = 1000 is substituted? (Note: The fact that the collector current changes by less than 10% for a large change in β illustrates that this is a good way to establish a specific collector current.) 6.25 A pnp power transistor operates with an emitter-to-collector voltage of 5 V, an emitter current of 5 A, and VEB = 0.8 V. For β = 20, what base current is required? What is IS for this transistor? Compare the emitter–base junction area of this transistor with that of a small-signal = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 6 PROBLEMS Problems 357 transistor that conducts iC = 1 mA with vEB = 0.70 V. How much larger is it? 6.26 While Fig. 6.5 provides four possible large-signal equivalent circuits for the npn transistor, only two equivalent circuits for the pnp transistor are provided in Fig. 6.11. Supply the missing two. 6.27 By analogy to the npn case shown in Fig. 6.9, give the equivalent circuit of a pnp transistor in saturation. Section 6.2: Current–Voltage Characteristics 6.28 For the circuits in Fig. P6.28, assume that the transistors have very large β. Some measurements have been made on these circuits, with the results indicated in the figure. Find the values of the other labeled voltages and currents. 6.29 Measurements on the circuits of Fig. P6.29 produce labeled voltages as indicated. Find the value of β for each transistor. 5k V2 5k (a) Figure P6.28 5.6 k V3 4V 2.4 k 2 15 k V4 I5 0V 20 k 9.1 k V7 0.7 V I6 3k (b) (c) (d) 200 k 2k (a) Figure P6.29 27 k⍀ + 3.0 V 750 ⍀ 6.3 V 45 k 7V 1.5 k (b) (c) = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 6 PROBLEMS 358 Chapter 6 Bipolar Junction Transistors (BJTs) 6.30 A very simple circuit for measuring β of an npn transistor is shown in Fig. P6.30. In a particular design, VCC is provided by a 9-V battery; M is a current meter with a 50-μA full scale and relatively low resistance that you can neglect for our purposes here. Assuming that the transistor has VBE = 0.7 V at IE = 1 mA, what value of RC would establish a resistor current of 1 mA? Now, to what value of β does a meter reading of full scale correspond? What is β if the meter reading is 1/5 of full scale? 1/10 of full scale? VCC D 6.33 Examination of the table of standard values for resistors with 5% tolerance in Appendix J reveals that the closest values to those found in the design of Example 6.2 are 5.1 k and 6.8 k . For these values, use approximate calculations (e.g., VBE 0.7 V and α 1) to determine the values of collector current and collector voltage that are likely to result. D 6.34 Design the circuit in Fig. P6.34 to establish IC = 0.2 mA and VC = 0.5 V. The transistor exhibits vBE of 0.8 V at iC = 1 mA, and β = 100. 1.5 V RC M RC IC VC RE Figure P6.30 6.31 Repeat Exercise 6.13 for the situation in which the power supplies are reduced to ±2.5 V. D 6.32 Design the circuit in Fig. P6.32 to establish a current of 0.5 mA in the emitter and a voltage of −0.5 V at the collector. The transistor vEB = 0.64 V at IE = 0.1 mA, and β = 100. To what value can RC be increased while the collector current remains unchanged? 1.5 V Figure P6.34 6.35 For each of the circuits shown in Fig. P6.35, find the emitter, base, and collector voltages and currents. Use β = 50, but assume VBE = 0.8 V independent of current level. 1.5 V 1.5 V 2.5 V 2.7 k 2k RE Q1 Q2 RC 2.5 V Figure P6.32 2.7 k 1.5 V (a) Figure P6.35 2k 1.5 V (b) = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem Problems 359 CHAPTER 6 PROBLEMS 3V 3V 1.0 V 10 k Q3 1.5 V 8.2 k Q4 5.6 k 4.7 k 6.41 Use Eq. (6.18) to plot iC versus vCE for an npn transistor having IS = 10−15 A and VA = 100 V. Provide curves for vBE = 0.65, 0.70, 0.72, 0.73, and 0.74 volts. Show the characteristics for vCE up to 15 V. *6.42 In the circuit shown in Fig. P6.42, current source I is 1.1 mA, and at 25°C vBE = 680 mV at iE = 1 mA. At 25°C with β = 100, what currents flow in R1 and R2? What voltage would you expect at node E? Noting that the temperature coefficient of vBE for IE constant is −2 mV/°C, what is the TC of vE? For an ambient temperature of 75°C, what voltage would you expect at node E? Clearly state any simplifying assumptions you make. (c) (d) Figure P6.35 continued R2 68 k 6.36 The current ICBO of a small transistor is measured to be 10 nA at 25°C. If the temperature of the device is raised to 125°C, what do you expect ICBO to become? 6.37 Augment the model of the npn BJT shown in Fig. 6.19(a) by a current source representing ICBO. Assume that ro is very large and thus can be neglected. In terms of this addition, what do the terminal currents iB, iC, and iE become? If the base lead is open-circuited while the emitter is connected to ground, and the collector is connected to a positive supply, find the emitter and collector currents. 6.38 A BJT whose emitter current is fixed at 1 mA has a base–emitter voltage of 0.70 V at 25°C. What base–emitter voltage would you expect at 0°C? At 100°C? 6.39 A particular pnp transistor operating at an emitter current of 0.5 mA at 20°C has an emitter–base voltage of 692 mV. R1 6.8 k E I Figure P6.42 6.43 For a particular npn transistor operating at a vBE of 680 mV and IC = 1 mA, the iC–vCE characteristic has a slope of 0.8 × 10−5 . To what value of output resistance does this correspond? What is the value of the Early voltage for this transistor? For operation at 10 mA, what would the output resistance become? (a) What does vEB become if the junction temperature rises to 50°C? (b) If the transistor is operated at a fixed emitter–base voltage of 700 mV, what emitter current flows at 20°C? At 50°C? 6.40 Consider a transistor for which the base–emitter voltage drop is 0.7 V at 10 mA. What current flows for vBE = 0.5 V? Evaluate the ratio of the slopes of the iC–vBE curve at vBE = 700 mV and at vBE = 500 mV. The large ratio confirms the point that the BJT has an “apparent threshold” at vBE 0.5 V. 6.44 For a BJT having an Early voltage of 50 V, what is its output resistance at 1 mA? At 100 μA? 6.45 Measurements of the iC–vCE characteristic of a small-signal transistor operating at vBE = 710 mV show that iC = 1.1 mA at vCE = 5 V and that iC = 1.3 mA at vCE = 15 V. What is the corresponding value of iC near saturation? At what value of vCE is iC = 1.2 mA? What is the value of the Early voltage for this transistor? What is the output resistance that corresponds to operation at vBE = 710 mV? = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 6 PROBLEMS 360 Chapter 6 Bipolar Junction Transistors (BJTs) 6.46 Give the pnp equivalent circuit models that correspond 5V to those shown in Fig. 6.19 for the npn case. 6.47 A BJT operating at iB = 10 μA and iC = 1.0 mA undergoes a reduction in base current of 1.0 μA. It is found that when vCE is held constant, the corresponding reduction in collector current is 0.08 mA. What are the values of β and the incremental β or βac that apply? If the base current is increased from 10 μA to 12 μA and vCE is increased from 8 V to 10 V, what collector current results? Assume VA = 100 V. 6.48 For the circuit in Fig. P6.48 let VCC = 10 V, RC = 1 k , and RB = 10 k . The BJT has β = 50. Find the value of VBB that results in the transistor operating RB 10 k VC 1k (a) in the active mode with VC = 2 V; (b) at the edge of saturation; (c) deep in saturation with βforced = 10. Assume VBE 0.7 V. VCC Figure P6.50 Section 6.3: BJT Circuits at DC 6.51 The transistor in the circuit of Fig. P6.51 has a very high β. Find VE and VC for VB (a) +2.0 V, (b) +1.7 V, and (c) 0 V. 3V VBB IC RC VC RB 1k VB VC Figure P6.48 VE 1k D *6.49 Consider the circuit of Fig. P6.48 for the case VBB = VCC. If the BJT is saturated, use the equivalent circuit of Fig. 6.21 to derive an expression for βforced in terms of VCC and RB/RC . Also derive an expression for the total power dissipated in the circuit. For VCC = 5 V, design the circuit to obtain operation at a forced β as close to 10 as possible while limiting the power dissipation to no larger than 20 mW. Use 1% resistors (see Appendix J). Figure P6.51 6.52 The transistor in the circuit of Fig. P6.51 has a very high β. Find the highest value of VB for which the transistor still operates in the active mode. Also, find the value of VB for which the transistor operates in saturation with a forced β of 2. 6.50 The pnp transistor in the circuit in Fig. P6.50 has β = 50. Show that the BJT is operating in the saturation mode and find βforced and VC. To what value should RB be increased in order for the transistor to operate at the edge of saturation? 6.53 Consider the operation of the circuit shown in Fig. P6.53 for VB at –1 V, 0 V, and +1 V. Assume that β is very high. What values of VE and VC result? At what value of VB does the emitter current reduce to one-tenth of = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 6 PROBLEMS Problems 361 its value for VB = 0 V? For what value of VB is the transistor just at the edge of conduction? (vBE = 0.5 V) What values of VE and VC correspond? For what value of VB does the transistor reach the edge of saturation? What values of VC and VE correspond? Find the value of VB for which the transistor operates in saturation with a forced β of 2. 3V D 6.55 Consider the circuit in Fig. P6.51 with the base voltage VB obtained using a voltage divider across the 3-V supply. Assuming the transistor β to be very large (i.e., ignoring the base current), design the voltage divider to obtain VB = 1.2 V. Design for a 0.1-mA current in the voltage divider. Now, if the BJT β = 100, analyze the circuit to determine the collector current and the collector voltage. 1k VC VB 6.56 A single measurement indicates the emitter voltage of the transistor in the circuit of Fig. P5.56 to be 1.0 V. Under the assumption that VBE = 0.7 V, what are VB, IB, IE, IC, VC, β, and α ? (Note: Isn’t it surprising what a little measurement can lead to?) VE 3V 1k 3V Figure P6.53 VB 6.54 For the transistor shown in Fig. P6.54, assume α 1 and vBE = 0.5 V at the edge of conduction. What are the values 50 k of VE and VC for VB = 0 V? For what value of VB does the transistor cut off? Saturate? In each case, what values of VE and VC result? +5 V Figure P6.56 5k VE VC 5k 3V 4 mA VC 1k VB 2 mA VE 1k –5 V Figure P6.54 D 6.57 Design a circuit using a pnp transistor for which α 1 using two resistors connected appropriately to ±3 V so that IE = 0.5 mA and VBC = 1 V. What exact values of RE and RC would be needed? Now, consult a table of standard 5% resistor values (e.g., that provided in Appendix J) to select suitable practical values. What values of resistors have you chosen? What are the values of IE and VBC that result? 6.58 In the circuit shown in Fig. P6.58, the transistor has β = 40. Find the values of VB, VE, and VC. If RB is raised to 100 k , what voltages result? With RB = 100 k , what value of β would return the voltages to the values first calculated? = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 6 PROBLEMS 362 Chapter 6 Bipolar Junction Transistors (BJTs) 3V RE 2.2 k 6.61 For the circuits in Fig. P6.61, find values for the labeled node voltages and branch currents. Assume β to be very high. VB RB 20 k VE VC RC 2.2 k 3V 3.6 k V2 43 k 3V 3.6 k V3 3V Figure P6.58 6.59 In the circuit shown in Fig. P6.58, the transistor has β = 50. Find the values of VB, VE, and VC, and verify that the transistor is operating in the active mode. What is the largest value that RC can have while the transistor remains in the active mode? V1 0.5 mA (a) I4 4.7 k 3V (b) 6.60 For the circuit in Fig. P6.60, find VB, VE, and VC for RB = 100 k , 10 k , and 1 k . Let β = 100. 3V 3V Figure P6.60 43 k V6 3.6 k V7 0.75 V V5 4.7 k 110 k 6.2 k V8 V9 10 k 3V (c) Figure P6.61 3V (d) = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 6 PROBLEMS 3V 180 k 6.2 k V11 V10 300 k V12 10 k Problems 363 D 6.64 The pnp transistor in the circuit of Fig. P6.64 has β = 50. Find the value for RC to obtain VC = +2 V. What happens if the transistor is replaced with another having β = 100? Give the value of VC in the latter case. 3V 3V (e) Figure P6.61 continued Figure P6.64 *6.62 Repeat the analysis of the circuits in Problem 6.61 using β = 100. Find all the labeled node voltages and branch currents. D **6.63 It is required to design the circuit in Fig. P6.63 so that a current of 1 mA is established in the emitter and a voltage of −1 V appears at the collector. The transistor type used has a nominal β of 100. However, the β value can be as low as 50 and as high as 150. Your design should ensure that the specified emitter current is obtained when β = 100 and that at the extreme values of β the emitter current does not change by more than 10% of its nominal value. Also, design for as large a value for RB as possible. Give the values of RB, RE, and RC to the nearest kilohm. What is the expected range of collector current and collector voltage corresponding to the full range of β values? +5V E ***6.65 Consider the circuit shown in Fig. P6.65. It resembles that in Fig. 6.30 but includes other features. First, note diodes D1 and D2 are included to make design (and analysis) easier and to provide temperature compensation for the emitter–base voltages of Q1 and Q2. Second, note resistor R, whose purpose is to provide negative feedback (more on this later in the book!). Using VBE and VD = 0.7 V independent of current, and β = ∞, find the voltages VB1, VE1, VC1, VB2, VE2, and VC2, initially with R open-circuited and then with R connected. Repeat for β = 100, with R open-circuited initially, then connected. 9V 2k 100 80 k D2 Q2 Q1 R D1 2k C 40 k 2k 100 –5V Figure P6.63 Figure P6.65 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 6 PROBLEMS 364 Chapter 6 Bipolar Junction Transistors (BJTs) *6.66 For the circuit shown in Fig. P6.66, find the labeled node voltages for: (a) β = ∞ (b) β = 100 are 0.5 mA, 0.5 mA, and 1 mA, respectively, and V3 = 0, V5 = −2 V, and V7 = 1 V. For each resistor, select the nearest standard value utilizing the table of standard values for 5% resistors in Appendix J. Now, for β = 100, find the values of V3, V4, V5, V6, and V7. 3V 9.1 k V2 5.1 k V1 Q1 V5 *6.68 For the circuit in Fig. P6.68, find VB and VE for vI = 0 V, +2 V, –2.5 V, and –5 V. The BJTs have β = 50. 2.5 V 100 k V3 Q2 9.1 k V4 4.3 k 3V Figure P6.66 D *6.67 Using β = ∞, design the circuit shown in Fig. P6.67 so that the emitter currents of Q1, Q2, and Q3 5V R2 Q1 V3 V2 R1 R3 V4 Q2 V5 R4 R5 V7 Q3 V6 R6 Figure P6.68 2.5 V **6.69 All the transistors in the circuits of Fig. P6.69 are specified to have a minimum β of 50. Find approximate values for the collector voltages and calculate forced β for each of the transistors. (Hint: Initially, assume all transistors are operating in saturation, and verify the assumption.) 5V Figure P6.67 = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem 5V 5V 5V Figure P6.69 Problems 365 5V 20 CHAPTER 6 PROBLEMS CHAPTER 7 Transistor Amplifiers Introduction 367 7.1 Basic Principles 368 7.2 Small-Signal Operation and Models 383 7.3 Basic Configurations 423 7.4 Biasing 454 7.5 Discrete-Circuit Amplifiers 467 Summary 479 Problems 480 IN THIS CHAPTER YOU WILL LEARN 1. How the transistor (a MOSFET or a BJT) can be used to make an amplifier. 2. How to obtain linear amplification from the fundamentally nonlinear MOS and bipolar transistor. 3. How to model the linear operation of a transistor around a bias point by an equivalent circuit that can be used in the analysis and design of transistor amplifiers. 4. The three basic ways to connect a MOSFET or a BJT to construct amplifiers with different properties. 5. Practical circuits for MOS and bipolar transistor amplifiers that can be constructed using discrete components. Introduction Having studied the two major transistor types, the MOSFET (Chapter 5) and the BJT (Chapter 6), we now begin the study of their application. There are two distinctly different kinds of transistor application: as a switch, in the design of digital circuits (Chapters 14–16) and as a controlled source, in the design of amplifiers for analog circuits. This chapter and the subsequent six focus on the latter application, namely, the use of the transistor in the design of a variety of amplifier types. Since the basic principles that underlie the use of the MOSFET and the BJT in amplifier design are the same, the two devices are studied together in this chapter. Besides providing some economy in presentation, this unified study enables us to make important comparisons between MOS and bipolar amplifiers. The bulk of this chapter is concerned with the fundamental principles and concepts that are the basis for the application of transistors in amplifier design: We study in detail the models that are used to represent both transistor types in the analysis and design of small-signal linear amplifiers. We also study the three basic configurations in which each of the two transistor types can be connected to realize an amplifier. The chapter concludes with examples of discrete-circuit amplifiers. These are circuits that can be assembled using discrete transistors, resistors, and capacitors on printed-circuit boards (PCBs). They predominantly use BJTs, and their design differs in significant ways from the design of integrated-circuit (IC) amplifiers. The latter predominantly use MOSFETs, and their study begins in Chapter 8. However, the fundamental principles and concepts introduced in this chapter apply equally well to both discrete and integrated amplifiers. 367 368 Chapter 7 Transistor Amplifiers 7.1 Basic Principles 7.1.1 The Basis for Amplifier Operation The basis for the application of the transistor (a MOSFET or a BJT) in amplifier design is that when the device is operated in the active region, a voltage-controlled current source is realized. Specifically, when a MOSFET is operated in the saturation or pinch-off region, also referred to in this chapter as the active region, the voltage between gate and source, vGS, controls the drain current iD according to the square-law relationship which, for an NMOS transistor, is expressed as iD = 1 2 kn (vGS − Vtn)2 (7.1) We note that in this first-order model of MOSFET operation, the drain current iD does not depend on the drain voltage vDS because the channel is pinched off at the drain end,1 thus “isolating” the drain. Similarly, when a BJT is operated in the active region, the base-emitter voltage vBE controls the collector current iC according to the exponential relationship which, for an npn transistor, is expressed as iC = IS evBE/VT (7.2) Here, this first-order model of BJT operation indicates that the collector current iC does not depend on the collector voltage vCE because the collector–base junction is reverse biased, thus “isolating” the collector. Figure 7.1 shows an NMOS transistor and an npn transistor operating in the active mode. Observe that for the NMOS transistor, the pinch-off condition is ensured by keeping vDS ≥ vOV . Since the overdrive voltage vOV = vGS −Vtn, this condition implies that vGD ≤ Vtn, which indeed ensures channel pinch-off at the drain end. Similarly, for the npn transistor in Fig. 7.1(b), the CBJ reverse-bias condition is ensured by keeping vCE ≥ 0.3 V. Since vBE is usually in the vicinity of 0.7 V, vBC is thus kept vGD ≤ Vtn Channel pinched off at drain iD vDS ≥ vOV vGS = Vtn vOV vBC ≤ 0.4 V CBJ reverse biased iC vCE ≥ 0.3 V vBE iD = 1 2 kn (vGS – Vtn)2 iC = IS evBE/VT (a) (b) Figure 7.1 Operating (a) an NMOS transistor and (b) an npn transistor in the active mode. Note that vGS = Vtn + vOV and vDS ≥ vOV ; thus vGD ≤ Vtn, which ensures channel pinch-off at the drain end. Similarly, vBE 0.7 V, and vCE ≥ 0.3 V results in vBC ≤ 0.4 V, which is sufficient to keep the CBJ from conducting. 1To focus on essentials, we shall neglect the Early effect until a later point. 7.1 Basic Principles 369 smaller than 0.4 V, which is sufficient to prevent this relatively large-area junction from conducting. Although we used NMOS and npn transistors to illustrate the conditions for active-mode operation, similar conditions apply for PMOS and pnp transistors, as studied in Chapters 5 and 6, respectively. Finally, we note that the control relationships in Eqs. (7.1) and (7.2) are nonlinear. Nevertheless, we shall shortly devise a technique for obtaining almost-linear amplification from these fundamentally nonlinear devices. 7.1.2 Obtaining a Voltage Amplifier From the above we see that the transistor is basically a transconductance amplifier: that is, an amplifier whose input signal is a voltage and whose output signal is a current. More commonly, however, one is interested in voltage amplifiers. A simple way to convert a transconductance amplifier to a voltage amplifier is to pass the output current through a resistor and take the voltage across the resistor as the output. Doing this for a MOSFET results in the simple amplifier circuit shown in Fig. 7.2(a). Here vGS is the input voltage, RD (known as a load resistance) converts the drain current iD to a voltage (iDRD), and VDD is the supply voltage that powers up the amplifier and, together with RD, establishes operation in the active region, as will be shown shortly. In the amplifier circuit of Fig. 7.2(a) the output voltage is taken between the drain and ground, rather than simply across RD. This is done because of the need to maintain a common ground reference between the input and the output. The output voltage vDS is given by vDS = VDD − iDRD (7.3) Thus it is an inverted version (note the minus sign) of iDRD that is shifted by the constant value of the supply voltage VDD. An exactly similar arrangement applies for the BJT amplifier, as illustrated in Fig. 7.2(c). Here the output voltage vCE is given by vCE = VCC − iC RC (7.4) Cut- Saturation off or Active region Triode iD A + ϩϪ vO = vDS VDSΗB B – C 0 Vt (a) (b) Figure 7.2 (a) An NMOS amplifier and (b) its VTC; and (c) an npn amplifier and (d) its VTC. 370 Chapter 7 Transistor Amplifiers VCC iC RC vCE Cutoff Active mode VCC A Saturation vBE ϩ Ϫ (c) Figure 7.2 continued Ӎ 0.3 V 0 0.5 V Edge of saturation B vBE (d) 7.1.3 The Voltage-Transfer Characteristic (VTC) A useful tool that provides insight into the operation of an amplifier circuit is its voltage-transfer characteristic (VTC). This is simply a plot (or a clearly labeled sketch) of the output voltage versus the input voltage. For the MOS amplifier in Fig. 7.2(a), this is the plot of vDS versus vGS shown in Fig. 7.2(b). Observe that for vGS < Vt, the transistor is cut off, iD = 0 and, from Eq. (7.3), vDS = VDD. As vGS exceeds Vt, the transistor turns on and vDS decreases. However, since initially vDS is still high, the MOSFET will be operating in saturation or the active region. This continues as vGS is increased until the value of vGS is reached that results in vDS becoming lower than vGS by Vt volts [point B on the VTC in Fig. 7.2(b)]. For vGS greater than that at point B, the transistor operates in the triode region and vDS decreases more slowly. The VTC in Fig. 7.2(b) indicates that the segment of greatest slope (hence potentially the largest amplifier gain) is that labeled AB, which corresponds to operation in the active region. When a MOSFET is operated as an amplifier, its operating point is confined to the segment AB at all times. An expression for the segment AB can be obtained by substituting for iD in Eq. (7.3) by its active-region value from Eq. (7.1), thus vDS = VDD − 1 2 knRD(vGS − Vt )2 (7.5) This is obviously a nonlinear relationship. Nevertheless, linear (or almost-linear) amplification can be obtained by using the technique of biasing the MOSFET. Before considering biasing, however, it is useful to determine the coordinates of point B, which is at the boundary between the saturation and the triode regions of operation. These can be obtained by substituting in Eq. (7.5), vGS = VGS B and vDS = VDS B = VGS B − Vt. The result is VGS B = Vt + 2knRDVDD + 1 − 1 knRD (7.6) Point B can be alternatively characterized by the overdrive voltage VOV B ≡ VGS B − Vt = 2knRDVDD + 1 − 1 knRD and VDS B = VOV B 7.1 Basic Principles 371 (7.7) (7.8) EXERCISE 7.1 Consider the amplifier of Fig. 7.2(a) with VDD = 1.8 V, RD = 17.5 k , and with a MOSFET specified to have Vt = 0.4 V, kn = 4 mA/V2, and λ = 0. Determine the coordinates of the end points of the active-region segment of the VTC. Also, determine VDS C assuming VGS C = VDD. Ans. A: 0.4 V, 1.8 V; B: 0.613 V, 0.213 V;VDS C = 18 mV An exactly similar development applies to the BJT case. This is illustrated in Fig. 7.2(c) and (d). In this case, over the active-region or amplifier segment AB, the output voltage vCE is related to the input voltage vBE by vCE = VCC − RC IS evBE/VT (7.9) Here also, the input–output relationship is nonlinear. Nevertheless, linear (or almost-linear) amplification can be obtained by using the biasing technique discussed next. 7.1.4 Obtaining Linear Amplification by Biasing the Transistor Biasing enables us to obtain almost-linear amplification from the MOSFET and the BJT. The technique is illustrated for the MOSFET case in Fig. 7.3(a). A dc voltage VGS is selected to obtain operation at a point Q on the segment AB of the VTC. How to select an appropriate location for the bias point Q will be discussed shortly. For the time being, observe that the coordinates of Q are the dc voltages VGS and VDS, which are related by VDS = VDD − 1 2 knRD(VGS − Vt )2 (7.10) Point Q is known as the bias point or the dc operating point. Also, since at Q no signal component is present, it is also known as the quiescent point (which is the origin of the symbol Q). Next, the signal to be amplified, vgs, a function of time t, is superimposed on the bias voltage VGS, as shown in Fig. 7.4(a). Thus the total instantaneous value of vGS becomes vGS(t) = VGS + vgs(t) The resulting vDS(t) can be obtained by substituting for vGS(t) into Eq. (7.5). Graphically, we can use the VTC to obtain vDS(t) point by point, as illustrated in Fig. 7.4(b). Here we show 372 Chapter 7 Transistor Amplifiers VDD ID RD vDS A VDD + VGS – (a) + VDS Q VDS – B 0 Vt VGS (b) C VDD vGS Figure 7.3 Biasing the MOSFET amplifier at a point Q located on the segment AB of the VTC. the case of vgs being a triangular wave of “small” amplitude. Specifically, the amplitude of vgs is small enough to restrict the excursion of the instantaneous operating point to a short, almost-linear segment of the VTC around the bias point Q. The shorter the segment, the greater the linearity achieved, and the closer to an ideal triangular wave the signal component at the output, vds, will be. This is the essence of obtaining linear amplification from the nonlinear MOSFET. Before leaving Fig. 7.4(b) we wish to draw the reader’s attention to the consequence of increasing the amplitude of the signal vgs. As the instantaneous operating point will no longer be confined to the almost-linear segment of the VTC, the output signal vds will deviate from its ideal triangular shape; that is, it will exhibit nonlinear distortion. Worse yet, if the input signal amplitude becomes sufficiently large, the instantaneous operating point may leave the segment AB altogether. If this happens at the negative peaks of vgs, the transistor will cut off for a portion of the cycle and the positive peaks of vds will be “clipped off.” If it occurs at the positive peaks of vgs, the transistor will enter the triode region for a portion of the cycle, and the negative peaks of vds will become flattened. It follows that the selection of the location of the bias point Q can have a profound effect on the maximum allowable amplitude of vds, referred to as the allowable signal swing at the output. We will have more to say later on this important point. An exactly parallel development can be applied to the BJT amplifier. In fact, all we need to do is replace the NMOS transistor in Figs. 7.3 and 7.4 with an npn transistor and change the voltage and current symbols to their BJT counterparts. The resulting bias point Q will be characterized by dc voltages VBE and VCE, which are related by VCE = VCC − RC IS eVBE/VT (7.11) and a dc current IC, IC = IS eVBE/VT (7.12) Also, superimposing a small-signal vbe on the dc bias voltage VBE results in vBE(t) = VBE + vbe(t) VDD iD RD vDS ϩ vgs ϩ Ϫ vGS VGS Ϫ (a) vDS VDD A VDS Slope at Q = voltage gain vds Q Time B Vt VGS vgs C VDD vGS 7.1 Basic Principles 373 Time (b) Figure 7.4 The MOSFET amplifier with a small time-varying signal vgs(t) superimposed on the dc bias voltage VGS. The MOSFET operates on a short almost-linear segment of the VTC around the bias point Q and provides an output voltage vds = Av vgs. 374 Chapter 7 Transistor Amplifiers which can be substituted into Eq. (7.9) to obtain the total instantaneous value of the output voltage vCE(t). Here again, almost-linear operation is obtained by keeping vbe small enough to restrict the excursion of the instantaneous operating point to a short, almost-linear segment of the VTC around the bias point Q. Similar comments also apply to the maximum allowable signal swing at the output. 7.1.5 The Small-Signal Voltage Gain The MOSFET Case Consider the MOSFET amplifier in Fig. 7.4(a). If the input signal vgs is kept small, the corresponding signal at the output vds will be nearly proportional to vgs with the constant of proportionality being the slope of the almost-linear segment of the VTC around Q. This is the voltage gain of the amplifier, and its value can be determined by evaluating the slope of the tangent to the VTC at the bias point Q, Utilizing Eq. (7.5) we obtain Av = dvDS dvGS vGS = VGS (7.13) Av = −kn(VGS − Vt)RD which can be expressed in terms of the overdrive voltage at the bias point, VOV , as Av = −knVOV RD (7.14) (7.15) We make the following observations on this expression for the voltage gain. 1. The gain is negative, which signifies that the amplifier is inverting; that is, there is a 180° phase shift between the input and the output. This inversion is obvious in Fig. 7.4(b) and should have been anticipated from Eq. (7.5). 2. The gain is proportional to the load resistance RD, to the transistor transconductance parameter kn, and to the overdrive voltage VOV . This all makes intuitive sense. Another simple and insightful expression for the voltage gain Av can be derived by recalling that the dc current in the drain at the bias point is related to VOV by ID = 1 2 kn VO2V This equation can be combined with Eq. (7.15) to obtain Av = − IDRD VOV /2 (7.16) That is, the gain is simply the ratio of the dc voltage drop across the load resistance RD to VOV /2. It can be expressed in the alternative form Av = − VDD − VDS VOV /2 (7.17) Since the maximum slope of the VTC in Fig. 7.4(b) occurs at point B, the maximum gain magnitude |Avmax| is obtained by biasing the transistor at point B, |Avmax | = VDD − VDS VOV /2 B B 7.1 Basic Principles 375 and since VDS B = VOV , B |Avmax | = VDD − VOV VOV /2 B B (7.18) where VOV is given by Eq. (7.7). Of course, this result is only of theoretical importance since B biasing at B would leave no room for negative signal swing at the output. Nevertheless, the result in Eq. (7.18) is valuable as it provides an upper bound on the magnitude of voltage gain achievable from this basic amplifier circuit. As an example, for a discrete-circuit amplifier operated with VDD = 5 V and VOV B = 0.5 V, the maximum achievable gain is 18 V/V. An integrated-circuit amplifier utilizing a modern submicron MOSFET operated with VDD = 1.3 V and with VOV B = 0.2 V realizes a maximum gain of 11 V/V. Finally, note that to maximize the gain, the bias point Q should be as close to point B as possible, consistent with the required signal swing at the output. This point will be explored further in the end-of-chapter problems. Example 7.1 Consider the amplifier circuit shown in Fig. 7.4(a). The transistor is specified to have Vt = 0.4 V, kn = 0.4 mA/V2, W/L = 10, and λ = 0. Also, let VDD = 1.8 V, RD = 17.5 k , and VGS = 0.6 V. (a) For vgs = 0 (and hence vds = 0), find VOV , ID, VDS, and Av . (b) What is the maximum symmetrical signal swing allowed at the drain? Hence, find the maximum allowable amplitude of a sinusoidal vgs. Solution (a) With VGS = 0.6 V, VOV = 0.6 − 0.4 = 0.2 V. Thus, 1 ID = 2 kn W L VO2V = 1 × 0.4 × 10 × 0.22 = 0.08 mA 2 VDS = VDD − RDID = 1.8 − 17.5 × 0.08 = 0.4 V Since VDS is greater than VOV , the transistor is indeed operating in saturation. The voltage gain can be found from Eq. (7.15), Av = −knVOV RD = −0.4 × 10 × 0.2 × 17.5 = −14 V/V An identical result can be found using Eq. (7.17). (b) Since VOV = 0.2 V and VDS = 0.4 V, we see that the maximum allowable negative signal swing at the drain is 0.2 V. In the positive direction, a swing of +0.2 V would not cause the transistor to 376 Chapter 7 Transistor Amplifiers Example 7.1 continued cut off (since the resulting vDS would be still lower than VDD) and thus is allowed. Thus the maximum symmetrical signal swing allowable at the drain is ±0.2 V. The corresponding amplitude of vgs can be found from vˆ gs = vˆ ds |Av | = 0.2 V 14 = 14.2 mV Since vˆgs VOV , the operation will be reasonably linear (more on this in later sections). Greater insight into the issue of allowable signal swing can be obtained by examining the signal waveforms shown in Fig. 7.5. Note that for the MOSFET to remain in saturation at the negative peak of vds, we must ensure that vDSmin ≥ vGSmax − Vt that is, 0.4 − |Av |vˆgs ≥ 0.6 + vˆgs − 0.4 which results in vˆ gs ≤ 0.2 |Av | + 1 = 13.3 mV This result differs slightly from the one obtained earlier. vGS vgs VGS vGSmax = VGS ϩ vˆgs Vt 0 vDS VDS t vds vDSmin = VDS Ϫ vˆds 0 t Figure 7.5 Signal waveforms at gate and drain for the amplifier in Example 7.1. Note that to ensure operation in the saturation region at all times, vDSmin ≥ vGSmax−Vt. 7.1 Basic Principles 377 EXERCISE D7.2 For the amplifier circuit studied in Example 7.1, create two alternative designs, each providing a voltage gain of −10 by (a) changing RD while keeping VOV constant and (b) changing VOV while keeping RD constant. For each design, specify VGS, ID, RD, and VDS. Ans. (a) 0.6 V, 0.08 mA, 12.5 k , 0.8 V; (b) 0.54 V, 0.04 mA, 17.5 k , 1.1 V The BJT Case A similar development can be used to obtain the small-signal voltage gain of the BJT amplifier shown in Fig. 7.6, Av = d vCE d vBE vBE =VBE Utilizing Eq. (7.9) together with Eq. (7.12), we obtain (7.19) Av = − IC VT RC (7.20) We make the following observations on this expression for the voltage gain: 1. The gain is negative, which signifies that the amplifier is inverting; that is, there is a 180° phase shift between the input and the output. This inversion should have been anticipated from Eq. (7.9). 2. The gain is proportional to the collector bias current IC and to the load resistance RC. Additional insight into the voltage gain Av can be obtained by expressing Eq. (7.20) as Av = − ICRC VT (7.21) VCC iC RC vbe VBE vBE vCE Figure 7.6 BJT amplifier biased at a point Q, with a small voltage signal vbe superimposed on the dc bias voltage VBE. The resulting output signal vce appears superimposed on the dc collector voltage VCE. The amplitude of vce is larger than that of vbe by the voltage gain Av . 378 Chapter 7 Transistor Amplifiers That is, the gain is the ratio of the dc voltage drop across the load resistance RC to the physical constant VT (recall that the thermal voltage VT 25 mV at room temperature). This relationship is similar in form to that for the MOSFET (Eq. 7.16) except that here the denominator is a physical constant (VT ) rather than a design parameter (VOV /2). Usually, VOV /2 is larger than (VT ), thus we can obtain higher voltage gain from the BJT amplifier than from the MOSFET amplifier. This should not be surprising, as the exponential iC–vBE relationship is much steeper than the square-law relationship iD–vGS. The gain Av in Eq. (7.21) can be expressed alternately as Av = − VCC − VCE VT (7.22) from which we see that maximum gain is achieved when VCE is at its minimum value of about 0.3 V, |Avmax | = VCC − VT 0.3 (7.23) Here again, this is only a theoretical maximum, since biasing the BJT at the edge of saturation leaves no room for negative signal swing at the output. Equation (7.23) nevertheless provides an upper bound on the voltage gain achievable from the basic BJT amplifier. As an example, for VCC = 5 V, the maximum gain is 188 V/V, considerably larger than in the MOSFET case. For modern low-voltage technologies, a VCC of 1.3 V provides a gain of 40 V/V, again much larger than the MOSFET case. The reader should not, however, jump to the conclusion that the BJT is preferred to the MOSFET in the design of modern integrated-circuit amplifiers; in fact, the opposite is true, as we shall see in Chapter 8 and beyond. Finally, we conclude from Eq. (7.22) that to maximize |Av | the transistor should be biased at the lowest possible VCE consistent with the desired value of negative signal swing at the output. Example 7.2 Consider an amplifier circuit using a BJT having IS = 10−15 A, a collector resistance RC = 6.8 k , and a power supply VCC = 10 V. (a) Determine the value of the bias voltage VBE required to operate the transistor at VCE = 3.2 V. What is the corresponding value of IC? (b) Find the voltage gain Av at this bias point. If an input sine-wave signal of 5-mV peak amplitude is superimposed on VBE, find the amplitude of the output sine-wave signal (assume linear operation). (c) Find the positive increment in vBE (above VBE) that drives the transistor to the edge of saturation, where vCE = 0.3 V. (d) Find the negative increment in vBE that drives the transistor to within 1% of cutoff (i.e., to vCE = 0.99VCC). Solution (a) IC = VCC − VCE RC = 10 − 3.2 = 1 mA 6.8 The value of VBE can be determined from 1 × 10−3 = 10−15 eVBE /VT which results in VBE = 690.8 mV (b) Av = − VCC − VCE VT = 10 − 3.2 = −272 V/V 0.025 vˆce = 272 × 0.005 = 1.36 V (c) For vCE = 0.3 V, 10 − 0.3 iC = 6.8 = 1.617 mA To increase iC from 1 mA to 1.617 mA, vBE must be increased by vBE = VT ln 1.617 1 = 12 mV (d) For vCE = 0.99VCC = 9.9 V, 10 − 9.9 iC = 6.8 = 0.0147 mA To decrease iC from 1 mA to 0.0147 mA, vBE must change by 0.0147 vBE = VT ln 1 = −105.5 mV 7.1 Basic Principles 379 380 Chapter 7 Transistor Amplifiers EXERCISE 7.3 For the situation described in Example 7.2, while keeping IC unchanged at 1 mA, find the value of RC that will result in a voltage gain of −320 V/V. What is the largest negative signal swing allowed at the output (assume that vCE is not to decrease below 0.3 V)? What (approximately) is the corresponding input signal amplitude? (Assume linear operation.) Ans. 8 k ; 1.7 V; 5.3 mV 7.1.6 Determining the VTC by Graphical Analysis Figure 7.7 shows a graphical method for determining the VTC of the amplifier of Fig. 7.4(a). Although graphical analysis of transistor circuits is rarely employed in practice, it is useful to us at this stage for gaining greater insight into circuit operation, especially in answering the question of where to locate the bias point Q. The graphical analysis is based on the observation that for each value of vGS, the circuit will be operating at the point of intersection of the iD−vDS graph corresponding to the particular Figure 7.7 Graphical construction to determine the voltage-transfer characteristic of the amplifier in Fig. 7.4(a). VDD 0 RD vDS = VDD vGS ≤ Vt (a) 7.1 Basic Principles 381 VDD RD vDS = VDSΗC rDS vGS = VDD (b) Figure 7.8 Operation of the MOSFET in Fig. 7.4(a) as a switch: (a) open, corresponding to point A in Fig. 7.7; (b) closed, corresponding to point C in Fig. 7.7. The closure resistance is approximately equal to rDS because VDS is usually very small. value of vGS and the straight line representing Eq. (7.3), which can be rewritten in the form iD = VDD RD − 1 RD vDS (7.24) The straight line representing this relationship is superimposed on the iD−vDS characteristics in Fig. 7.7. It intersects the horizontal axis at vDS = VDD and has a slope of −1/RD. Since this straight line represents in effect the load resistance RD, it is called the load line. The VTC is then determined point by point. Note that we have labeled four important points: point A at which vGS = Vt, point Q at which the MOSFET can be biased for amplifier operation (vGS = VGS and vDS = VDS), point B at which the MOSFET leaves saturation and enters the triode region, and point C, which is deep into the triode region and for which vGS = VDD. If the MOSFET is to be used as a switch, then operating points A and C are applicable: At A the transistor is off (open switch), and at C the transistor operates as a low-valued resistance rDS and has a small voltage drop (closed switch). The incremental resistance at point C is also known as the closure resistance. The operation of the MOSFET as a switch is illustrated in Fig. 7.8. A detailed study of the application of the MOSFET as a switch is undertaken in Chapter 14, dealing with CMOS digital logic circuits. The graphical analysis method above can be applied to determine the VTC of the BJT amplifier in Fig. 7.2(c). Here point A, Fig. 7.2(d), corresponds to the BJT just turning on (vBE 0.5 V) and point B corresponds to the BJT leaving the active region and entering the saturation region. If the BJT is to be operated as a switch, the two modes of operation are cutoff (open switch) and saturation (closed switch). As discussed in Section 6.2, in saturation, the BJT has a small closure resistance RCEsat as well as an offset voltage. More seriously, switching the BJT out of its saturation region can require a relatively long delay time to ensure the removal of the charge stored in the BJT base region. This phenomenon has made the BJT much less attractive in digital logic applications relative to the MOSFET.2 7.1.7 Deciding on a Location for the Bias Point Q For the MOSFET amplifier, the bias point Q is determined by the value of VGS and that of the load resistance RD. Two important considerations in deciding on the location of Q 2The only exception is a nonsaturating form of BJT logic circuits known as emitter-coupled logic (ECL). 382 Chapter 7 Transistor Amplifiers iD Q2 Q1 vGS VGS 0 VDD vDS Figure 7.9 Two load lines and corresponding bias points. Bias point Q1 does not leave sufficient room for positive signal swing at the drain (too close to VDD). Bias point Q2 is too close to the boundary of the triode region and might not allow for sufficient negative signal swing. are the required gain and the desired signal swing at the output. To illustrate, consider the VTC shown in Fig. 7.4(b). Here the value of RD is fixed and the only variable remaining is the value of VGS. Since the slope increases as we move closer to point B, we obtain higher gain by locating Q as close to B as possible. However, the closer Q is to the boundary point B, the smaller the allowable magnitude of negative signal swing. Thus, as often happens in engineering design, we encounter a situation requiring a trade-off. The answer here is relatively simple: For a given RD, locate Q as close to the triode region (point B) as possible to obtain high gain but sufficiently distant to allow for the required negative signal swing. In deciding on a value for RD, it is useful to refer to the iD−vDS plane. Figure 7.9 shows two load lines resulting in two extreme bias points: Point Q1 is too close to VDD, resulting in a severe constraint on the positive signal swing of vds. Exceeding the allowable positive maximum results in the positive peaks of the signal being clipped off, since the MOSFET will turn off for the part of each cycle near the positive peak. We speak of this situation by saying that the circuit does not have sufficient “headroom.” Similarly, point Q2 is too close to the boundary of the triode region, thus severely limiting the allowable negative signal swing of vds. Exceeding this limit would result in the transistor entering the triode region for part of each cycle near the negative peaks, resulting in a distorted output signal. In this situation we say that the circuit does not have sufficient “legroom.” We will have more to say on bias design in Section 7.4. Finally, we note that exactly similar considerations apply to the case of the BJT amplifier. 7.2 Small-Signal Operation and Models 383 7.2 Small-Signal Operation and Models In our study of the operation of the MOSFET and BJT amplifiers in Section 7.1 we learned that linear amplification can be obtained by biasing the transistor to operate in the active region and by keeping the input signal small. In this section, we explore the small-signal operation in greater detail. 7.2.1 The MOSFET Case Consider the conceptual amplifier circuit shown in Fig. 7.10. Here the MOS transistor is biased by applying a dc voltage3 VGS, and the input signal to be amplified, vgs, is superimposed on the dc bias voltage VGS. The output voltage is taken at the drain. The DC Bias Point The dc bias current ID can be found by setting the signal vgs to zero; thus, ID = 1 2 kn (VGS − Vt)2 = 1 2 kn VO2V (7.25) where we have neglected channel-length modulation (i.e., we have assumed λ = 0). Here VOV = VGS − Vt is the overdrive voltage at which the MOSFET is biased to operate. The dc voltage at the drain, VDS, will be VDS = VDD − RDID (7.26) To ensure saturation-region operation, we must have VDS > VOV Furthermore, since the total voltage at the drain will have a signal component superimposed on VDS, VDS has to be sufficiently greater than VOV to allow for the required negative signal swing. VDD iD RD vDS ϩ vgs ϩ Ϫ vGS VGS Ϫ Figure 7.10 Conceptual circuit utilized to study the operation of the MOSFET as a small-signal amplifier. 3Practical biasing arrangements will be studied in Section 7.4. 384 Chapter 7 Transistor Amplifiers The Signal Current in the Drain Terminal Next, consider the situation with the input signal vgs applied. The total instantaneous gate-to-source voltage will be vGS = VGS + vgs (7.27) resulting in a total instantaneous drain current iD, iD = 1 2 kn VGS + vgs − Vt 2 = 1 2 kn (VGS − Vt)2 + kn(VGS − Vt)vgs + 1 2 knv 2 gs (7.28) The first term on the right-hand side of Eq. (7.28) can be recognized as the dc bias current ID (Eq. 7.25). The second term represents a current component that is directly proportional to the input signal vgs. The third term is a current component that is proportional to the square of the input signal. This last component is undesirable because it represents nonlinear distortion. To reduce the nonlinear distortion introduced by the MOSFET, the input signal should be kept small so that 1 2 kn v 2 gs kn(VGS − Vt)vgs resulting in vgs 2(VGS − Vt) (7.29) or, equivalently, vgs 2VOV (7.30) If this small-signal condition is satisfied, we may neglect the last term in Eq. (7.28) and express iD as iD ID + id (7.31) where id = kn(VGS − Vt)vgs The parameter that relates id and vgs is the MOSFET transconductance gm, gm ≡ id v gs = kn(VGS − Vt) or in terms of the overdrive voltage VOV , gm = knVOV (7.32) (7.33) Figure 7.11 presents a graphical interpretation of the small-signal operation of the MOSFET amplifier. Note that gm is equal to the slope of the iD–vGS characteristic at the bias point, gm ≡ ∂ iD ∂ v GS vGS =VGS (7.34) This is the formal definition of gm, which can be shown to yield the expressions given in Eqs. (7.32) and (7.33). 7.2 Small-Signal Operation and Models 385 An almost-line-ar segment Q VGS 0 VOV Figure 7.11 Small-signal operation of the MOSFET amplifier. The Voltage Gain Returning to the circuit of Fig. 7.10, we can express the total instantaneous drain voltage vDS as follows: vDS = VDD − RDiD Under the small-signal condition, we have vDS = VDD − RD(ID + id ) which can be rewritten as vDS = VDS − RDid Thus the signal component of the drain voltage is vds = −id RD = −gmvgsRD (7.35) which indicates that the voltage gain is given by Av ≡ v ds v gs = −gmRD (7.36) The minus sign in Eq. (7.36) indicates that the output signal vds is 180° out of phase with respect to the input signal vgs. This is illustrated in Fig. 7.12, which shows vGS and vDS. The input signal is assumed to have a triangular waveform with an amplitude much smaller than 2(VGS – Vt), the small-signal condition in Eq. (7.29), to ensure linear operation. For operation in the saturation (active) region at all times, the minimum value of vDS should not fall below the corresponding value of vGS by more than Vt. Also, the maximum value of vDS should be 386 Chapter 7 Transistor Amplifiers vGS VGS vGS 0 vDS vDSmax ≤ VDD V V 2 2(VGS Ϫ Vt) t (gm RD )V VDS vDS min ≥ vGS max Ϫ Vt 0 t Figure 7.12 Total instantaneous voltages vGS and vDS for the circuit in Fig. 7.10. smaller than VDD; otherwise the FET will enter the cutoff region and the peaks of the output signal waveform will be clipped off. Finally, we note that by substituting for gm from Eq. (7.33) the voltage-gain expression in Eq. (7.36) becomes identical to that derived in Section 7.1—namely, Eq. (7.15). Separating the DC Analysis and the Signal Analysis From the preceding analysis, we see that under the small-signal approximation, signal quantities are superimposed on dc quantities. For instance, the total drain current iD equals the dc current ID plus the signal current id, the total drain voltage vDS = VDS + vds, and so on. It follows that the analysis and design can be greatly simplified by separating dc or bias calculations from small-signal calculations. That is, once a stable dc operating point has been established and all dc quantities calculated, we may then perform signal analysis ignoring dc quantities. Small-Signal Equivalent-Circuit Models From a signal point of view, the FET behaves as a voltage-controlled current source. It accepts a signal vgs between gate and source and provides a current gmvgs at the drain terminal. The input resistance of this controlled source is very high—ideally, infinite. The output resistance—that is, the resistance looking into the G ϩ vgs Ϫ DG ϩ gmvgs vgs Ϫ 7.2 Small-Signal Operation and Models 387 gm vgs D ro S (a) S (b) Figure 7.13 Small-signal models for the MOSFET: (a) neglecting the dependence of iD on vDS in the active region (the channel-length modulation effect) and (b) including the effect of channel-length modulation, modeled by output resistance ro = VA /ID. These models apply equally well for both NMOS and PMOS transistors. drain—also is high, and we have assumed it to be infinite thus far. Putting all of this together, we arrive at the circuit in Fig. 7.13(a), which represents the small-signal operation of the MOSFET and is thus a small-signal model or a small-signal equivalent circuit. In the analysis of a MOSFET amplifier circuit, the transistor can be replaced by the equivalent-circuit model shown in Fig. 7.13(a). The rest of the circuit remains unchanged except that ideal constant dc voltage sources are replaced by short circuits. This is a result of the fact that the voltage across an ideal constant dc voltage source does not change, and thus there will always be a zero voltage signal across a constant dc voltage source. A dual statement applies for constant dc current sources; namely, the signal current of an ideal constant dc current source will always be zero, and thus an ideal constant dc current source can be replaced by an open circuit in the small-signal equivalent circuit of the amplifier. The circuit resulting can then be used to perform any required signal analysis, such as calculating voltage gain. The most serious shortcoming of the small-signal model of Fig. 7.13(a) is that it assumes the drain current in saturation to be independent of the drain voltage. From our study of the MOSFET characteristics in saturation, we know that the drain current does in fact depend on vDS in a linear manner. Such dependence was modeled by a finite resistance ro between drain and source, whose value was given by Eq. (5.27) in Section 5.2.4, which we repeat here (with the prime on ID dropped) as ro = |VA| ID (7.37) where VA = 1/λ is a MOSFET parameter that either is specified or can be measured. It should be recalled that for a given process technology, VA is proportional to the MOSFET channel length. The current ID is the value of the dc drain current without the channel-length modulation taken into account; that is, ID = 1 2 kn VO2V (7.38) Typically, ro is in the range of 10 k to 1000 k . It follows that the accuracy of the small-signal model can be improved by including ro in parallel with the controlled source, as shown in Fig. 7.13(b). 388 Chapter 7 Transistor Amplifiers It is important to note that the small-signal model parameters gm and ro depend on the dc bias point of the MOSFET. Returning to the amplifier of Fig. 7.10, we find that replacing the MOSFET with the small-signal model of Fig. 7.13(b) results in the voltage-gain expression Av = v ds v gs = −gm(RD ro) (7.39) Thus, the finite output resistance ro results in a reduction in the magnitude of the voltage gain. Although the analysis above is performed on an NMOS transistor, the results, and the equivalent-circuit models of Fig. 7.13, apply equally well to PMOS devices, except for using |VGS|, |Vt|,|VOV |, and |VA| and replacing kn with kp. The Transconductance gm We shall now take a closer look at the MOSFET transconductance given by Eq. (7.32), which we rewrite with kn = kn(W/L) as follows: gm = kn(W/L)(VGS − Vt) = kn(W/L)VOV (7.40) This relationship indicates that gm is proportional to the process transconductance parameter kn = μnCox and to the W/L ratio of the MOS transistor; hence to obtain relatively large transconductance the device must be short and wide. We also observe that for a given device the transconductance is proportional to the overdrive voltage, VOV = VGS − Vt, the amount by which the bias voltage VGS exceeds the threshold voltage Vt. Note, however, that increasing gm by biasing the device at a larger VGS has the disadvantage of reducing the allowable voltage signal swing at the drain. Another useful expression for gm can be obtained by substituting for VOV in Eq. (7.40) by 2ID/(kn(W/L)) [from Eq. (7.25)]: √ gm = 2kn W/L ID (7.41) This expression shows two things: 1. For a given MOSFET, gm is proportional to th√e square root of the dc bias current. 2. At a given bias current, gm is proportional to W/L. In contrast, as we shall see shortly, the transconductance of the bipolar junction transistor (BJT) is proportional to the bias current and is independent of the physical size and geometry of the device. To gain some insight into the values of gm obtained in MOSFETs consider an integrated-circuit device operating at ID = 0.5 mA and having kn = 120 μA/V2. Equation (7.41) shows that for W/L = 1, gm = 0.35 mA/V, whereas a device for which W/L = 100 has gm = 3.5 mA/V. In contrast, a BJT operating at a collector current of 0.5 mA has gm = 20 mA/V. Yet another useful expression for gm of the MOSFET can be obtained by substituting for kn(W/L) in Eq. (7.40) by 2ID/(VGS – Vt)2: gm = 2ID VGS − Vt = 2I D VOV (7.42) 7.2 Small-Signal Operation and Models 389 iD ID Q Slope ϵgm = ID 1 2 VOV 0 1 2 VOV VOV Figure 7.14 The slope of the tangent at vOV the bias point Q intersects the vOV axis at 1 2 VOV . Thus, gm = ID /( 1 2 VOV ). A convenient graphical construction that clearly illustrates this relationship is shown in Fig. 7.14. In summary, there are three different relationships for determining gm—Eqs. (7.40), (7.41), and (7.42)—and there are three design parameters—(W/L), VOV , and ID, any two of which can be chosen independently. That is, the designer may choose to operate the MOSFET with a certain overdrive voltage VOV and at a particular current ID; the required W/L ratio can then be found and the resulting gm determined.4 Example 7.3 Figure 7.15(a) shows a discrete MOSFET amplifier utilizing a drain-to-gate resistance RG for biasing purposes. Such a biasing arrangement will be studied in Section 7.4. The input signal vi is coupled to the gate via a large capacitor, and the output signal at the drain is coupled to the load resistance RL via another large capacitor. We wish to analyze this amplifier circuit to determine its small-signal voltage gain, its input resistance, and the largest allowable input signal. The transistor has Vt = 1.5 V, kn (W/L) = 0.25 mA/V2, and VA = 50 V. Assume the coupling capacitors to be sufficiently large so as to act as short circuits at the signal frequencies of interest. 4This assumes that the circuit designer is also designing the device, as is typically the case in IC design. On the other hand, a circuit designer working with a discrete-circuit MOSFET obviously does not have the freedom to change its W /L ratio. Thus, in this case there are only two design parameters—VOV and ID, and only one can be specified by the designer. 390 Chapter 7 Transistor Amplifiers Example 7.3 continued VDD = (a) VDD ID RD IG = 0 RG ϩ VGS Ϫ ϩ ID VDS Ϫ (b) (c) Figure 7.15 Example 7.3: (a) amplifier circuit; (b) circuit for determining the dc operating point; (c) the amplifier small-signal equivalent circuit; (d) a simplified version of the circuit in (c). 7.2 Small-Signal Operation and Models 391 ii RG ( ) ii−gmvgs ϩ ϩ vi ϩ Ϫ vgs =vi gmvgs RЈL vo Ϫ Ϫ Rin = vi ii RЈL= RL ʈ RD ʈ ro (d) Figure 7.15 continued Solution We first determine the dc operating point. For this purpose, we eliminate the input signal vi, and open-circuit the two coupling capacitors (since they block dc currents). The result is the circuit shown in Fig. 7.14(b). We note that since IG = 0, the dc voltage drop across RG will be zero, and VGS = VDS = VDD − RDID (7.43) With VDS = VGS, the NMOS transistor will be operating in saturation. Thus, ID = 1 2 kn VGS − Vt 2 (7.44) where, for simplicity, we have neglected the effect of channel-length modulation on the dc operating point. Substituting VDD = 15 V, RD = 10 k , kn = 0.25 mA/V2, and Vt = 1.5 V in Eqs. (7.43) and (7.44), and substituting for VGS from Eq. (7.43) into Eq. (7.44) results in a quadratic equation in ID. Solving the latter and discarding the root that is not physically meaningful yields the solution ID = 1.06 mA which corresponds to VGS = VDS = 4.4 V and VOV = 4.4 − 1.5 = 2.9 V Next we proceed with the small-signal analysis of the amplifier. Toward that end we replace the MOSFET with its small-signal model to obtain the small-signal equivalent circuit of the amplifier, shown in Fig. 7.15(c). Observe that we have replaced the coupling capacitors with short circuits. The dc voltage supply VDD has also been replaced with a short circuit to ground. 392 Chapter 7 Transistor Amplifiers Example 7.3 continued The values of the transistor small-signal parameters gm and ro can be determined by using the dc bias quantities found above, as follows: gm = knVOV = 0.25 × 2.9 = 0.725 mA/V ro = VA ID = 50 1.06 = 47 k Next we use the equivalent circuit of Fig. 7.15(c) to determine the input resistance Rin ≡ vi/ii and the voltage gain Av = vo/vi. Toward that end we simplify the circuit by combining the three parallel resistances ro, RD, and RL in a single resistance RL, RL = RL||RD||ro = 10||10||47 = 4.52 k as shown in Fig. 7.15(d). For the latter circuit we can write the two equations vo = ii − gmvgs RL (7.45) and ii = vgs − vo RG (7.46) Substituting for ii from Eq. (7.46) into Eq. (7.45) results in the following expression for the voltage gain Av ≡ vo/vi = vo/vgs: Av = −gm RL 1− 1+ 1/gmRG RL /RG Since RG is very large, gmRG 1 and RL/RG 1 (the reader can easily verify this), and the gain expression can be approximated as Av −gmRL (7.47) Substituting gm = 0.725 mA/V and RL = 4.52 k yields Av = −3.3 V/V To obtain the input resistance, we substitute in Eq. (7.46) for vo = Avvgs = −gmRLvgs, then use Rin ≡ vi/ii = vgs/ii to obtain Rin = RG 1 + gmRL (7.48) 7.2 Small-Signal Operation and Models 393 This is an interesting relationship: The input resistance decreases as the gain gmRL is increased. The value of Rin can now be determined; it is 10 M Rin = 1 + 3.3 = 2.33 M which is still very large. The largest allowable input signal vˆi is constrained by the need to keep the transistor in saturation at all times; that is, vDS ≥ vGS − Vt Enforcing this condition with equality at the point vGS is maximum and vDS is minimum, we write vDS min = vGS max − Vt VDS − Av vˆi = VGS + vˆi − Vt Since VDS = VGS, we obtain vˆi = Vt Av + 1 This is a general relationship that applies to this circuit irrespective of the component values. Observe that it simply states that the maximum signal swing is determined by the fact that the bias arrangement makes VD = VG and thus, to keep the MOSFET out of the triode region, the signal between D and G is constrained to be equal to Vt. For our particular design, vˆi = 1.5 3.3 + 1 = 0.35 V Since VOV = 2.9 V, a vi of 0.35 is indeed much smaller than 2VOV = 5.8 V; thus the assumption of linear operation is justified. A modification of this circuit that increases the allowable signal swing is investigated in Problem 7.103. EXERCISE D7.4 Consider the amplifier circuit of Fig. 7.15(a) without the load resistance RL and with channel-length modulation neglected. Let VDD = 5 V, Vt = 0.7 V, and kn = 1 mA/V2. Find VOV , ID, RD, and RG to obtain a voltage gain of −25 V/V and an input resistance of 0.5 M . What is the maximum allowable input signal, vˆi? Ans. 0.319 V; 50.9 μA; 78.5 k ; 13 M ; 27 mV 394 Chapter 7 Transistor Amplifiers The T Equivalent-Circuit Model Through a simple circuit transformation it is possible to develop an alternative equivalent-circuit model for the MOSFET. The development of such a model, known as the T model, is illustrated in Fig. 7.16. Figure 7.16(a) shows the equivalent circuit studied above without ro. In Fig. 7.16(b) we have added a second gmvgs current source in series with the original controlled source. This addition obviously does not change the terminal currents and is thus allowed. The newly created circuit node, labeled X, is joined to the gate terminal G in Fig. 7.16(c). Observe that the gate current does not change—that is, it remains equal to zero—and thus this connection does not alter the terminal characteristics. We now note that we have a controlled current source gmvgs connected across its control voltage vgs. We can replace this controlled source by a resistance as long as this resistance draws an equal current as the source. (See the source-absorption theorem in Appendix D.) Thus the value of the resistance is vgs/gmvgs = 1/gm. This replacement is shown in Fig. 7.16(d), which depicts ig = 0 G ϩ vgs Ϫ id D gmvgs is S (a) ig = 0 G ϩ vgs Ϫ gmvgs X gmvgs D id is S (b) D id ig = 0 G ϩ vgs Ϫ gmvgs 1/gm is S ig = 0 G ϩ vgs Ϫ gmvgs X gmvgs D id is S (d) (c) Figure 7.16 Development of the T equivalent-circuit model for the MOSFET. For simplicity, ro has been omitted; however, it may be added between D and S in the T model of (d). 7.2 Small-Signal Operation and Models 395 D D gmvgs 1ϫi G ϩ ro G vgs 1/gm Ϫ i ro 1/gm S (a) S (b) Figure 7.17 (a) The T model of the MOSFET augmented with the drain-to-source resistance ro. (b) An alternative representation of the T model. the alternative model. Observe that ig is still zero, id = gmvgs, and is = vgs/(1/gm) = gmvgs, all the same as in the original model in Fig. 7.16(a). The model of Fig. 7.16(d) shows that the resistance between gate and source looking into the source is 1/gm. This observation and the T model prove useful in many applications. Note that the resistance between gate and source, looking into the gate, is infinite. In developing the T model we did not include ro. If desired, this can be done by incorporating in the circuit of Fig. 7.16(d) a resistance ro between drain and source, as shown in Fig. 7.17(a). An alternative representation of the T model, in which the voltage-controlled current source is replaced with a current-controlled current source, is shown in Fig. 7.17(b). Finally, we should note that in order to distinguish the model of Fig. 7.13(b) from the equivalent T model, the former is sometimes referred to as the hybrid-π model, a carryover from the bipolar transistor literature. The origin of this name will be explained shortly. Example 7.4 Figure 7.18(a) shows a MOSFET amplifier biased by a constant-current source I. Assume that the values of I and RD are such that the MOSFET operates in the saturation region. The input signal vi is coupled to the source terminal by utilizing a large capacitor CC1. Similarly, the output signal at the drain is taken through a large coupling capacitor CC2. Find the input resistance Rin and the voltage gain vo/vi. Neglect channel-length modulation. 396 Chapter 7 Transistor Amplifiers Example 7.4 continued VDD D RD 1ϫi vo G ϩ RD vo CC2 Ϫ CC1 i S 1 gm vi ϩ Ϫ I vi ϩ Ϫ Rin (a) −VSS Rin (b) Figure 7.18 (a) Amplifier circuit for Example 7.4. (b) Small-signal equivalent circuit of the amplifier in (a). Solution Replacing the MOSFET with its T equivalent-circuit model results in the amplifier equivalent circuit shown in Fig. 7.18(b). Observe that the dc current source I is replaced with an open circuit and the dc voltage source VDD is replaced by a short circuit. The large coupling capacitors have been replaced by short circuits. From the equivalent-circuit model we determine Rin = vi −i = 1/gm and vo = −iRD = vi 1/gm RD = gmRDvi Thus, Av ≡ vo vi = gm RD We note that this amplifier, known as the common-gate amplifier because the gate at ground potential is common to both the input and output ports, has a low input resistance 1/gm and a noninverting gain. We shall study this amplifier type in Section 7.3.5. 7.2 Small-Signal Operation and Models 397 EXERCISE 7.5 Use the T model of Fig. 7.17(b) to show that a MOSFET whose drain is connected to its gate exhibits an incremental resistance equal to [ 1/gm ro]. Ans. See Fig. E7.5. 0i ro Q 1 i gm (a) ( )ʈ r = 1 gm ro (b) Figure E7.5 Circuits for Exercise 7.5. Note that the bias arrangement of Q is not shown. Modeling the Body Effect As mentioned earlier (see Section 5.4), the body effect occurs in a MOSFET when the source is not tied to the substrate (which is always connected to the most negative power supply in the integrated circuit for n-channel devices and to the most positive for p-channel devices). Thus the substrate (body) will be at signal ground, but since the source is not, a signal voltage vbs develops between the body (B) and the source (S). The substrate then acts as a “second gate” or a backgate for the MOSFET. Thus the signal vbs gives rise to a drain-current component, which we shall write as gmbvbs, where gmb is the body transconductance, defined as gmb ≡ ∂ iD ∂ v BS vGS = constant vDS = constant (7.49) Recalling that iD depends on vBS through the dependence of Vt on VBS, we can show that gmb = χ gm (7.50) where χ ≡ ∂Vt = γ ∂VSB 2 2φf + VSB (7.51) Typically the value of χ lies in the range 0.1 to 0.3. Figure 7.19 shows the MOSFET model augmented to include the controlled source gmbvbs that models the body effect. Ideally, this is the model to be used whenever the source is not connected to the substrate. It has been found, however, that except in some very particular 398 Chapter 7 Transistor Amplifiers D D G B S (a) (b) Figure 7.19 Small-signal, equivalent-circuit model of a MOSFET in which the source is not connected to the body. situations, the body effect can generally be ignored in the initial, pencil-and-paper design of MOSFET amplifiers. Finally, although the analysis above was performed on an NMOS transistor, the results and the equivalent circuit of Fig. 7.19 apply equally well to PMOS transistors, except for using |VGS|, |Vt|, |VOV |, |VA|, |VSB|, |γ |, and |λ| and replacing kn with kp in the appropriate formula. EXERCISES 7.6 For the amplifier in Fig. 7.4, let VDD = 5 V, RD = 10 k , Vt = 1 V, kn = 20 μA/V2, W/L = 20, VGS = 2 V, and λ = 0. (a) Find the dc current ID and the dc voltage VDS. (b) Find gm. (c) Find the voltage gain. (d) If vgs = 0.2 sin ωt volts, find vds assuming that the small-signal approximation holds. What are the minimum and maximum values of vDS? (e) Use Eq. (7.28) to determine the various components of iD. Using the identity (sin2ωt = 1 2 − 1 2 cos 2 ωt), show that there is a slight shift in ID (by how much?) and that there is a second-harmonic component (i.e., a component with frequency 2 ω). Express the amplitude of the second-harmonic component as a percentage of the amplitude of the fundamental. (This value is known as the second-harmonic distortion.) Ans. (a) 200 μA, 3 V; (b) 0.4 mA/V; (c) –4 V/V; (d) vds = −0.8 sin ωt volts, 2.2 V, 3.8 V; (e) iD = (204 + 80 sin ωt – 4 cos 2 ωt) μA, 5% 7.7 An NMOS transistor has μnCox = 60 μA/V2, W/L = 40, Vt = 1 V, and VA = 15 V. Find gm and ro when (a) the bias voltage VGS = 1.5 V, (b) the bias current ID = 0.5 mA. Ans. (a) 1.2 mA/V, 50 k ; (b) 1.55 mA/V, 30 k 7.2 Small-Signal Operation and Models 399 7.8 A MOSFET is to operate at ID = 0.1 mA and is to have gm = 1 mA/V. If kn = 50 μA/V2, find the required W/L ratio and the overdrive voltage. Ans. 100; 0.2 V 7.9 For a fabrication process for which μp 0.4 μn, find the ratio of the width of a PMOS transistor to the width of an NMOS transistor so that the two devices have equal gm for the same bias conditions. The two devices have equal channel lengths. Ans. 2.5 7.10 A PMOS transistor has Vt = −1 V, kp = 60 μA/V2, and W/L = 16 μm/0.8 μm. Find ID and gm when the device is biased at VGS = −1.6 V. Also, find the value of ro if λ (at L = 1 μm) = –0.04 V−1. Ans. 216 μA; 0.72 mA/V; 92.6 k 7.11 Derive an expression for (gmro) in terms of VA and VOV . As we shall see in Chapter 8, this is an important transistor parameter and is known as the intrinsic gain. Evaluate the value of gmro for an NMOS transistor fabricated in a 0.8-μm CMOS process for which VA = 12.5 V/μm of channel length. Let the device have minimum channel length and be operated at an overdrive voltage of 0.2 V. Ans. gmro = 2VA/VOV ; 100 V/V 7.2.2 The BJT Case We next consider the small-signal operation of the BJT and develop small-signal equivalent-circuit models that represent its operation at a given bias point. The following development parallels what we used for the MOSFET except that here we have an added complication: The BJT draws a finite base current. As will be seen shortly, this phenomenon (finite β) manifests itself as a finite input resistance looking into the base of the BJT (as compared to the infinite input resistance looking into the gate of the MOSFET). Consider the conceptual amplifier circuit shown in Fig. 7.20(a). Here the base–emitter junction is forward biased by a dc voltage VBE. The reverse bias of the collector–base junction is established by connecting the collector to another power supply of voltage VCC through a resistor RC. The input signal to be amplified is represented by the voltage source vbe that is superimposed on VBE. The DC Bias Point We consider first the dc bias conditions by setting the signal vbe to zero. The circuit reduces to that in Fig. 7.20(b), and we can write the following relationships for the dc currents and voltages: IC = I eVBE /VT S IE = IC/α IB = IC/β VCE = VCC − IC RC (7.52) (7.53) (7.54) (7.55) For active-mode operation, VCE should be greater than (VBE −0.4) by an amount that allows for the required negative signal swing at the collector. 400 Chapter 7 Transistor Amplifiers VCC iC RC VCC IC RC iB vbe vBE VBE vCE iE IB VBE VCE IE (a) (b) Figure 7.20 (a) Conceptual circuit to illustrate the operation of the transistor as an amplifier. (b) The circuit of (a) with the signal source vbe eliminated for dc (bias) analysis. The Collector Current and the Transconductance If a signal vbe is applied as shown in Fig. 7.20(a), the total instantaneous base–emitter voltage vBE becomes vBE = VBE + vbe Correspondingly, the collector current becomes iC = IS evBE /VT = IS e(VBE +vbe )/VT = I e e VBE /VT vbe /VT S Use of Eq. (7.52) yields iC = I evbe /VT C (7.56) Now, if vbe VT , we may approximate Eq. (7.56) as iC IC 1 + vbe VT (7.57) Here we have expanded the exponential in Eq. (7.56) in a series and retained only the first two terms. That is, we have assumed that vbe VT (7.58) so that we can neglect the higher-order terms in the exponential series expansion. The condition in Eq. (7.58) is the small-signal approximation for the BJT and corresponds to that in Eq. (7.29) for the MOSFET case. The small-signal approximation for the BJT is valid only for vbe less than 5 mV or 10 mV, at most. Under this approximation, the total collector current is given by Eq. (7.57) and can be rewritten iC = IC + IC VT v be (7.59) 7.2 Small-Signal Operation and Models 401 Thus the collector current is composed of the dc bias value IC and a signal component ic, ic = IC VT v be (7.60) This equation relates the signal current in the collector to the corresponding base–emitter signal voltage. It can be rewritten as ic = gmvbe (7.61) where gm is the transconductance, and from Eq. (7.60), it is given by gm = IC VT (7.62) We observe that the transconductance of the BJT is directly proportional to the collector bias current IC. Thus to obtain a constant predictable value for gm, we need a constant predictable IC. Also, we note that BJTs have relatively high transconductance in comparison to MOSFETs: for instance, at IC = 1 mA, gm 40 mA/V. Finally, unlike the MOSFET, whose gm depends on the device dimensions (W and L), gm of a BJT depends only on the dc collector current at which it is biased to operate. A graphical interpretation for gm is given in Fig. 7.21, where it is shown that gm is equal to the slope of the tangent to the iC–vBE characteristic curve at iC = IC (i.e., at the bias point Q). Thus, gm = ∂ iC ∂ v BE iC = IC (7.63) The small-signal approximation implies keeping the signal amplitude sufficiently small that operation is restricted to an almost-linear segment of the iC–vBE exponential curve. Increasing the signal amplitude will result in the collector current having components nonlinearly related to vbe. EXERCISES 7.12 Use Eq. (7.63) to derive the expression for gm in Eq. (7.62). 7.13 Calculate the value of gm for a BJT biased at IC = 0.5 mA. Ans. 20 mA/V 402 Chapter 7 Transistor Amplifiers Q Figure 7.21 Linear operation of the transistor under the small-signal condition: A small-signal vbe with a triangular waveform is superimposed on the dc voltage VBE. It gives rise to a collector-signal current ic, also of triangular waveform, superimposed on the dc current IC. Here, ic = gmvbe, where gm is the slope of the iC–vBE curve at the bias point Q. The Base Current and the Input Resistance at the Base To determine the resistance seen by vbe, we first evaluate the total base current iB using Eq. (7.59), as follows: iB = iC β = IC β + 1 β IC VT v be Thus, iB = IB + ib where IB is equal to IC/β and the signal component ib is given by ib = 1 β IC VT v be Substituting for IC/VT by gm gives ib = gm β v be (7.64) (7.65) (7.66) 7.2 Small-Signal Operation and Models 403 The small-signal input resistance between base and emitter, looking into the base, is denoted by rπ and is defined as rπ ≡ v be ib (7.67) Using Eq. (7.66) gives rπ = β gm (7.68) Thus rπ is directly dependent on β and is inversely proportional to the bias current IC. Substituting for gm in Eq. (7.68) from Eq. (7.62) and replacing IC/β by IB gives an alternative expression for rπ , rπ = VT IB (7.69) Here, we recall that because the gate current of the MOSFET is zero (at dc and low frequencies) the input resistance at the gate is infinite; that is, in the MOSFET there is no counterpart to rπ .5 EXERCISE 7.14 A BJT amplifier is biased to operate at a constant collector current IC = 0.5 mA irrespective of the value β. If the transistor manufacturer specifies β to range from 50 to 200, give the expected range of gm, IB, and rπ . Ans. gm is constant at 20 mA/V; IB = 10 μA to 2.5 μA; rπ = 2.5 k to 10 k The Emitter Current and the Input Resistance at the Emitter The total emitter current iE can be determined using Eq. (7.59) as iE = iC α = IC α + ic α Thus, iE = IE + ie where IE is equal to IC/α and the signal current ie is given by ie = ic α = IC αVT v be = IE VT v be (7.70) (7.71) 5At high frequencies, the input capacitance at the MOSFET gate makes the input current finite (see Chapter 10). 404 Chapter 7 Transistor Amplifiers If we denote the small-signal resistance between base and emitter looking into the emitter by re, it can be defined as re ≡ v be ie (7.72) Using Eq. (7.71) we find that re, called the emitter resistance, is given by re = VT IE (7.73) Comparison with Eq. (7.62) reveals that α re = gm 1 gm (7.74) The relationship between rπ and re can be found by combining their respective definitions in Eqs. (7.67) and (7.72) as Thus, vbe = ibrπ = iere which yields rπ = (ie/ib)re rπ = (β + 1)re (7.75) Figure 7.22 illustrates the definition of rπ and re. ib vbe vbe ie r ϵ vbe ib r = (bϩ1)re re ϵ vbe ie Figure 7.22 Illustrating the definition of rπ and re. Finally, a comparison with the MOSFET would be useful: For the MOSFET, α = 1 and the resistance looking into the source is simply 1/gm. SHOCKLEY AND SILICON VALLEY: 7.2 Small-Signal Operation and Models 405 In 1956 William Bradford Shockley started a new company, Shockley Semiconductor Laboratory in Mountain View, California (near Stanford, his birthplace). While at Bell Labs, together with John Bardeen and Walter Brattain, he had invented the BJT. At Shockley, the initial concentration was on developing semiconductor devices, particularly a new four-layer diode. But Shockley’s scientific genius and ability to select and attract good team members, first demonstrated at Bell Labs, was not accompanied by comparable talent for management. Consequently, in 1957, eight of his team members (the so-called Traitorous Eight, including Gordon Moore and Robert Noyce) left Shockley to create Fairchild Semiconductor. It was a propitious time: The first Sputnik was launched a month later, and the ensuing space race accelerated demand for solid-state circuits. Decades passed, and in 2002 a group of some 30 individuals who had been associated with Silicon Valley since 1956 met at Stanford University to reminisce about Shockley’s contributions to the information technology age. They unanimously concluded that Shockley was the man who brought silicon to Silicon Valley! EXERCISE 7.15 A BJT having β = 100 is biased at a dc collector current of 1 mA. Find the value of gm, re, and rπ at the bias point. Ans. 40 mA/V; 25 ; 2.5 k The Voltage Gain The total collector voltage vCE is vCE = VCC − iC RC = VCC − (IC + ic)RC = (VCC − ICRC) − icRC = VCE − icRC (7.76) Thus, superimposed on the collector bias voltage VCE we have signal voltage vce given by vce = −icRC = −gmvbeRC = (−gmRC)vbe (7.77) from which we find the voltage gain Av of this amplifier as Av ≡ v ce v be = −gmRC (7.78) Here again we note that because gm is directly proportional to the collector bias current, the gain will be as stable as the collector bias current is made. Substituting for gm from Eq. (7.62) 406 Chapter 7 Transistor Amplifiers enables us to express the gain in the form Av = − ICRC VT (7.79) which is identical to the expression we derived in Section 7.1 (Eq. 7.21). Finally, we note that the gain expression in Eq. (7.78) is identical in form to that for the MOSFET amplifier (namely, −gmRD). EXERCISE 7.16 In the circuit of Fig. 7.20(a), VBE is adjusted to yield a dc collector current of 1 mA. Let VCC = 15 V, RC = 10 k , and β = 100. Find the voltage gain vce/vbe. If vbe = 0.005 sin ωt volt, find vC(t) and iB(t). Ans. −400 V/V; 5 – 2 sin ωt volts; 10 + 2 sin ωt μA Separating the Signal and the DC Quantities The analysis above indicates that every current and voltage in the amplifier circuit of Fig. 7.20(a) is composed of two components: a dc component and a signal component. For instance, vBE = VBE + vbe, IC = IC + ic, and so on. The dc components are determined from the dc circuit given in Fig. 7.20(b) and from the relationships imposed by the transistor (Eqs. 7.52 through 7.54). On the other hand, a representation of the signal operation of the BJT can be obtained by eliminating the dc sources, as shown in Fig. 7.23. Observe that since the voltage of an ideal dc supply does not change, the signal voltage across it will be zero. For this reason we have replaced VCC and VBE with short circuits. Had the circuit contained ideal dc current sources, these would have been replaced by open circuits. Note, however, that the circuit of Fig. 7.23 is useful only insofar as it shows the various signal currents and voltages; it is not an actual amplifier circuit, since the dc bias circuit is not shown. Figure 7.23 also shows the expressions for the current increments (ic, ib, and ie) obtained when a small signal vbe is applied. These relationships can be represented by a circuit. Such RC C ic gm vbe ib vbe /r B vce vbe vbe vbe E ie re Figure 7.23 The amplifier circuit of Fig. 7.20(a) with the dc sources (VBE and VCC) eliminated (short-circuited). Thus only the signal components are present. Note that this is a representation of the signal operation of the BJT and not an actual amplifier circuit. 7.2 Small-Signal Operation and Models 407 a circuit should have three terminals—C, B, and E—and should yield the same terminal currents indicated in Fig. 7.23. The resulting circuit is then equivalent to the transistor as far as small-signal operation is concerned, and thus it can be considered an equivalent small-signal circuit model. The Hybrid-π Model An equivalent-circuit model for the BJT is shown in Fig. 7.24(a). This model represents the BJT as a voltage-controlled current source and explicitly includes the input resistance looking into the base, rπ . The model obviously yields ic = gmvbe and ib = vbe/rπ . Not so obvious, however, is the fact that the model also yields the correct expression for ie. This can be shown as follows: At the emitter node we have ie = v be rπ + gmvbe = v be rπ 1 + gmrπ = vbe (1 + β) rπ = v be rπ 1+β = vbe/re A slightly different equivalent-circuit model can be obtained by expressing the current of the controlled source (gmvbe) in terms of the base current ib as follows: gmvbe = gm ibrπ = gmrπ ib = βib This results in the alternative equivalent-circuit model shown in Fig. 7.24(b). Here the transistor is represented as a current-controlled current source, with the control current being ib. As we have done in the case of the MOSFET’s small-signal models, we can account for the Early effect (the slight dependence of iC on vCE due to basewidth modulation) by adding the resistance ro = VA/IC between collector and emitter, as shown in Fig. 7.25. Note that to conform with the literature, we have renamed vbe as vπ . The two models of Fig. 7.25 are versions of the hybrid-π model, the most widely used model for the BJT. The equivalent circuit of Fig. 7.25(a) corresponds to that of the MOSFET [Fig. 7.13(b)] except for rπ , which accounts for the finite base current (or finite β) of the BJT. However, the equivalent circuit of Fig. 7.25(b) has no MOS counterpart. ib B + vbe rp – ic C gmvbe ib B + vbe rp – ic C bib ie E (a) gm = IC / VT rp = VT / IB = b/ gm ie E (b) Figure 7.24 Two slightly different versions of the hybrid-π model for the small-signal operation of the BJT. The equivalent circuit in (a) represents the BJT as a voltage-controlled current source (a transconductance amplifier), and that in (b) represents the BJT as a current-controlled current source (a current amplifier). 408 Chapter 7 Transistor Amplifiers Figure 7.25 The hybrid-π small-signal model, in its two versions, with the resistance ro included. It is important to note that the small-signal equivalent circuits of Fig. 7.25 model the operation of the BJT at a given bias point. This should be obvious from the fact that the model parameters gm, rπ , and ro depend on the value of the dc bias current IC, as indicated in Fig. 7.25. That is, these equivalent circuits model the incremental operation of the BJT around the bias point. As in the case of the MOSFET amplifier, including ro in the BJT model causes the voltage gain of the conceptual amplifier of Fig. 7.20(a) to become vo vbe = −gm(RC ro) (7.80) Thus, the magnitude of the gain is reduced somewhat. EXERCISE 7.17 For the model in Fig. 7.24(b) show that ic = gmvbe and ie = vbe/re. The T Model Although the hybrid-π model (in one of its two variants shown in Fig. 7.24) can be used to carry out small-signal analysis of any transistor circuit, there are situations in which an alternative model, shown in Fig. 7.26, is much more convenient. This model, called, as in the case of the MOSFET, the T model, is shown in two versions in Fig. 7.26. The model of Fig. 7.26(a) represents the BJT as a voltage-controlled current source with the control voltage being vbe. Here, however, the resistance between base and emitter, looking into the emitter, is explicitly shown. From Fig. 7.26(a) we see clearly that the model yields the correct expressions for ic and ie. It can also be shown to yield the correct expression for ib (see Exercise 7.18 on the next page). C ic e ib B vbe ie re 7.2 Small-Signal Operation and Models 409 E (a) (b) Figure 7.26 Two slightly different versions of what is known as the T model of the BJT. The circuit in (a) is a voltage-controlled current source representation and that in (b) is a current-controlled current source representation. These models explicitly show the emitter resistance re rather than the base resistance rπ featured in the hybrid-π model. If in the model of Fig. 7.26(a) the current of the controlled source is expressed in terms of the emitter current as gmvbe = gm(iere) = (gmre)ie = αie we obtain the alternative T model shown in Fig. 7.26(b). Here the BJT is represented as a current-controlled current source but with the control signal being ie. Finally, the T models can be augmented by ro to account for the dependence of ic to vce (the Early effect) to obtain the equivalent circuits shown in Fig. 7.27. EXERCISE 7.18 Show that for the T model in Fig. 7.24(a), ib = vbe/rπ . Small-Signal Models of the pnp Transistor Although the small-signal models in Figs. 7.25 and 7.27 were developed for the case of the npn transistor, they apply equally well to the pnp transistor with no change in polarities. 410 Chapter 7 Transistor Amplifiers C C gmvp gm = IC/VT B ro re = VT IE = a gm B ai ro vp re ro = VA/IC re i E E (a) (b) Figure 7.27 The T models of the BJT. Example 7.5 We wish to analyze the transistor amplifier shown in Fig. 7.28(a) to determine its voltage gain vo/vi. Assume β = 100 and neglect the Early effect. VCC 10 V ϩ10 V RBB 100 k RC 3 k VC vo 2.3 mA ϩ3 V 100 k⍀ 3 k⍀ ϩ3.1 V vi VBB 3 V 0.023 mA ϩ0.7 V 2.323 mA (a) (b) Figure 7.28 Example 7.5: (a) amplifier circuit; (b) circuit for dc analysis; (c) amplifier circuit with dc sources replaced by short circuits; (d) amplifier circuit with transistor replaced by its hybrid-π , small-signal model. 7.2 Small-Signal Operation and Models 411 RC RBB vi ϩϪ (c) Figure 7.28 continued RBB 100 k vo vi B r vbe E C gm vbe vo RC 3 k (d) Solution We shall follow a five-step process: 1. The first step in the analysis consists of determining the quiescent operating point. For this purpose we assume that vi = 0 and thus obtain the dc circuit shown in Fig. 7.28(b). The dc base current will be The dc collector current will be IB = VBB − VBE RBB 3 − 0.7 = 0.023 mA 100 IC = βIB = 100 × 0.023 = 2.3 mA The dc voltage at the collector will be VC = VCC − IC RC = +10 − 2.3 × 3 = +3.1 V Since VB at +0.7 V is less than VC, it follows that in the quiescent condition the transistor will be operating in the active mode. The dc analysis is illustrated in Fig. 7.28(b). 412 Chapter 7 Transistor Amplifiers Example 7.5 continued 2. Having determined the operating point, we can now proceed to determine the small-signal model parameters: re = VT IE = 25 mV (2.3/0.99) mA = 10.8 gm = IC VT = 2.3 mA 25 mV = 92 mA/V rπ = β gm = 100 92 = 1.09 k 3. Replacing VBB and VCC with short circuits results in the circuit in Fig. 7.28(c). 4. To carry out the small-signal analysis it is equally convenient to employ either of the two hybrid-π, equivalent-circuit models of Fig. 7.24 to replace the transistor in the circuit of Fig. 7.28(c). Using the first results in the amplifier equivalent circuit given in Fig. 7.28(d). 5. Analysis of the equivalent circuit in Fig. 7.28(d) proceeds as follows: v be = vi rπ rπ + RBB 1.09 = vi 101.09 = 0.011vi (7.81) The output voltage vo is given by vo = −gmvbeRC = −92 × 0.011vi × 3 = −3.04vi Thus the voltage gain will be Av = vo vi = −3.04 V/V (7.82) Example 7.6 To gain more insight into the operation of transistor amplifiers, we wish to consider the waveforms at various points in the circuit analyzed in the previous example. For this purpose assume that vi has a triangular waveform. First determine the maximum amplitude that vi is allowed to have. Then, with the amplitude of vi set to this value, give the waveforms of the total quantities iB(t), vBE(t), iC(t), and vC(t). 7.2 Small-Signal Operation and Models 413 Solution One constraint on signal amplitude is the small-signal approximation, which stipulates that vbe should not exceed about 10 mV. If we take the triangular waveform vbe to be 20 mV peak-to-peak and work backward, Eq. (7.81) can be used to determine the maximum possible peak of vi, vˆi = vˆbe 0.011 = 10 0.011 = 0.91 V To check whether the transistor remains in the active mode with vi having a peak value vˆi = 0.91V, we have to evaluate the collector voltage. The voltage at the collector will consist of a triangular wave vo superimposed on the dc value VC = 3.1 V. The peak voltage of the triangular waveform will be vˆo = vˆi × gain = 0.91 × 3.04 = 2.77 V It follows that when the output swings negative, the collector voltage reaches a minimum of 3.1 − 2.77 = 0.33 V, which is lower than the base voltage by less than 0.4 V. Thus the transistor will remain in the active mode with vi having a peak value of 0.91 V. Nevertheless, to be on the safe side, we will use a somewhat lower value for vˆi of approximately 0.8 V, as shown in Fig. 7.29(a), and complete the analysis of this problem utilizing the equivalent circuit in Fig. 7.28(d). The signal current in the base will be triangular, with a peak value iˆb of iˆb = vˆi RBB + rπ = 0.8 100 + 1.09 = 0.008 mA This triangular-wave current will be superimposed on the quiescent base current IB, as shown in Fig. 7.29(b). The base–emitter voltage will consist of a triangular-wave component superimposed on the dc VBE that is approximately 0.7 V. The peak value of the triangular waveform will be vˆbe = vˆi rπ rπ + RBB = 0.8 1.09 100 + 1.09 = 8.6 mV The total vBE is sketched in Fig. 7.29(c). The signal current in the collector will be triangular in waveform, with a peak value iˆc given by iˆc = βiˆb = 100 × 0.008 = 0.8 mA This current will be superimposed on the quiescent collector current IC (= 2.3 mA), as shown in Fig. 7.29(d). The signal voltage at the collector can be obtained by multiplying vi by the voltage gain; that is, vˆo = 3.04 × 0.8 = 2.43 V Figure 7.29(e) shows a sketch of the total collector voltage vC versus time. Note the phase reversal between the input signal vi and the output signal vo. Finally, we observe that each of the total quantities is the sum of a dc quantity (found from the dc circuit in Fig. 7.28b), and a signal quantity (found from the circuit in Fig. 7.28d). 414 Chapter 7 Transistor Amplifiers Example 7.6 continued vi 0.8 V vˆi 0 t 0.8 V iB (mA) 0.03 0.02 0.01 0 vBE (a) iˆb 0.008 mA ib iB (b) IB 0.023 mA t vbe 0.7 V vˆbe 8.6 mV vBE VBE 0 t (c) iC (mA) IC 2.3 mA ic 3 2 1 iˆc 0.8 mA iC 0 IC t (d) Figure 7.29 Signal waveforms in the circuit of Fig. 7.28. 7.2 Small-Signal Operation and Models 415 vC (V) 6 4 2 0 Figure 7.29 continued vˆo 2.43 V vo vo VC 3.1 V vC t 0.67 V (e) Example 7.7 We need to analyze the circuit of Fig. 7.30(a) to determine the voltage gain and the signal waveforms at various points. The capacitor CC1 is a coupling capacitor whose purpose is to couple the signal vi to the emitter while blocking dc. In this way the dc bias established by V + and V − together with RE and RC will V 10 V RE 10 k 10 V 0.93 mA 10 k CC1 vi 0.7 V CC2 vo 0.92 mA 5.4 V RC 5 k 5k V 10 V 10 V (a) (b) Figure 7.30 Example 7.7: (a) circuit; (b) dc analysis; (c) circuit with the dc sources eliminated; (d) small-signal analysis using the T model for the BJT. 416 Chapter 7 Transistor Amplifiers Example 7.7 continued RE E vi RE ie re ie vi re vo ie RC B RC re vi vi ie C vo vo RC RC (c ) (d) Figure 7.30 continued not be disturbed when the signal vi is connected. For the purpose of this example, CC1 will be assumed to be very large so as to act as a perfect short circuit at signal frequencies of interest. Similarly, another very large capacitor CC2 is used to couple the output signal vo to other parts of the system. You may neglect the Early effect. Solution Here again we shall follow a five-step process: 1. Figure 7.30(b) shows the circuit with the signal source and the coupling capacitors eliminated. The dc operating point can be determined as follows: IE = +10 − VE RE +10 − 0.7 = 0.93 mA 10 Assuming β = 100, then α = 0.99, and IC = 0.99IE = 0.92 mA VC = −10 + ICRC = −10 + 0.92 × 5 = −5.4 V Thus the transistor is in the active mode. 7.2 Small-Signal Operation and Models 417 2. We now determine the small-signal parameters as follows: gm = IC VT = 0.92 0.025 = 36.8 mA/V re = VT IE = 0.025 0.92 = 27.2 β = 100 α = 0.99 rπ = β gm = 100 36.8 = 2.72 k 3. To prepare the circuit for small-signal analysis, we replace the dc sources with short circuits. The resulting circuit is shown in Fig. 7.30(c). Observe that we have also eliminated the two coupling capacitors, since they are assumed to be acting as perfect short circuits. 4. We are now ready to replace the BJT with one of the four equivalent-circuit models of Figs. 7.24 and 7.26. Although any of the four will work, the T models of Fig. 7.26 will be more convenient because the base is grounded. Selecting the version in Fig. 7.26(b) results in the amplifier equivalent circuit shown in Fig. 7.30(d). 5. Analysis of the circuit in Fig. 7.30(d) to determine the output voltage vo and hence the voltage gain vo/vi is straightforward and is given in the figure. The result is Av = vo vi = αRC re = 0.99 × 5 0.0272 = 182 V/V Note that the voltage gain is positive, indicating that the output is in phase with the input signal. This property is due to the fact that the input signal is applied to the emitter rather than to the base, as was done in Example 7.5. We should emphasize that the positive gain has nothing to do with the fact that the transistor used in this example is of the pnp type. Returning to the question of allowable signal magnitude, we observe from Fig. 7.30(d) that veb = vi. Thus, if small-signal operation is desired (for linearity), then the peak of vi should be limited to approximately 10 mV. With Vˆi set to this value, as shown for a sine-wave input in Fig. 7.31, the peak amplitude at the collector, Vˆo, will be Vˆo = 182 × 0.01 = 1.82 V 418 Chapter 7 Transistor Amplifiers Example 7.7 continued 1.82 Figure 7.31 Input and output waveforms for the circuit of Fig. 7.30. Observe that this amplifier is noninverting, a property of the grounded-base configuration. EXERCISE 7.19 To increase the voltage gain of the amplifier analyzed in Example 7.7, the collector resistance RC is increased to 7.5 k . Find the new values of VC, Av, and the peak amplitude of the output sine wave corresponding to an input sine wave vi of 10-mV peak. Ans. –3.1 V; 276 V/V; 2.76 V Performing Small-Signal Analysis Directly on the Circuit Diagram In most cases one should explicitly replace each BJT with its small-signal model and analyze the resulting circuit, as we have done in the examples above. This systematic procedure is particularly recommended for beginning students. Experienced circuit designers, however, often perform a first-order analysis directly on the circuit. Figure 7.32 illustrates this process for the two 7.2 Small-Signal Operation and Models 419 circuits we analyzed in Examples 7.5 and 7.7. The reader is urged to follow this direct analysis procedure (the steps are numbered). Observe that the equivalent-circuit model is implicitly utilized; we are only saving the step of drawing the circuit with the BJT replaced by its model. Direct analysis, however, has an additional very important benefit: It provides insight regarding the signal transmission through the circuit. Such insight can prove invaluable in design, particularly at the stage of selecting a circuit configuration appropriate for a given application. Direct analysis can be utilized also for MOS amplifier circuits. ic RC 3 ib RBB vi ϩϪ 4 bRC vi r 1 vi 2 5 Av ϵ vo vi b RC (a) vi ϩ Ϫ 3 RE 1 ie re 2 4 vi re re vi re vi RC 5 Av ϵ vo vi re (b) Figure 7.32 Performing signal analysis directly on the circuit diagram with the BJT small-signal model implicitly employed: (a) circuit for Example 7.5; (b) circuit for Example 7.7. 420 Chapter 7 Transistor Amplifiers EXERCISE 7.20 The transistor in Fig. E7.20 is biased with a constant current source I = 1 mA and has β = 100 and VA = 100 V. (a) Neglecting the Early effect, find the dc voltages at the base, emitter, and collector. (b) Find gm, rπ , and ro. (c) If terminal Z is connected to ground, X to a signal source vsig with a source resistance Rsig = 2 k , and Y to an 8-k load resistance, use the hybrid-π model shown earlier (Fig. 7.25) to draw the small-signal equivalent circuit of the amplifier. (Note that the current source I should be replaced with an open circuit.) Calculate the overall voltage gain vy/vsig. If ro is neglected, what is the error in estimating the gain magnitude? (Note: An infinite capacitance is used to indicate that the capacitance is sufficiently large that it acts as a short circuit at all signal frequencies of interest. However, the capacitor still blocks dc.) ϱ ϱ ϱ Figure E7.20 Ans. (a) –0.1 V, –0.8 V, +2.1 V; (b) 40 mA/V, 2.5 k , 100 k ; (c) –77 V/V, +3.9% 7.2.3 Summary Tables We conclude this section by presenting three useful summary tables: Table 7.1 lists the five steps to be followed in the analysis of a MOSFET or a BJT amplifier circuit. Table 7.2 presents the MOSFET small-signal, equivalent-circuit models, together with the formulas for calculating the parameter values of the models. Finally, Table 7.3 supplies the corresponding data for the BJT. 7.2 Small-Signal Operation and Models 421 Table 7.1 Systematic Procedure for the Analysis of Transistor Amplifier Circuits 1. Eliminate the signal source and determine the dc operating point of the transistor. 2. Calculate the values of the parameters of the small-signal model. 3. Eliminate the dc sources by replacing each dc voltage source by a short circuit and each dc current source by an open circuit. 4. Replace the transistor with one of its small-signal, equivalent-circuit models. Although any of the models can be used, one might be more convenient than the others for the particular circuit being analyzed. This point will be made clearer in the next section. 5. Analyze the resulting circuit to determine the required quantities (e.g., voltage gain, input resistance). Table 7.2 Small-Signal Models of the MOSFET Small-Signal Parameters NMOS transistors Transconductance: g =μC W L V = 2μ C W L I = 2I V Output resistance: r = V /I = 1/λI PMOS transistors Same formulas as for NMOS except using |V |, V , |λ| and replacing μ with μ . Small-Signal , Equivalent-Circuit Models D D G gv v i gv DG r G r r v 1 g i1 g S Hybrid-π model S S T models 422 Chapter 7 Transistor Amplifiers Table 7.3 Small-Signal Models of the BJT Hybrid-π Model (gmvπ ) Version B C vr r gv (βib) Version i B r C r i E T Model (gmvπ ) Version C E (αi) Version C gmv B ro v re i B ro re i E Model Parameters in Terms of DC Bias Currents gm = IC VT re = VT IE = α VT IC In Terms of gm re = α gm rπ = β gm In Terms of re α gm = re rπ = (β + 1)re Relationships between α and β β = 1 α − α α = β β + 1 E rπ = VT IB = β VT IC gm + 1 rπ = 1 re β + 1 = 1 1 − α ro = |VA| IC 7.3 Basic Configurations 423 7.3 Basic Configurations It is useful at this point to take stock of where we are and where we are going in our study of transistor amplifiers. In Section 7.1 we examined the underlying principle for the application of the MOSFET, and of the BJT, as an amplifier. There we found that almost-linear amplification can be obtained by dc biasing the transistor at an appropriate point in its active region of operation, and by keeping the input signal (vgs or vbe) small. We then developed, in Section 7.2, circuit models that represent the small-signal operation of each of the two transistor types (Tables 7.2 and 7.3), thus providing a systematic procedure (Table 7.1) for the analysis of transistor amplifiers. We are now ready to consider the various possible configurations of MOSFET and BJT amplifiers, and we will do that in the present section. To focus our attention on the salient features of the various configurations, we shall present them in their most simple, or “stripped-down,” version. Thus, we will not show the dc biasing arrangements, leaving the study of bias design to the next section. Finally, in Section 7.5 we will bring everything together and present practical discrete-circuit amplifiers, namely, amplifier circuits that can be constructed using discrete components. The study of integrated-circuit amplifiers begins in Chapter 8. 7.3.1 The Three Basic Configurations There are three basic configurations for connecting a MOSFET or a BJT as an amplifier. Each of these configurations is obtained by connecting one of the device terminals to ground, thus creating a two-port network with the grounded terminal being common to the input and output ports. The resulting configurations are shown in Fig. 7.33(a–c) for the MOSFET and in Fig. 7.33(d–f) for the BJT. In the circuit of Fig. 7.33(a) the source terminal is connected to ground, the input voltage signal vi is applied between the gate and ground, and the output voltage signal vo is taken between the drain and ground, across the resistance RD. This configuration, therefore, is called the grounded-source or common-source (CS) amplifier. It is by far the most popular MOS amplifier configuration, and we utilized it in Sections 7.1 and 7.2 to study MOS amplifier operation. A parallel set of remarks apply to the BJT counterpart, the grounded-emitter or common-emitter (CE) amplifier in Fig. 7.33(d). The common-gate (CG) or grounded-gate amplifier is shown in Fig. 7.33(b), and its BJT counterpart, the common-base (CB) or grounded-base amplifier in Fig. 7.33(e). Here the gate (base) is grounded, the input signal vi is applied to the source (emitter), and the output signal vo is taken at the drain (collector) across the resistance RD (RC). We encountered a CG amplifier in Example 7.4 and a CB amplifier in Example 7.7. Finally, Fig. 7.33(c) shows the common drain (CD) or grounded-drain amplifier, and Fig. 7.31(f) shows its BJT counterpart, the common-collector (CC) or grounded collector amplifier. Here the drain (collector) terminal is grounded, the input signal vi is applied between gate (base) and ground, and the output voltage vo is taken between the source (emitter) and ground, across a resistance RL. For reasons that will become apparent shortly, this pair of configurations is more commonly called the source follower and the emitter follower. Our study of the three basic amplifier configurations of the MOSFET and of the BJT will reveal that each has distinctly different attributes, hence areas of application. As well, it will be shown that although each pair of configurations, (e.g., CS and CE), has many common attributes, important differences remain. 424 Chapter 7 Transistor Amplifiers vi ϩ Ϫ ϩ RD vo Ϫ (a) Common Source (CS) vi Ϫϩ ϩ RD vo Ϫ (b) Common Gate (CG) vi ϩ Ϫ ϩ RL vo Ϫ (c) Common Drain (CD) or Source Follower vi ϩϪ RC vo vi ϩϪ RC vo vi ϩϪ RL vo (d) Common-Emitter (CE) (e) Common-Base (CB) (f) Common-Collector (CC) or Emitter Follower Figure 7.33 The basic configurations of transistor amplifiers. (a)–(c) For the MOSFET; (d)–(f) for the BJT. Our next step is to replace the transistor in each of the six circuits in Fig. 7.33 by an appropriate equivalent-circuit model (from Tables 7.2 and 7.3) and analyze the resulting circuits to determine important characteristic parameters of the particular amplifier configuration. To simplify matters, we shall not include ro in the initial analysis. At the end of the section we will offer a number of comments about when to include ro in the analysis, and on the expected magnitude of its effect. 7.3.2 Characterizing Amplifiers Before we begin our study of the different transistor amplifier configurations, we consider how to characterize the performance of an amplifier as a circuit building block. An introduction to this topic was presented in Section 1.5. Figure 7.34(a) shows an amplifier fed with a signal source having an open-circuit voltage vsig and an internal resistance Rsig. These can be the parameters of an actual signal source or, in a cascade amplifier, the The´venin equivalent of the output circuit of another amplifier stage preceding the one under study. The amplifier is shown with a load resistance RL connected to the output terminal. Here, RL can be an actual load resistance or the input resistance of a succeeding amplifier stage in a cascade amplifier. Figure 7.34(b) shows the amplifier circuit with the amplifier block replaced by its equivalent-circuit model. The input resistance Rin represents the loading effect of the amplifier Rsig vsig ϩ Ϫ ii ϩ vi Ϫ io ϩ RL vo Ϫ (a) Rsig ii Ro io vsig ϩ Ϫ ϩ vi Rin Ϫ ϩ Ϫ Avovi ϩ RL vo Ϫ (b) 7.3 Basic Configurations 425 ϩ vi = 0 Ϫ ϩ vx RL ix Ϫ Ro (c) Figure 7.34 Characterization of the amplifier as a functional block: (a) An amplifier fed with a voltage signal vsig having a source resistance Rsig, and feeding a load resistance RL; (b) equivalent-circuit representation of the circuit in (a); (c) determining the amplifier output resistance Ro. input on the signal source. It is found from Rin ≡ vi ii and together with the resistance Rsig forms a voltage divider that reduces vsig to the value vi that appears at the amplifier input, vi = Rin Rin + Rsig v sig (7.83) Most of the amplifier circuits studied in this section are unilateral. That is, they do not contain internal feedback, and thus Rin will be independent of RL. However, in general Rin may depend on the load resistance RL. Indeed one of the six configurations studied in this section, the emitter follower, exhibits such dependence. The second parameter in characterizing amplifier performance is the open-circuit voltage gain Av o, defined as Av o ≡ vo vi RL = ∞ 426 Chapter 7 Transistor Amplifiers The third and final parameter is the output resistance Ro. Observe from Fig. 7.34(b) that Ro is the resistance seen looking back into the amplifier output terminal with vi set to zero. Thus Ro can be determined, at least conceptually, as indicated in Fig. 7.34(c) with Ro = vx ix Because Ro is determined with vi = 0, the value of Ro does not depend on Rsig. The controlled source Av ovi and the output resistance Ro represent the The´venin equivalent of the amplifier output circuit, and the output voltage vo can be found from vo = RL RL + Ro Av ovi (7.84) Thus the voltage gain of the amplifier proper, Av, can be found as Av ≡ vo vi = Av o RL RL + Ro and the overall voltage gain, Gv, Gv ≡ vo v sig (7.85) can be determined by combining Eqs. (7.83) and (7.85): Gv = Rin Rin + Rsig Av o RL RL + Ro (7.86) 7.3.3 The Common-Source (CS) and Common-Emitter (CE) Amplifiers Of the three basic transistor amplifier configurations, the common-source (common-emitter, for BJT), is the most widely used. Typically, in an amplifier formed by cascading a number of gain stages, the bulk of the voltage gain is obtained by using one or more common-source (or common-emitter, for BJT) stages in cascade. Characteristic Parameters of the CS Amplifier Figure 7.35(a) shows a common-source amplifier (with the biasing arrangement omitted) fed with a signal source vsig having a source resistance Rsig. We wish to analyze this circuit to determine Rin, Av o, and Ro. For this purpose, we assume that RD is part of the amplifier; thus if a load resistance RL is connected to the amplifier output, RL appears in parallel with RD. In such a case, we wish to determine Av and Gv as well. Replacing the MOSFET with its hybrid-π model (without ro), we obtain the CS amplifier equivalent circuit in Fig. 7.35(b) for which, tracing the signal from input to output, we can write by inspection Rin = ∞ vi = vsig vgs = vi vo = −gmvgsRD (7.87) 7.3 Basic Configurations 427 vsig ϩ Ϫ Rsig ϩ vi Ϫ Rin RD (a) ϩ vo Ϫ Ro Rsig vsig ϩϪ ϩ vgs ϭ vi gmvgs Ϫ ϩ RD vo Ϫ Rin = ϱ Ro = RD (b) Figure 7.35 (a) Common-source amplifier fed with a signal vsig from a generator with a resistance Rsig. The bias circuit is omitted. (b) The common-source amplifier with the MOSFET replaced with its hybrid-π model. Thus, Av o ≡ vo vi = −gmRD Ro = RD (7.88) (7.89) If a load resistance RL is connected across RD, the voltage gain Av can be obtained from Av = Av o RL RL + Ro (7.90) where Av o is given by Eq. (7.88) and Ro by Eq. (7.89), or alternatively by simply adding RL in parallel with RD in Eq. (7.88), thus Av = −gm(RD RL) (7.91) The reader can easily show that the expression obtained from Eq. (7.90) is identical to that in Eq. (7.91). Finally, since Rin = ∞ and thus vi = vsig, the overall voltage gain Gv is equal to Av , Gv ≡ vo vsig = −gm(RD RL ) (7.92) 428 Chapter 7 Transistor Amplifiers EXERCISE 7.21 A CS amplifier utilizes a MOSFET biased at ID = 0.25 mA with VOV = 0.25 V and RD = 20 k . The amplifier is fed with a signal source having Rsig = 100 k , and a 20-k load is connected to the output. Find Rin, Av o, Ro, Av , and Gv . If, to maintain reasonable linearity, the peak of the input sine-wave signal is limited to 10% of 2VOV , what is the peak of the sine-wave voltage at the output? Ans. ∞; −40 V/V; 20 k ; −20 V/V; −20 V/V; 1 V Characteristic Parameters of the CE Amplifier Figure 7.36(a) shows a common-emitter amplifier. Its equivalent circuit, obtained by replacing the BJT with its hybrid-π model (without ro), is shown in Fig. 7.36(b). The latter circuit can be analyzed to obtain the characteristic parameters of the CE amplifier. The analysis parallels that for the MOSFET above except that here we have the added complexity of a finite input resistance rπ . Tracing the signal through the amplifier from input to output, we can write by inspection Rin = rπ Rsig vsig ϩϪ vi RC vo Rin Ro (a) Rsig vsig ϩ Ϫ r RC vo (b) Figure 7.36 (a) Common-emitter amplifier fed with a signal vsig from a generator with a resistance Rsig. The bias circuit is omitted. (b) The common-emitter amplifier circuit with the BJT replaced by its hybrid-π model. 7.3 Basic Configurations 429 Then we write Thus, vi = rπ rπ + Rsig vsig vπ = vi vo = −gmvπ RC Av o ≡ vo vi = −gmRC (7.93) (7.94) Ro = RC (7.95) With a load resistance RL connected across RC, Av = −gm(RC RL) (7.96) and the overall voltage gain Gv can be found from Gv ≡ vo vsig = vi vsig vo vi Thus, Gv = − rπ rπ + Rsig gm (RC RL ) (7.97) It is important to note here the effect of the finite input resistance (rπ ) in reducing the magnitude of the voltage gain by the voltage-divider ratio rπ /(rπ + Rsig). The extent of the gain reduction depends on the relative values of rπ and Rsig. However, there is a compensating effect in the CE amplifier: gm of the BJT is usually much higher than the corresponding value of the MOSFET. Example 7.8 A CE amplifier utilizes a BJT with β = 100 is biased at IC = 1 mA and has a collector resistance RC = 5 k . Find Rin, Ro, and Av o. If the amplifier is fed with a signal source having a resistance of 5 k , and a load resistance RL = 5 k is connected to the output terminal, find the resulting Av and Gv. If vˆπ is to be limited to 5 mV, what are the corresponding vˆsig and vˆo with the load connected? Solution At IC = 1 mA, gm = IC VT = 1 mA 0.025 V = 40 mA/V β 100 rπ = gm = 40 mA/V = 2.5 k 430 Chapter 7 Transistor Amplifiers Example 7.8 continued The amplifier characteristic parameters can now be found as Rin = rπ = 2.5 k Av o = −gmRC = −40 mA/V × 5 k = −200 V/V Ro = RC = 5 k With a load resistance RL = 5 k connected at the output, we can find Av by either of the following two approaches: Av = Av o RL RL + Ro = −200 × 5 5+5 = −100 V/V or Av = −gm(RC RL) = −40(5 5) = −100 V/V The overall voltage gain Gv can now be determined as Gv = Rin Rin + Rsig Av = 2.5 2.5 + 5 × −100 = −33.3 V/V If the maximum amplitude of vπ is to be 5 mV, the corresponding value of vˆsig will be vˆsig = Rin + Rsig Rin 2.5 + 5 vˆπ = 2.5 × 5 = 15 mV and the amplitude of the signal at the output will be vˆo = Gvvˆsig = 33.3 × 0.015 = 0.5 V 7.3 Basic Configurations 431 EXERCISE 7.22 The designer of the amplifier in Example 7.8 decides to lower the bias current to half its original value in order to raise the input resistance and hence increase the fraction of vsig that appears at the input of the amplifier proper. In an attempt to maintain the voltage gain, the designer decides to double the value of RC. For the new design, determine Rin, Av o, Ro, Av, and Gv. If the peak amplitude of vπ is to be limited to 5 mV, what are the corresponding values of vˆsig and vˆo (with the load connected)? Ans. 5 k ; −200 V/V; 10 k ; −66.7 V/V; −33.3 V/V; 10 mV; 0.33 V Comment: Although a larger fraction of the input signal reaches the amplifier input, linearity considerations cause the output signal to be in fact smaller than in the original design! Final Remarks 1. The CS and CE amplifiers are the most useful of all transistor amplifier configurations. They exhibit a moderate to high input resistance (infinite for the CS), a moderate to high output resistance, and reasonably high voltage gain. 2. The input resistance of the CE amplifier, Rin = rπ = β/gm, is inversely proportional to the dc bias current IC. To increase Rin one is tempted to lower the bias current IC; however, this also lowers gm and hence the voltage gain. This is a significant design trade-off. If a much higher input resistance is desired, then a modification of the CE configuration (to be discussed in Section 7.3.4) can be applied, or an emitter-follower stage can be inserted between the signal source and the CE amplifier (see Section 7.3.6). 3. Reducing RD or RC to lower the output resistance of the CS or CE amplifier, respectively, is usually not a viable proposition because the voltage gain is also reduced. Alternatively, if a very low output resistance (in the ohms or tens-of-ohms range) is needed, a source-follower or an emitter-follower stage can be utilized between the output of the CS or CE amplifier and the load resistance (see Section 7.3.6). 4. Although the CS and the CE configurations are the workhorses of transistor amplifiers, both suffer from a limitation on their high-frequency response. As will be shown in Chapter 10, combining the CS (CE) amplifier with a CG (CB) amplifier can extend the bandwidth considerably. The CG and CB amplifiers are studied in Section 7.3.5. 7.3.4 The Common-Source (Common-Emitter) Amplifier with a Source (Emitter) Resistance It is often beneficial to insert a resistance Rs (a resistance Re) in the source lead (the emitter lead) of a common-source (common-emitter) amplifier. Figure 7.37(a) shows a CS amplifier with a resistance Rs in its source lead. The corresponding small-signal equivalent circuit is shown 432 Chapter 7 Transistor Amplifiers Figure 7.37 The CS amplifier with a source resistance Rs: (a) circuit without bias details; (b) equivalent circuit with the MOSFET represented by its T model. in Fig. 7.37(b), where we have utilized the T model for the MOSFET. The T model is used in preference to the hybrid-π model because it makes the analysis in this case considerably simpler. In general, whenever a resistance is connected in the source lead, the T model is preferred. The source resistance then simply appears in series with the model resistance 1/gm and can be added to it. From Fig. 7.37(b) we see that as expected, the input resistance Rin is infinite and thus vi = vsig. Unlike the CS amplifier, however, here only a fraction of vi appears between gate and source as vgs. The voltage divider composed of 1/gm and Rs, which appears across the amplifier input, can be used to determine vgs, as follows: v gs = v i 1/gm 1/gm + Rs = vi 1 + gmRs (7.98) Thus we can use the value of Rs to control the magnitude of the signal vgs and thereby ensure that vgs does not become too large and cause unacceptably high nonlinear distortion. This is the first benefit of including resistor Rs. Other benefits will be encountered in later sections and chapters. For instance, it will be shown in Chapter 10 that Rs causes the useful bandwidth of the amplifier to be extended. The mechanism by which Rs causes such improvements in amplifier performance is negative feedback. To see how Rs introduces negative feedback, refer to Fig. 7.37(a): If with vsig and hence vi kept constant, the drain current increases for some 7.3 Basic Configurations 433 reason, the source current also will increase, resulting in an increased voltage drop across Rs. Thus the source voltage rises, and the gate-to-source voltage decreases. The latter effect causes the drain current to decrease, counteracting the initially assumed change, an indication of the presence of negative feedback. In Chapter 11 we shall study negative feedback formally. There we will learn that the improvements that negative feedback provides are obtained at the expense of a reduction in gain. We will now show this to be the case in the circuit of Fig. 7.37. The output voltage vo is obtained by multiplying the controlled-source current i by RD, vo = −iRD The current i in the source lead can be found by dividing vi by the total resistance in the source, i = vi = 1/gm + Rs gm 1 + gmRs vi (7.99) Thus, the voltage gain Av o can be found as Av o ≡ vo vi = − gmRD 1 + gmRs which can also be expressed as (7.100) Av o = − RD 1/gm + Rs (7.101) Equation (7.100) indicates that including the resistance Rs reduces the voltage gain by the factor (1 + gmRs). This is the price paid for the improvements that accrue as a result of Rs. It is interesting to note that in Chapter 11, we will find that the factor (1 + gmRs) is the “amount of negative feedback” introduced by Rs. It is also the same factor by which linearity, bandwidth, and other performance parameters improve. Because of the negative-feedback action of Rs it is known as a source-degeneration resistance. There is another useful interpretation of the expression for the drain current in Eq. (7.99): The quantity between brackets on the right-hand side can be thought of as the “effective transconductance with Rs included.” Thus, including Rs reduces the transconductance by the factor (1 + gmRs). This, of course, is simply the result of the fact that only a fraction 1/(1 + gmRs) of vi appears as vgs (see Eq. 7.98). The alternative gain expression in Eq. (7.101) has a powerful and insightful interpretation: The voltage gain between gate and drain is equal to the ratio of the total resistance in the drain (RD) to the total resistance in the source (1/gm + Rs), Voltage gain from gate to drain = − Total resistance in drain Total resistance in source (7.102) This is a general expression. For instance, setting Rs = 0 in Eq. (7.101) yields Avo of the CS amplifier. 434 Chapter 7 Transistor Amplifiers Finally, we consider the situation of a load resistance RL connected at the output. We can obtain the gain Av using the open-circuit voltage gain Avo together with the output resistance Ro, which can be found by inspection to be Ro = RD Alternatively, Av can be obtained by simply replacing RD in Eq. (7.101) or (7.100) by (RD RL); thus, Av = − gm(RD RL) 1 + gmRs (7.103) or Av = − RD 1/gm RL + Rs (7.104) Observe that Eq. (7.104) is a direct application of the ratio of total resistance rule of Eq. (7.102). Finally, note that because Rin is infinite, vi = vsig and the overall voltage gain Gv is equal to Av. EXERCISE 7.23 In Exercise 7.21 we applied an input signal vsig of 50 mV peak and obtained an output signal of approximately 1 V peak. Assume that for some reason we now have an input signal vsig that is 0.2 V peak and that we wish to modify the circuit to keep vgs unchanged, and thus keep the nonlinear distortion from increasing. What value should we use for Rs? What value of Gv will result? What will the peak signal at the output become? Assume ro = ∞. Ans. 1.5 k ; −5 V/V; 1 V We next turn our attention to the BJT case. Figure 7.38(a) shows a CE amplifier with a resistance Re in its emitter. The corresponding equivalent circuit, utilizing the T model, is shown in Fig. 7.38(b). Note that in the BJT case also, as a general rule, the T model results in a simpler analysis and should be employed whenever there is a resistance in series with the emitter. To determine the amplifier input resistance Rin, we note from Fig. 7.38(b) that Rin ≡ vi ib where ib = (1 − α)ie = β ie + 1 (7.105) and ie = re vi + Re (7.106) 7.3 Basic Configurations 435 Rsig RC vsig vi Re Rin C ic vo Ro ie Rsig B ib RC vo ie re Ro vsig vi E Re Rin (b) Figure 7.38 The CE amplifier with an emitter resistance Re; (a) circuit without bias details; (b) equivalent circuit with the BJT replaced with its T model. Thus, Rin = (β + 1)(re + Re) (7.107) This is a very important result. It states that the input resistance looking into the base is (β + 1) times the total resistance in the emitter, and is known as the resistance-reflection rule. The factor (β + 1) arises because the base current is 1/(β + 1) times the emitter current. The expression for Rin in Eq. (7.107) shows clearly that including a resistance Re in the emitter can substantially increase Rin, a very desirable result. Indeed, the value of Rin is increased by the ratio Rin(with Re included) = (β + 1)(re + Re) Rin(without Re) (β + 1)re = 1 + Re re 1 + gmRe (7.108) Thus the circuit designer can use the value of Re to control the value of Rin. 436 Chapter 7 Transistor Amplifiers To determine the voltage gain Avo, we see from Fig. 7.38(b) that vo = −icRC = −αieRC Substituting for ie from Eq. (7.106) gives Av o = −α re RC + Re (7.109) This is a very useful result: It states that the gain from base to collector is α times the ratio of the total resistance in the collector to the total resistance in the emitter (in this case, re + Re), Voltage gain from base to collector = −α Total resistance in collector Total resistance in emitter (7.110) This is the BJT version of the MOSFET expression in Eq. (7.102) except that here we have the additional factor α. This factor arises because ic = αie, unlike the MOSFET case where id = is. Usually, α 1 and can be dropped from Eq. (7.110). The open-circuit voltage gain in Eq. (7.109) can be expressed alternatively as Av o = −α re 1 RC + Re/re Thus, Av o = − 1 gm RC + Re/re − gmRC 1 + gmRe (7.111) Thus, including Re reduces the voltage gain by the factor (1 + gmRe), which is the same factor by which Rin is increased. This points out an interesting trade-off between gain and input resistance, a trade-off that the designer can exercise through the choice of an appropriate value for Re. The output resistance Ro can be found from the circuit in Fig. 7.38(b) by inspection: Ro = RC If a load resistance RL is connected at the amplifier output, Av can be found as Av = Av o RL RL + Ro = −α RC RL re + Re RL + RC = −α RC RL re + Re (7.112) which could have been written directly using Eq. (7.110). The overall voltage gain Gv can now be found: Gv = Rin Rin + Rsig × −α RC RL re + Re 7.3 Basic Configurations 437 Substituting for Rin from Eq. (7.107) and replacing α with β/(β + 1) results in Gv = −β Rsig + RC (β + RL 1)(re + Re) (7.113) Careful examination of this expression reveals that the denominator comprises the total resistance in the base circuit [recall that (β + 1)(re + Re) is the reflection of (re + Re) from the emitter side to the base side]. Thus the expression in Eq. (7.113) states that the voltage gain from base to collector is equal to β times the ratio of the total resistance in the collector to the total resistance in the base. The factor β appears because it is the ratio of the collector current to the base current. This general and useful expression has no counterpart in the MOS case. We observe that the overall voltage gain Gv is lower than the value without Re, namely, Gv = −β RC RL Rsig + (β + 1)re (7.114) because of the additional term (β + 1)Re in the denominator. The gain, however, is now less sensitive to the value of β, a desirable result because of the typical wide variability in the value of β. Another important consequence of including the resistance Re in the emitter is that it enables the amplifier to handle larger input signals without incurring nonlinear distortion. This is because only a fraction of the input signal at the base, vi, appears between the base and the emitter. Specifically, from the circuit in Fig. 7.38(b), we see that vπ = re vi re + Re 1 1 + gmRe (7.115) Thus, for the same vπ , the signal at the input terminal of the amplifier, vi, can be greater than for the CE amplifier by the factor (1 + gmRe). To summarize, including a resistance Re in the emitter of the CE amplifier results in the following characteristics: 1. The input resistance Rin is increased by the factor (1 + gmRe). 2. The voltage gain from base to collector, Av, is reduced by the factor (1 + gmRe). 3. For the same nonlinear distortion, the input signal vi can be increased by the factor (1 + gmRe). 4. The overall voltage gain is less dependent on the value of β. 5. The high-frequency response is significantly improved (as we shall see in Chapter 10). With the exception of gain reduction, these characteristics represent performance improvements. Indeed, the reduction in gain is the price paid for obtaining the other performance improvements. In many cases this is a good bargain; it is the underlying philosophy for the use of negative feedback. That the resistance Re introduces negative feedback in the amplifier circuit can be verified by utilizing a procedure similar to that we used above for the MOSFET case. In Chapter 11, where we shall study negative feedback formally, we will find that the factor (1 + gmRe), which appears repeatedly, is the “amount of negative feedback” introduced by Re. Finally, we note that the negative-feedback action of Re gives it the name emitter degeneration resistance. 438 Chapter 7 Transistor Amplifiers Example 7.9 For the CE amplifier specified in Example 7.8, what value of Re is needed to raise Rin to a value four times that of Rsig? With Re included, find Av o, Ro, Av, and Gv. Also, if vˆπ is limited to 5 mV, what are the corresponding values of vˆsig and vˆo? Solution To obtain Rin = 4Rsig = 4 × 5 = 20 k , the required Re is found from 20 = (β + 1) re + Re With β = 100, Thus, re + Re 200 For vˆπ = 5 mV, Re = 200 − 25 = 175 Av o = −α re RC + Re − 5000 25 + 175 = −25 V/V Ro = RC = 5 k (unchanged) Av = Av o RL RL + Ro = −25 × 5 5+5 = −12.5 V/V Gv = Rin Rin + Rsig Av = − 20 20 + 5 × 12.5 = −10 V/V vˆi = vˆπ re + Re re = 5 1 + 175 25 vˆsig = vˆi Rin + Rsig Rin = 40 1 + 5 20 = 40 mV = 50 mV vˆo = vˆsig × Gv = 50 × 10 = 500 mV = 0.5 V Thus, while Gv has decreased to about a third of its original value, the amplifier is able to produce as large an output signal as before for the same nonlinear distortion. 7.3 Basic Configurations 439 EXERCISE 7.24 Show that with Re included, and vπ limited to a maximum value vˆπ , the maximum allowable input signal, vˆsig, is given by vˆsig = vˆπ 1 + Re + Rsig re rπ If the transistor is biased at IC = 0.5 mA and has a β of 100, what value of Re is needed to permit an input signal vˆsig of 100 mV from a source with a resistance Rsig = 10 k while limiting vˆπ to 10 mV? What is Rin for this amplifier? If the total resistance in the collector is 10 k , what Gv value results? Ans. 350 ; 40.4 k ; −19.8 V/V 7.3.5 The Common-Gate (CG) and the Common-Base (CB) Amplifiers Figure 7.39(a) shows a common-gate amplifier with the biasing circuit omitted. The amplifier is fed with a signal source characterized by vsig and Rsig. Since Rsig appears in series with the source, it is more convenient to represent the transistor with the T model than with the π model. Doing this, we obtain the amplifier equivalent circuit shown in Fig. 7.39(b). From inspection of the equivalent circuit of Fig. 7.39(b), we see that the input resistance Rin = 1 gm (7.116) Figure 7.39 (a) Common-gate (CG) amplifier with bias arrangement omitted. (b) Equivalent circuit of the CG amplifier with the MOSFET replaced with its T model. 440 Chapter 7 Transistor Amplifiers This should have been expected, since we are looking into the source and the gate is grounded. Typically 1/gm is a few hundred ohms; thus the CG amplifier has a low input resistance. To determine the voltage gain Av o, we write at the drain node vo = −iRD and substitute for the source current i from i = − vi 1/gm to obtain Av o ≡ vo vi = gmRD (7.117) which except for the positive sign is identical to the expression for Avo of the CS amplifier. The output resistance of the CG circuit can be found by inspection of the circuit in Fig. 7.39(b) as Ro = RD (7.118) which is the same as in the case of the CS amplifier. Although the gain of the CG amplifier proper has the same magnitude as that of the CS amplifier, this is usually not the case as far as the overall voltage gain is concerned. The low input resistance of the CG amplifier can cause the input signal to be severely attenuated. Specifically, vi = Rin = 1/gm vsig Rin + Rsig 1/gm + Rsig (7.119) from which we see that except for situations in which Rsig is on the order of 1/gm, the signal transmission factor vi/vsig can be very small and the overall voltage gain Gv can be correspondingly small. Specifically, with a resistance RL connected at the output Gv = 1/gm Rsig + 1/gm [gm(RD RL )] Thus, Gv = (RD RL) Rsig + 1/gm (7.120) Observe that the overall voltage gain is simply the ratio of the total resistance in the drain circuit to the total resistance in the source circuit. If Rsig is of the same order as RD and RL, Gv will be very small. Because of its low input resistance, the CG amplifier alone has very limited application. One such application is to amplify high-frequency signals that come from sources with relatively low resistances. These include cables, where it is usually necessary for the input resistance of the amplifier to match the characteristic resistance of the cable. As will be shown in Chapter 10, the CG amplifier has excellent high-frequency response. Thus it can be combined with the CS amplifier in a very beneficial way that takes advantage of the best features of each of the two configurations. A very significant circuit of this kind will be studied in Chapter 8. 7.3 Basic Configurations 441 EXERCISE 7.25 A CG amplifier is required to match a signal source with Rsig = 100 . At what current ID should the MOSFET be biased if it is operated at an overdrive voltage of 0.20 V? If the total resistance in the drain circuit is 2 k , what overall voltage gain is realized? Ans. 1 mA; 10 V/V Very similar results can be obtained for the CB amplifier shown in Fig. 7.40(a). Specifically, from the equivalent circuit in Fig. 7.40(b) we can find α Rin = re = gm 1/gm (7.121) α Av o = re RC = gmRC (7.122) Ro = RC (7.123) and with a load resistance RL connected to the output, the overall voltage gain is given by Gv ≡ vo vsig = α RC RL Rsig + re (7.124) RC vo RC vo Rsig re Rsig ie vsig ϩ Ϫ vi Ro vsig ϩϪ vi Rin (a) (b) Figure 7.40 (a) CB amplifier with bias details omitted; (b) amplifier equivalent circuit with the BJT represented by its T model. 442 Chapter 7 Transistor Amplifiers Since α 1, we see that as in the case of the CG amplifier, the overall voltage gain is simply the ratio of the total resistance in the collector to the total resistance in the emitter. We also note that the overall voltage gain is almost independent of the value of β (except through the small dependence of α on β), a desirable property. Observe that for Rsig of the same order as RC and RL, the gain will be very small. In summary, the CB and CG amplifiers exhibit a very low input resistance (1/gm), an open-circuit voltage gain that is positive and equal in magnitude to that of the CE (CG) amplifier (gmRC or gmRD), and, like the CE (CS) amplifier, a relatively high output resistance (RC or RD). Because of its very low input resistance, the CB (CG) circuit alone is not attractive as a voltage amplifier except in specialized applications, such as the cable amplifier mentioned above. The CB (CG) amplifier has excellent high-frequency performance, which as we shall see in Chapters 8 and 10, makes it useful in combination with other circuits in the implementation of high-frequency amplifiers. EXERCISES 7.26 Consider a CB amplifier utilizing a BJT biased at IC = 1 mA and with RC = 5 k . Determine Rin, Av o, and Ro. If the amplifier is loaded in RL = 5 k , what value of Av results? What Gv is obtained if Rsig = 5 k ? Ans. 25 ; 200 V/V; 5 k ; 100 V/V; 0.5 V/V 7.27 A CB amplifier is required to amplify a signal delivered by a coaxial cable having a characteristic resistance of 50 . What bias current IC should be utilized to obtain Rin that is matched to the cable resistance? To obtain an overall voltage gain of Gv of 40 V/V, what should the total resistance in the collector (i.e., RC RL) be? Ans. 0.5 mA; 4 k 7.3.6 The Source and Emitter Followers The last of the basic transistor amplifier configurations is the common-drain (common-collector) amplifier, an important circuit that finds application in the design of both small-signal amplifiers and amplifiers that are required to handle large signals and deliver substantial amounts of signal power to a load. This latter variety will be studied in Chapter 12. The common-drain amplifier is more commonly known as the source follower, and the common-collector amplifier is more commonly known as the emitter follower. The reason behind these names will become apparent shortly. The Need for Voltage Buffers Before embarking on the analysis of the source and the emitter followers, it is useful to look at one of their more common applications. Consider the situation depicted in Fig. 7.41(a). A signal source delivering a signal of reasonable strength (1 V) with an internal resistance of 1 M is to be connected to a 1-k load resistance. Connecting the source to the load directly as in Fig. 7.41(b) would result in severe attenuation 7.3 Basic Configurations 443 Rsig = 1 M⍀ vsig = 1V ϩ Ϫ RL 1 k⍀ Rsig ϭ 1 M⍀ vsig = 1V ϩ Ϫ RL 1 k⍀ ϩ vo Ӎ 1 mV Ϫ (a) (b) vsig = 1 V Rsig = 1 M⍀ ϩ Ϫ Ӎ 1V Avoϭ1 Ro = 100 ⍀ RL 1 k⍀ ϩ vo Ӎ 0.9 V Ϫ Rin very large (c) Figure 7.41 Illustrating the need for a unity-gain voltage buffer amplifier. of the signal; the signal appearing across the load will be only 1/(1000 + 1) of the input signal, or about 1 mV. An alternative course of action is suggested in Fig. 7.41(c). Here we have interposed an amplifier between the source and the load. Our amplifier, however, is unlike the amplifiers we have been studying in this chapter thus far; it has a voltage gain of only unity. This is because our signal is already of sufficient strength and we do not need to increase its amplitude. Note, however, that our amplifier has a very high input resistance, thus almost all of vsig (i.e., 1 V) will appear at the input of the amplifier proper. Since the amplifier has a low output resistance (100 ), 90% of this signal (0.9 V) will appear at the output, obviously a very significant improvement over the situation without the amplifier. As will be seen next, the source follower can easily implement the unity-gain buffer amplifier shown in Fig. 7.41(c). Characteristic Parameters of the Source Follower Figure 7.42(a) shows a source follower with the bias circuit omitted. The source follower is fed with a signal generator (vsig, Rsig) and has a load resistance RL connected between the source terminal and ground. We shall assume that RL includes both the actual load and any other resistance that may be present between the source terminal and ground (e.g., for biasing purposes). Normally, the actual load resistance would be much lower in value than such other resistances and thus would dominate. Since the MOSFET has a resistance RL connected in its source terminal, it is most convenient to use the T model, as shown in Fig. 7.40(b). From the latter circuit we can write by inspection Rin = ∞ 444 Chapter 7 Transistor Amplifiers Rsig ϩ vsig ϩ Ϫ vi Ϫ Rin (a) ϩ RL vo Ϫ Ro i Rsig 0 vsig ϩ Ϫ i ϩ 1 gm vi RL Ϫ Rin = ϱ Ro = 1 gm ϩ vo Ϫ (b) Figure 7.42 (a) Common-drain amplifier or source follower with the bias circuit omitted. (b) Equivalent circuit of the source follower obtained by replacing the MOSFET with its T model. and obtain Av from the voltage divider formed by 1/gm and RL as Setting RL = ∞ we obtain Av ≡ vo vi = RL RL + 1/gm Av o = 1 (7.125) (7.126) The output resistance Ro is found by setting vi = 0 (i.e., by grounding the gate). Now looking back into the output terminal, excluding RL, we simply see 1/gm, thus Ro = 1/gm (7.127) The unity open-circuit voltage gain together with Ro in Eq. (7.127) can be used to find Av when a load resistance RL is connected. The result is simply the expression in Eq. (7.125). 7.3 Basic Configurations 445 Finally, because of the infinite Rin, vi = vsig, and the overall voltage gain is Gv = Av = RL RL + 1/gm (7.128) Thus Gv will be lower than unity. However, because 1/gm is usually low, the voltage gain can be close to unity. The unity open-circuit voltage gain in Eq. (7.126) indicates that the voltage at the source terminal will follow that at the input, hence the name source follower. In conclusion, the source follower features a very high input resistance (ideally, infinite), a relatively low output resistance (1/gm), and an open-circuit voltage gain that is near unity (ideally, unity). Thus the source follower is ideally suited for implementing the unity-gain voltage buffer of Fig. 7.41(c). The source follower is also used as the output (i.e., last) stage in a multistage amplifier, where its function is to equip the overall amplifier with a low output resistance, thus enabling it to supply relatively large load currents without loss of gain (i.e., with little reduction of output signal level). The design of output stages is studied in Chapter 12. EXERCISES D7.28 It is required to design a source follower that implements the buffer amplifier shown in Fig. 7.41(c). If the MOSFET is operated with an overdrive voltage VOV = 0.25 V, at what drain current should it be biased? Find the output signal amplitude and the signal amplitude between gate and source. Ans. 1.25 mA; 0.91 V; 91 mV D7.29 A MOSFET is connected in the source-follower configuration and employed as the output stage of a cascade amplifier. It is required to provide an output resistance of 200 . If the MOSFET has kn = 0.4 mA/V2 and is operated at VOV = 0.25 V, find the required W/L ratio. Also specify the dc bias current ID. If the amplifier load resistance varies over the range 1 k to 10 k , what is the range of Gv of the source follower? Ans. 50; 0.625 mA; 0.83 V/V to 0.98 V/V Characteristic Parameters of the Emitter Follower Although the emitter follower does not have an infinite input resistance (as in the case of the source follower), it is still widely used as a voltage buffer. In fact, it is a very versatile and popular circuit. We will therefore study it in some detail. Figure 7.43(a) shows an emitter follower with the equivalent circuit shown in Fig. 7.43(b). The input resistance Rin is found from Rin = vi ib Substituting for ib = ie/(β + 1) where ie is given by ie = re vi + RL 446 Chapter 7 Transistor Amplifiers Rsig vsig ϩ Ϫ vi Rin (a) RL vo Ro Rsig vsig ϩ Ϫ ie re vi RL vo (b) Figure 7.43 (a) Common-collector amplifier or emitter follower with the bias circuit omitted. (b) Equivalent circuit obtained by replacing the BJT with its T model. we obtain Rin = (β + 1)(re + RL) (7.129) a result that we could have written directly, utilizing the resistance-reflection rule. Note that as expected the emitter follower takes the low load resistance and reflects it to the base side, where the signal source is, after increasing its value by a factor (β + 1). It is this impedance transformation property of the emitter follower that makes it useful in 7.3 Basic Configurations 447 connecting a low-resistance load to a high-resistance source, that is, to implement a buffer amplifier. The voltage gain Av is given by Av ≡ vo vi = RL RL + re (7.130) Setting RL = ∞ yields Av o, Av o = 1 (7.131) Thus, as expected, the open-circuit voltage gain of the emitter follower proper is unity, which means that the signal voltage at the emitter follows that at the base, which is the origin of the name “emitter follower.” To determine Ro, refer to Fig. 7.43(b) and look back into the emitter (i.e., behind or excluding RL) while setting vi = 0 (i.e., grounding the base). You will see re of the BJT, thus Ro = re (7.132) This result together with Avo = 1 yields Av in Eq. (7.130), thus confirming our earlier analysis. We next determine the overall voltage gain Gv, as follows: vi = Rin vsig Rin + Rsig = (β + 1)(re + RL) (β + 1)(re + RL) + Rsig Gv ≡ vo v sig = vi v sig × Av Substituting for Av from Eq. (7.130) results in Gv = (β + (β + 1)RL 1)RL + (β + 1)re + Rsig (7.133) This equation indicates that the overall gain, though lower than one, can be close to one if (β + 1)RL is larger or comparable in value to Rsig. This again confirms the action of the emitter follower in delivering a large proportion of vsig to a low-valued load resistance RL even though Rsig can be much larger than RL. The key point is that RL is multiplied by (β + 1) before it is “presented to the source.” Figure 7.44(a) shows an equivalent circuit of the emitter follower obtained by simply reflecting re and RL to the base side. The overall voltage gain Gv ≡ vo/vsig can be determined directly and very simply from this circuit by using the voltage divider rule. The result is the expression for Gv already given in Eq. (7.133). Dividing all resistances in the circuit of Fig. 7.44(a) by β + 1 does not change the voltage ratio vo/vsig. Thus we obtain another equivalent circuit, shown in Fig. 7.44(b), that can be used to determine Gv ≡ vo/vsig of the emitter follower. A glance at this circuit reveals that it is simply the equivalent circuit obtained by reflecting vsig and Rsig from the base side to the emitter side. In this reflection, vsig does not change, but Rsig is divided by β + 1. Thus, we 448 Chapter 7 Transistor Amplifiers Rsig vsig ϩ Ϫ vsig ϩϪ vo re RL vo (a) (b) Figure 7.44 Simple equivalent circuits for the emitter follower obtained by (a) reflecting re and RL to the base side, and (b) reflecting vsig and Rsig to the emitter side. Note that the circuit in (b) can be obtained from that in (a) by simply dividing all resistances by (β + 1). either reflect to the base side and obtain the circuit in Fig. 7.44(a) or reflect to the emitter side and obtain the circuit in Fig. 7.44(b). From the latter, Gv can be found as Gv ≡ vo v sig = RL + re RL + Rsig/(β + 1) (7.134) Observe that this expression is the same as that in Eq. (7.133) except for dividing both the numerator and denominator by β + 1. The expression for Gv in Eq. (7.134) has an interesting interpretation: The emitter follower reduces Rsig by the factor (β + 1) before “presenting it to the load resistance RL” : an impedance transformation that has the same buffering effect. At this point it is important to note that although the emitter follower does not provide voltage gain it has a current gain of β + 1. Thévenin Representation of the Emitter-Follower Output A more general representation of the emitter-follower output is shown in Fig. 7.45(a). Here Gvo is the overall open-circuit voltage gain that can be obtained by setting RL = ∞ in the circuit of Fig. 7.44(b), as illustrated in Fig. 7.45(b). The result is Gv o = 1. The output resistance Rout is different from Ro. To determine Rout we set vsig to zero (rather than setting vi to zero). Again we can use the equivalent circuit in Fig. 7.44(b) to do this, as illustrated in Fig. 7.45(c). We see that Rout = re + Rsig β +1 (7.135) Finally, we show in Fig. 7.45(d) the emitter-follower circuit together with its Rin and Rout. Observe that Rin is determined by reflecting re and RL to the base side (by multiplying their values by β + 1). To determine Rout, grab hold of the emitter and walk (or just look!) backward while vsig = 0. You will see re in series with Rsig, which because it is in the base must be divided by (β + 1). We note that unlike the amplifier circuits we studied earlier, the emitter follower is not unilateral. This is manifested by the fact that Rin depends on RL and Rout depends on Rsig. 7.3 Basic Configurations 449 vsig ϩ re Ϫ (b) Rsig re E ϩ vsig Ϫ Rsig re RL (c) (d) Figure 7.45 (a) The´venin representation of the output of the emitter follower. (b) Obtaining Gvo from the equivalent circuit in Fig. 7.44(b). (c) Obtaining Rout from the equivalent circuit in Fig. 7.44(b) with vsig set to zero. (d) The emitter follower with Rin and Rout determined simply by looking into the input and output terminals, respectively. Example 7.10 It is required to design an emitter follower to implement the buffer amplifier of Fig. 7.46(a). Specify the required bias current IE and the minimum value the transistor β must have. Determine the maximum allowed value of vsig if vπ is to be limited to 5 mV in order to obtain reasonably linear operation. With vsig = 200 mV, determine the signal voltage at the output if RL is changed to 2 k , and to 0.5 k . Rsig 100 k vsig 200 mV Ro 10 Avo 1 RL 1k vo Figure 7.46 Circuit for Example 7.10. Rin 100 k (a) 450 Chapter 7 Transistor Amplifiers Example 7.10 continued Rsig 100 k vsig 200 mV voRo 10 RL 1 k Rin 100 k (b) Figure 7.46 continued Solution The emitter-follower circuit is shown in Fig. 7.46(b). To obtain Ro = 10 , we bias the transistor to obtain re = 10 . Thus, 10 = VT IE IE = 2.5 mA The input resistance Rin will be Rin = (β + 1) re + RL 100 = (β + 1)(0.01 + 1) Thus, the BJT should have a β with a minimum value of 98. A higher β would obviously be beneficial. The overall voltage gain can be determined from Gv ≡ vo v sig = RL RL + re + Rsig (β + 1) Assuming β = 100, the value of Gv obtained is Gv = 0.5 7.3 Basic Configurations 451 Thus when vsig = 200 mV, the signal at the output will be 100 mV. Since the 100 mV appears across the 1-k load, the signal across the base–emitter junction can be found from vπ = vo RL × re = 100 × 10 = 1 mV 1000 If vˆπ = 5 mV then vsig can be increased by a factor of 5, resulting in vˆsig = 1 V. To obtain vo as the load is varied, we use the The´venin equivalent of the emitter follower, shown in Fig. 7.45(a) with Gv o = 1 and Rout = Rsig β +1 + re = 100 101 + 0.01 = 1 k to obtain For RL = 2 k , vo = v sig RL RL + Rout 2 vo = 200 mV × 2 + 1 = 133.3 mV and for RL = 0.5 k , 0.5 vo = 200 mV × 0.5 + 1 = 66.7 mV EXERCISE 7.30 An emitter follower utilizes a transistor with β = 100 and is biased at IC = 5 mA. It operates between a source having a resistance of 10 k and a load of 1 k . Find Rin, Gv o, Rout, and Gv. What is the peak amplitude of vsig that results in vπ having a peak amplitude of 5 mV? Find the resulting peak amplitude at the output. Ans. 101.5 k ; 1 V/V; 104 ; 0.91 V/V; 1.1 V; 1 V 452 Chapter 7 Transistor Amplifiers 7.3.7 Summary Tables and Comparisons For easy reference and to enable comparisons, we present in Tables 7.4 and 7.5 the formulas for determining the characteristic parameters for the various configurations of MOSFET and BJT amplifiers, respectively. In addition to the remarks made throughout this section about the characteristics and areas of applicability of the various configurations, we make the following concluding points: 1. MOS amplifiers provide much higher, ideally infinite input resistances (except, of course, for the CG configuration). This is a definite advantage over BJT amplifiers. 2. BJTs exhibit higher gm values than MOSFETs, resulting in higher gains. 3. For discrete-circuit amplifiers—that is, those that are assembled from discrete components on a printed-circuit board (PCB)—the BJT remains the device of choice. This is because discrete BJTs are much easier to handle physically than discrete MOSFETs and, more important, a very wide variety of discrete BJTs is available commercially. The remainder of this chapter is concerned with discrete-circuit amplifiers. 4. Integrated-circuit (IC) amplifiers predominantly use MOSFETs, although BJTs are utilized in certain niche areas. Chapters 8 to 13 are mainly concerned with IC amplifiers. 5. The CS and CE configurations are the best suited for realizing the bulk of the gain required in an amplifier. Depending on the magnitude of the gain required, either a single stage or a cascade of two or three stages can be used. 6. Including a resistance Rs in the source of the CS amplifier (a resistance Re in the emitter of the CE amplifier) provides a number of performance improvements at the expense of gain reduction. Table 7.4 Characteristics of MOSFET Amplifiers Amplifier type Rin Av o Common source (Fig. 7.35) ∞ −gm RD Common source with Rs (Fig. 7.37) ∞ − 1 gm RD + gmRs Common gate (Fig. 7.39) Source follower (Fig. 7.42) 1 gm gm RD ∞1 a For the interpretation of Rin, Av o, and Ro, refer to Fig. 7.34(b). Characteristicsa Ro Av RD −gm RD RL RD −gm RD RL 1 + gmRs − RD 1/gm RL + Rs RD gm RD RL 1 RL gm RL + 1/gm Gv −gm RD RL − gm 1 RD + gm RL Rs − RD 1/gm RL + Rs RD RL Rsig + 1/gm RL RL + 1/gm 7.3 Basic Configurations 453 Table 7.5 Characteristics of BJT Amplifiersa,b Rin Av o Ro Av Gv Common emitter (Fig. 7.36) (β + 1)re −gm RC RC −gm RC RL −α RC RL re −β RC RL Rsig + (β + 1)re Common emitter with Re (Fig. 7.38) (β + 1) re + Re − gmRC 1 + gmRe RC −gm RC RL 1 + gmRe −α RC RL re + Re −β RC RL Rsig + (β + 1) re + Re Common base re (Fig. 7.40) gm RC RC gm RC RL α RC RL re α RC Rsig RL + re Emitter follower (Fig. 7.43) (β + 1) re + RL 1 re RL RL + re RL RL + re + Rsig/(β + 1) Gv o = 1 Rout = re + Rsig β +1 a For the interpretation of Rm, Av o, and Ro refer to Fig. 7.34. b Setting β = ∞ (α = 1) and replacing re with 1/gm, RC with RD, and Re with Rs results in the corresponding formulas for MOSFET amplifiers (Table 7.4). 7. The low input resistance of the CG and CB amplifiers makes them useful only in specific applications. As we shall see in Chapter 10, these two configurations exhibit a much better high-frequency response than that available from the CS and CE amplifiers. This makes them useful as high-frequency amplifiers, especially when combined with the CS or CE circuit. We shall study one such combination in Chapter 8. 8. The source follower (emitter follower) finds application as a voltage buffer for connecting a high-resistance source to a low-resistance load, and as the output stage in a multistage amplifier, where its purpose is to equip the amplifier with a low output resistance. 7.3.8 When and How to Include the Output Resistance ro So far we have been neglecting the output resistance ro of the MOSFET and the BJT. We have done this for two reasons: 1. To keep things simple and focus attention on the significant features of each of the basic configurations, and 2. Because our main interest in this chapter is discrete-circuit design, where the circuit resistances (e.g., RC, RD, and RL) are usually much smaller than ro. 454 Chapter 7 Transistor Amplifiers LEE DE FOREST—A FATHER OF THE ELECTRONICS AGE: In 1906 self-employed inventor Lee de Forest (1873–1961) created a three-terminal vacuum tube; it was the first electronic amplifier of weak signals. The device was known initially as the de Forest valve. The patent filed in 1907, however, used the name Audion, with the “-ion” indicating that the device was not completely evacuated. By 1919, engineers had realized that complete evacuation of internal gases produced a more reliable device. De Forest’s first amplifier became known as the vacuum tube triode. Through its impact on radio, telephony, motion picture sound, and television, this invention, one of de Forest’s 180 patents, is credited with introducing the electronics age. The vacuum tube, in a variety of types, remained the device for implementing amplifiers until the appearance of transistors in the early 1950s. Nevertheless, in some instances it is relatively easy to include ro in the analysis. Specifically: 1. In the CS and CE amplifiers, it can be seen that ro of the transistor appears in parallel with RD and RC, respectively, and can be simply included in the corresponding formulas in Tables 7.4 and 7.5 by replacing RD with (RD ro) and RC with (RC ro). The effect will be a reduction in the magnitude of gain, of perhaps 5% to 10%. 2. In the source and emitter followers, it can be seen that the transistor ro appears in parallel with RL and can be taken into account by replacing RL in the corresponding formulas with (RL ro). Thus, here too, the effect of taking ro into account is a small reduction in gain. More significant, however, taking ro into account reduces the open-circuit voltage gain Av o from unity to Av o = ro + ro (1/gm) (7.136) There are configurations in which taking ro into account complicates the analysis considerably. These are the CS (CE) amplifiers with a source (emitter) resistance, and the CG (CB) amplifier. Fortunately, for discrete implementation of these configurations, the effect of neglecting ro is usually small (which can be verified by computer simulation). Finally, a very important point: In the analysis and design of IC amplifiers, ro must always be taken into account. This is because, as will be seen in the next chapter, all the circuit resistances are of the same order of magnitude as ro; thus, neglecting ro can result in completely erroneous results. 7.4 Biasing As discussed in Section 7.1, an essential step in the design of a transistor amplifier is the establishment of an appropriate dc operating point for the transistor. This is the step known as biasing or bias design. In this section, we study the biasing methods commonly employed in discrete-circuit amplifiers. Biasing of integrated-circuit amplifiers will be studied in Chapter 8. Bias design aims to establish in the drain (collector) a dc current that is predictable and insensitive to variations in temperature and to the large variations in parameter values between devices of the same type. For instance, discrete BJTs belonging to the same manufacturer’s part number can exhibit β values that range, say, from 50 to 150. Nevertheless, the bias design 7.4 Biasing 455 for an amplifier utilizing this particular transistor type may specify that the dc collector current shall always be within, say, ±10% of the nominal value of, say, 1 mA. A similar statement can be made about the desired insensitivity of the dc drain current to the wide variations encountered in Vt of discrete MOSFETs. A second consideration in bias design is locating the dc operating point in the active region of operation of the transistor so as to obtain high voltage gain while allowing for the required output signal swing without the transistor leaving the active region at any time (in order to avoid nonlinear distortion). We discussed this point in Section 7.1.7. Although we shall consider the biasing of MOSFET and BJT amplifiers separately, the resulting circuits are very similar. Also, it will be seen that good bias designs incorporate a feedback mechanism that works to keep the dc bias point as constant as possible. In order to keep matters simple and thus focus our attention on significant issues, we will neglect the Early effect; that is assume λ = 0 or VA = ∞. This is certainly allowed in initial designs of discrete circuits. Of course, the design can be fine-tuned at a later point with the assistance of a circuit-simulation program such as SPICE. 7.4.1 The MOSFET Case Biasing by Fixing VGS The most straightforward approach to biasing a MOSFET is to fix its gate-to-source voltage VGS to the value required6 to provide the desired ID. This voltage value can be derived from the power-supply voltage VDD through the use of an appropriate voltage divider, as shown in Fig. 7.47(a). Alternatively, it can be derived from another suitable reference voltage that might be available in the system. Independent of how the voltage VGS may be generated, this is not a good approach to biasing a MOSFET. To understand the reason for this statement, recall that ID = 1 2 μn Co x W L (VGS − Vt )2 and note that the values of the threshold voltage Vt, the oxide-capacitance Cox, and (to a lesser extent) the transistor aspect ratio W/L vary widely among devices of supposedly the same size and type. This is certainly the case for discrete devices, in which large spreads in the values of these parameters occur among devices of the same manufacturer’s part number. The spread can also be large in integrated circuits, especially among devices fabricated on different wafers and certainly between different batches of wafers. Furthermore, both Vt and μn depend on temperature, with the result that if we fix the value of VGS, the drain current ID becomes very much temperature dependent. To emphasize the point that biasing by fixing VGS is not a good technique, we show in Fig. 7.47 two iD–vGS characteristic curves representing extreme values in a batch of MOSFETs of the same type. Observe that for the fixed value of VGS, the resultant spread in the values of the drain current can be substantial. Biasing by Fixing VG and Connecting a Resistance in the Source An excellent biasing technique for discrete MOSFET circuits consists of fixing the dc voltage at the gate, VG, and connecting a resistance in the source lead, as shown in Fig. 7.48(a). For this circuit 6That is indeed what we were doing in Section 7.1. However, the amplifier circuits studied there were conceptual ones, not actual practical circuits. Our purpose in this section is to study the latter. 456 Chapter 7 Transistor Amplifiers iD Device 2 VDD ID2 Device 1 RG1 RD ID ϩ RG2 VGS Ϫ ID1 0 VGS vGS (a) (b) Figure 7.47 (a) Biasing the MOSFET with a constant VGS generated from VDD using a voltage divider (RG1, RG2); (b) the use of fixed bias (constant VGS) can result in a large variability in the value of ID. Devices 1 and 2 represent extremes among units of the same type. we can write VG = VGS + RSID (7.137) Now, if VG is much greater than VGS, ID will be mostly determined by the values of VG and RS. However, even if VG is not much larger than VGS, resistor RS provides negative feedback, which acts to stabilize the value of the bias current ID. To see how this comes about, consider what happens when ID increases for whatever reason. Equation (7.137) indicates that since VG is constant, VGS will have to decrease. This in turn results in a decrease in ID, a change that is opposite to that initially assumed. Thus the action of RS works to keep ID as constant as possible.7 Figure 7.48(b) provides a graphical illustration of the effectiveness of this biasing scheme. Here too we show the iD–vGS characteristics for two devices that represent the extremes of a batch of MOSFETs. Superimposed on the device characteristics is a straight line that represents the constraint imposed by the bias circuit—namely, Eq. (7.137). The intersection of this straight line with the iD–vGS characteristic curve provides the coordinates (ID and VGS) of the bias point. Observe that compared to the case of fixed VGS, here the variability obtained in ID is much smaller. Also, note that the variability decreases as VG and RS are made larger (thus providing a bias line that is less steep). Two possible practical discrete implementations of this bias scheme are shown in Fig. 7.48(c) and (e). The circuit in Fig. 7.48(c) utilizes one power-supply VDD and derives VG 7The action of RS in stabilizing the value of the bias current ID is not unlike that of the resistance Rs, which we included in the source lead of a CS amplifier in Section 7.3.4. In the latter case also, Rs works to reduce the change in iD with the result that the amplifier gain is reduced. 7.4 Biasing 457 iD ID ϩ ϩ VGS Ϫ ID VG RS Ϫ ID2 ID1 0 (a) VDD RG1 RD ID 0 VG ϩ VGS ID Ϫ RG2 RS Rsig vsig ϩ Ϫ Device 2 Device 1 Slope = Ϫ1͞RS VGS2 VGS1 (b) VG vGS VDD VDD RD RG1 CC1 RG2 RD 0 ID ϩ RG VGS ID Ϫ RS RS ϪVSS (c) (d) (e) Figure 7.48 Biasing using a fixed voltage at the gate, VG, and a resistance in the source lead, RS: (a) basic arrangement; (b) reduced variability in ID; (c) practical implementation using a single supply; (d) coupling of a signal source to the gate using a capacitor CC1; (e) practical implementation using two supplies. through a voltage divider (RG1, RG2). Since IG = 0, RG1 and RG2 can be selected to be very large (in the megohm range), allowing the MOSFET to present a large input resistance to a signal source that may be connected to the gate through a coupling capacitor, as shown in Fig. 7.48(d). Here capacitor CC1 blocks dc and thus allows us to couple the signal vsig to the amplifier input without disturbing the MOSFET dc bias point. The value of CC1 should be selected large enough to approximate a short circuit at all signal frequencies of interest. We shall study capacitively coupled MOSFET amplifiers, which are suitable only in discrete-circuit design, in Section 7.5. Finally, note that in the circuit of Fig. 7.48(c), resistor RD is selected to be as large as possible to obtain high gain but small enough to allow for the desired signal swing at the drain while keeping the MOSFET in saturation at all times. When two power supplies are available, as is often the case, the somewhat simpler bias arrangement of Fig. 7.48(e) can be utilized. This circuit is an implementation of Eq. (7.137), with VG replaced by VSS. Resistor RG establishes a dc ground at the gate and presents a high input resistance to a signal source that may be connected to the gate through a coupling capacitor. 458 Chapter 7 Transistor Amplifiers Example 7.11 It is required to design the circuit of Fig. 7.48(c) to establish a dc drain current ID = 0.5 mA. The MOSFET is specified to have Vt = 1 V and knW/L = 1 mA/V2. For simplicity, neglect the channel-length modulation effect (i.e., assume λ = 0). Use a power-supply VDD = 15 V. Calculate the percentage change in the value of ID obtained when the MOSFET is replaced with another unit having the same knW/L but Vt = 1.5 V. Solution As a rule of thumb for designing this classical biasing circuit, we choose RD and RS to provide one-third of the power-supply voltage VDD as a drop across each of RD, the transistor (i.e., VDS), and RS. For VDD = 15 V, this choice makes VD = +10 V and VS = +5 V. Now, since ID is required to be 0.5 mA, we can find the values of RD and RS as follows: RD = VDD − VD ID = 15 − 10 0.5 = 10 k RS = VS RS = 5 0.5 = 10 k The required value of VGS can be determined by first calculating the overdrive voltage VOV from which yields VOV = 1 V, and thus, ID = 1 2 kn (W/L)VO2V 0.5 = 1 2 × 1 × VO2V VGS = Vt + VOV = 1 + 1 = 2 V Now, since VS = +5 V, VG must be VG = VS + VGS = 5 + 2 = 7 V To establish this voltage at the gate we may select RG1 = 8 M and RG2 = 7 M . The final circuit is shown in Fig. 7.49. Observe that the dc voltage at the drain (+10 V) allows for a positive signal swing of +5 V (i.e., up to VDD) and a negative signal swing of 4 V [i.e., down to (VG – Vt)]. If the NMOS transistor is replaced with another having Vt = 1.5 V, the new value of ID can be found as follows: ID = 1 2 ×1× VGS − 1.5 2 VG = VGS + IDRS 7 = VGS + 10ID (7.138) (7.139) 7.4 Biasing 459 VDD = ϩ15 V 8 M⍀ ID = 0.5 mA RD = 10 k⍀ VG = ϩ7 V VD = ϩ10 V 7 M⍀ ID = 0.5 mA VS = ϩ5 V RS = 10 k⍀ Figure 7.49 Circuit for Example 7.11. Solving Eqs. (7.138) and (7.139) together yields ID = 0.455 mA Thus the change in ID is ID = 0.455 − 0.5 = −0.045 mA which is −0.045 × 100 = −9% change. 0.5 EXERCISES 7.31 Consider the MOSFET in Example 7.11 when fixed-VGS bias is used. Find the required value of VGS to establish a dc bias current ID = 0.5 mA. Recall that the device parameters are Vt = 1 V, knW/L = 1 mA/V2, and λ = 0. What is the percentage change in ID obtained when the transistor is replaced with another having Vt = 1.5 V? Ans. VGS = 2 V; –75% D7.32 Design the circuit of Fig. 7.48(e) to operate at a dc drain current of 0.5 mA and VD = +2 V. Let Vt = 1 V, knW/L = 1 mA/V2, λ = 0, VDD = VSS = 5 V. Use standard 5% resistor values (see Appendix J), and give the resulting values of ID, VD, and VS. Ans. RD = RS = 6.2 k ; ID = 0.49 mA, VS = −1.96 V, and VD = +1.96 V. RG can be selected in the range of 1 M to 10 M . 460 Chapter 7 Transistor Amplifiers VDD RD 0 RG ID ID ϩ ϩ VDS VGS ϪϪ Figure 7.50 Biasing the MOSFET using a large drain-to-gate feedback resistance, RG. Biasing Using a Drain-to-Gate Feedback Resistor A simple and effective discrete- circuit biasing arrangement utilizing a feedback resistor connected between the drain and the gate is shown in Fig. 7.50. Here the large feedback resistance RG (usually in the megohm range) forces the dc voltage at the gate to be equal to that at the drain (because IG = 0). Thus we can write VGS = VDS = VDD − RDID which can be rewritten in the form VDD = VGS + RDI D (7.140) which is identical in form to Eq. (7.137), which describes the operation of the bias scheme discussed above [that in Fig. 7.48(a)]. Thus, here too, if ID for some reason changes, say increases, then Eq. (7.140) indicates that VGS must decrease. The decrease in VGS in turn causes a decrease in ID, a change that is opposite in direction to the one originally assumed. Thus the negative feedback or degeneration provided by RG works to keep the value of ID as constant as possible. The circuit of Fig. 7.50 can be utilized as an amplifier by applying the input voltage signal to the gate via a coupling capacitor so as not to disturb the dc bias conditions already established. The amplified output signal at the drain can be coupled to another part of the circuit, again via a capacitor. We considered such an amplifier circuit in Section 7.2 (Example 7.3). EXERCISE D7.33 Design the circuit in Fig. 7.50 to operate at a dc drain current of 0.5 mA. Assume VDD = +5 V, knW/L = 1 mA/V2, Vt = 1 V, and λ = 0. Use a standard 5% resistance value for RD, and give the actual values obtained for ID and VD. Ans. RD = 6.2 k ; ID 0.49 mA; VD 1.96 V VCC RB1 RC IC VCE IB RB2 VBE VCC RB RC IC VCE IB VBE 7.4 Biasing 461 (a) (b) Figure 7.51 Two obvious schemes for biasing the BJT: (a) by fixing VBE; (b) by fixing IB. Both result in wide variations in IC and hence in VCE and therefore are considered to be “bad.” Neither scheme is recommended. 7.4.2 The BJT Case Before presenting the “good” biasing schemes, we should point out why two obvious arrangements are not good. First, attempting to bias the BJT by fixing the voltage VBE by, for instance, using a voltage divider across the power supply VCC, as shown in Fig. 7.51(a), is not a viable approach: The very sharp exponential relationship iC−vBE means that any small and inevitable differences in VBE from the desired value will result in large differences in IC and in VCE. Second, biasing the BJT by establishing a constant current in the base, as shown in Fig. 7.51(b), where IB (VCC − 0.7)/RB, is also not a recommended approach. Here the typically large variations in the value of β among units of the same device type will result in correspondingly large variations in IC and hence in VCE. The Classical Discrete-Circuit Bias Arrangement Figure 7.52(a) shows the arrangement most commonly used for biasing a discrete-circuit transistor amplifier if only a single power supply is available. The technique consists of supplying the base of the transistor with a fraction of the supply voltage VCC through the voltage divider R1, R2. In addition, a resistor RE is connected to the emitter. This circuit is very similar to one we used for the MOSFET [Fig. 7.48(c)]. Here, however, the design must take into account the finite base current. Figure 7.52(b) shows the same circuit with the voltage-divider network replaced by its The´venin equivalent, VBB = R2 R1 + R2 VCC (7.141) RB = R1R2 R1 + R2 (7.142) The current IE can be determined by writing a Kirchhoff loop equation for the base–emitter– ground loop, labeled L, and substituting IB = IE/(β + 1): IE = RE VBB − VBE + RB/(β + 1) (7.143) 462 Chapter 7 Transistor Amplifiers VCC R1 RC R2 RE VBB VCC R2 VCC R1 R2 RC IC IB RB R1 R2 IE L RE (a) (b) Figure 7.52 Classical biasing for BJTs using a single power supply: (a) circuit; (b) circuit with the voltage divider supplying the base replaced with its The´venin equivalent. To make IE insensitive to temperature and β variation,8 we design the circuit to satisfy the following two constraints: VBB VBE RE RB β +1 (7.144) (7.145) Condition (7.144) ensures that small variations in VBE ( 0.7 V) will be swamped by the much larger VBB. There is a limit, however, on how large VBB can be: For a given value of the supply voltage VCC, the higher the value we use for VBB, the lower will be the sum of voltages across RC and the collector–base junction (VCB). On the other hand, we want the voltage across RC to be large in order to obtain high voltage gain and large signal swing (before transistor cutoff). We also want VCB (or VCE) to be large, to provide a large signal swing (before transistor saturation). Thus, as is the case in any design, we have a set of conflicting requirements, and the solution must be a trade-off. As a rule of thumb, one designs for VBB about 1 3 VCC , VCB (or VCE ) about 1 3 VCC , and IC RC about 1 3 VCC . Condition (7.145) makes IE insensitive to variations in β and could be satisfied by selecting RB small. This in turn is achieved by using low values for R1 and R2. Lower values for R1 and R2, however, will mean a higher current drain from the power supply, and will result in a lowering of the input resistance of the amplifier (if the input signal is coupled to the base),9 which is the trade-off involved in this part of the design. It should be noted that condition (7.145) means that we want to make the base voltage independent of the value of β and determined solely by the voltage divider. This will obviously be satisfied if the current in the divider is made much larger than the base current. Typically one selects R1 and R2 such that their current is in the range of IE to 0.1IE. Further insight regarding the mechanism by which the bias arrangement of Fig. 7.52(a) stabilizes the dc emitter (and hence collector) current is obtained by considering the feedback 8Bias design seeks to stabilize either IE or IC since IC = αIE and α varies very little. That is, a stable IE will result in an equally stable IC, and vice versa. 9If the input signal is coupled to the transistor base, the two bias resistances R1 and R2 effectively appear in parallel between the base and ground. Thus, low values for R1 and R2 will result in lowering Rin. 7.4 Biasing 463 action provided by RE. Consider that for some reason the emitter current increases. The voltage drop across RE, and hence VE, will increase correspondingly. Now, if the base voltage is determined primarily by the voltage divider R1, R2, which is the case if RB is small, it will remain constant, and the increase in VE will result in a corresponding decrease in VBE. This in turn reduces the collector (and emitter) current, a change opposite to that originally assumed. Thus RE provides a negative feedback action that stabilizes the bias current. This should remind the reader of the resistance Re that we included in the emitter lead of the CE amplifier in Section 7.3.4. Also, the feedback action of RE in the circuit of Fig. 7.52(a) is similar to the feedback action of RS in the circuit of Fig. 7.48(c). We shall study negative feedback formally in Chapter 11. Example 7.12 We wish to design the bias network of the amplifier in Fig. 7.52 to establish a current IE = 1 mA using a power supply VCC = +12 V. The transistor is specified to have a nominal β value of 100. Solution We shall follow the rule of thumb mentioned above and allocate one-third of the supply voltage to the voltage drop across R2 and another one-third to the voltage drop across RC, leaving one-third for possible negative signal swing at the collector. Thus, VB = +4 V VE = 4 − VBE 3.3 V and RE is determined from RE = VE IE = 3.3 1 = 3.3 k From the discussion above we select a voltage-divider current of 0.1IE = 0.1 × 1 = 0.1 mA. Neglecting the base current, we find R1 + R2 = 12 0.1 = 120 k and R2 R1 + R2 VCC = 4 V Thus R2 = 40 k and R1 = 80 k . At this point, it is desirable to find a more accurate estimate for IE, taking into account the nonzero base current. Using Eq. (7.143), 4 − 0.7 IE = 3.3( k ) + (80 40)( k ) = 0.93 mA 101 464 Chapter 7 Transistor Amplifiers Example 7.12 continued This is quite a bit lower than 1 mA, the value we are aiming for. It is easy to see from the above equation that a simple way to restore IE to its nominal value would be to reduce RE from 3.3 k by the magnitude of the second term in the denominator (0.267 k ). Thus a more suitable value for RE in this case would be RE = 3 k , which results in IE = 1.01 mA 1 mA.10 It should be noted that if we are willing to draw a higher current from the power supply and to accept a lower input resistance for the amplifier, then we may use a voltage-divider current equal, say, to IE (i.e., 1 mA), resulting in R1 = 8 k and R2 = 4 k . We shall refer to the circuit using these latter values as design 2, for which the actual value of IE using the initial value of RE of 3.3 k will be 4 − 0.7 IE = 3.3 + 0.027 = 0.99 1 mA In this case, design 2, we need not change the value of RE. Finally, the value of RC can be determined from RC = 12 − VC IC Substituting IC = αIE = 0.99 × 1 = 0.99 mA 1 mA results, for both designs, in RC = 12 − 8 1 = 4 k EXERCISE 7.34 For design 1 in Example 7.12, calculate the expected range of IE if the transistor used has β in the range of 50 to 150. Express the range of IE as a percentage of the nominal value (IE 1 mA) obtained for β = 100. Repeat for design 2. Ans. For design 1: 0.94 mA to 1.04 mA, a 10% range; for design 2: 0.984 mA to 0.995 mA, a 1.1% range. A Two-Power-Supply Version of the Classical Bias Arrangement A somewhat simpler bias arrangement is possible if two power supplies are available, as shown in Fig. 7.53. 10Although reducing RE restores IE to the design value of 1 mA, it does not solve the problem of the dependence of the value of IE on β. See Exercise 7.34. 7.4 Biasing 465 Figure 7.53 Biasing the BJT using two power supplies. Resistor RB is needed only if the signal is to be capacitively coupled to the base. Otherwise, the base can be connected directly to ground, or to a grounded signal source, resulting in almost total β-independence of the bias current. Writing a loop equation for the loop labeled L gives IE = RE VEE − VBE + RB/(β + 1) (7.146) This equation is identical to Eq. (7.143) except for VEE replacing VBB. Thus the two constraints of Eqs. (7.144) and (7.145) apply here as well. Note that if the transistor is to be used with the base grounded (i.e., in the common-base configuration), then RB can be eliminated altogether. On the other hand, if the input signal is to be coupled to the base, then RB is needed. We shall study complete circuits of the various BJT amplifier configurations in Section 7.5. Finally, observe that the circuit in Fig. 7.53 is the counterpart of the MOS circuit in Fig. 7.48(e). EXERCISE D7.35 The bias arrangement of Fig. 7.53 is to be used for a common-base amplifier. Design the circuit to establish a dc emitter current of 1 mA and provide the highest possible voltage gain while allowing for a signal swing at the collector of ±2 V. Use +10-V and –5-V power supplies. Ans. RB = 0; RE = 4.3 k ; RC = 8.4 k Biasing Using a Collector-to-Base Feedback Resistor In the BJT case, there is a counterpart to the MOSFET circuit of Fig. 7.50. Figure 7.54(a) shows such a simple but effective biasing arrangement that is suitable for common-emitter amplifiers. The circuit employs a resistor RB connected between the collector and the base. Resistor RB provides negative feedback, which helps to stabilize the bias point of the BJT. 466 Chapter 7 Transistor Amplifiers + VBE (a) (b) Figure 7.54 (a) A common-emitter transistor amplifier biased by a feedback resistor RB. (b) Analysis of the circuit in (a). Analysis of the circuit is shown in Fig. 7.54(b), from which we can write VCC = IE RC + IBRB + VBE = IE RC + β IE + 1 RB + VBE Thus the emitter bias current is given by IE = RC VCC − VBE + RB/(β + 1) (7.147) It is interesting to note that this equation is identical to Eq. (7.143), which governs the operation of the traditional bias circuit, except that VCC replaces VBB and RC replaces RE. It follows that to obtain a value of IE that is insensitive to variation of β, we select RB/(β + 1) RC. Note, however, that the value of RB determines the allowable negative signal swing at the collector since VCB = IBRB = IE RB β+ 1 (7.148) EXERCISE D7.36 Design the circuit of Fig. 7.54 to obtain a dc emitter current of 1 mA, maximum gain, and a ±2-V signal swing at the collector; that is, design for VCE = +2.3 V. Let VCC = 10 V and β = 100. Ans. RB = 162 k ; RC = 7.7 k . Note that if standard 5% resistor values are used (Appendix J), we select RB = 160 k and RC = 7.5 k . This results in IE = 1.02 mA and VC = +2.3 V. 7.5 Discrete-Circuit Amplifiers 467 7.5 Discrete-Circuit Amplifiers With our study of transistor amplifier basics complete, we now put everything together by presenting practical circuits for discrete-circuit amplifiers. These circuits, which utilize the amplifier configurations studied in Section 7.3 and the biasing methods of Section 7.4, can be assembled using off-the-shelf discrete transistors, resistors, and capacitors. Though practical and carefully selected to illustrate some important points, the circuits presented in this section should be regarded as examples of discrete-circuit transistor amplifiers. Indeed, there is a great variety of such circuits, a number of which are explored in the end-of-chapter problems. As mentioned earlier, the vast majority of discrete-circuit amplifiers utilize BJTs. This is reflected in this section where all the circuits presented except for one utilize BJTs. Of course, if desired, one can utilize MOSFETs in the same amplifier configurations presented here. Also, the MOSFET is the device of choice in the design of integrated-circuit (IC) amplifiers. We begin our study of IC amplifiers in Chapter 8. As will be seen shortly, the circuits presented in this section utilize large capacitors (in the μF range) to couple the signal source to the input of the amplifier, and to couple the amplifier output signal to a load resistance or to the input of another amplifier stage. As well, a large capacitor is employed to establish a signal ground at the desired terminal of the transistor (e.g., at the emitter of a CE amplifier). The use of capacitors for these purposes simplifies the design considerably: Since capacitors block dc, one is able to first carry out the dc bias design and then connect the signal source and load to the amplifier without disturbing the dc design. There amplifiers are therefore known as capacitively coupled amplifiers. 7.5.1 A Common-Source (CS) Amplifier As mentioned in Section 7.3, the common-source (CS) configuration is the most widely used of all MOSFET amplifier circuits. A common-source amplifier realized using the bias circuit of Fig. 7.48(c) is shown in Fig. 7.55(a). Observe that to establish a signal ground, or an ac ground as it is sometimes called, at the source, we have connected a large capacitor, CS, between the source and ground. This capacitor, usually in the microfarad range, is required to provide a very small impedance (ideally, zero impedance—i.e., in effect, a short circuit) at all signal frequencies of interest. In this way, the signal current passes through CS to ground and thus bypasses the resistance RS; hence, CS is called a bypass capacitor. Obviously, the lower the signal frequency, the less effective the bypass capacitor becomes. This issue will be studied in Section 10.1. For our purposes here we shall assume that CS is acting as a perfect short circuit and thus is establishing a zero signal voltage at the MOSFET source. To prevent disturbances to the dc bias current and voltages, the signal to be amplified, shown as voltage source vsig with an internal resistance Rsig, is connected to the gate through a large capacitor CC1. Capacitor CC1, known as a coupling capacitor, is required to act as a perfect short circuit at all signal frequencies of interest while blocking dc. Here again, we note that as the signal frequency is lowered, the impedance of CC1 (i.e., 1/jωCC1) will increase and its effectiveness as a coupling capacitor will be correspondingly reduced. This problem, too, will be considered in Section 10.1 in connection with the dependence of the amplifier 468 Chapter 7 Transistor Amplifiers Rsig vsig ϩ Ϫ VDD (0 V) RG1 CC1 0 RD CC2 vd ϩϩ 0V vgs = vi Ϫ vi RG2 RS CS Ϫ RG1 ϩ VG RL vo Ϫ RG2 Rin Ro (a) VDD ID RD VD VS ID RS (b) Figure 7.55 (a) A common-source amplifier using the classical biasing arrangement of Fig. 7.48(c). (b) Circuit for determining the bias point. (c) Equivalent circuit and analysis. operation on frequency. For our purposes here we shall assume that CC1 is acting as a perfect short circuit as far as the signal is concerned. The voltage signal resulting at the drain is coupled to the load resistance RL via another coupling capacitor CC2. We shall assume that CC2 acts as a perfect short circuit at all signal frequencies of interest and thus that the output voltage vo = vd. Note that RL can be either an actual load resistor, to which the amplifier is required to provide its output voltage signal, or it can be the input resistance of another amplifier stage in cases where more than one stage of amplification is needed. (We will study multistage amplifiers in Chapter 9). Since a capacitor behaves as an open circuit at dc, the circuit for performing the dc bias design and analysis is obtained by open-circuiting all capacitors. The resulting circuit is shown in Fig. 7.55(b) and can be designed as discussed in Section 7.4.1. 7.5 Discrete-Circuit Amplifiers 469 To determine the terminal characteristics of the CS amplifier of Fig. 7.55(a)—that is, its input resistance, voltage gain, and output resistance—we replace the MOSFET with its hybrid-π small-signal model, replace VDD with a signal ground, and replace all coupling and bypass capacitors with short circuits. The result is the circuit in Fig. 7.55(c). Analysis is straightforward and is shown on the figure, thus Rin = RG1 RG2 (7.149) which shows that to keep Rin high, large values should be used for RG1 and RG2, usually in the megohm range. The overall voltage gain Gv is Gv = − Rin Rin + Rsig gm(RD RL ro) (7.150) Observe that we have taken ro into account, simply because it is easy to do so. Its effect, however, is usually small (this is not the case for IC amplifiers, as will be explained in Chapter 8). Finally, to encourage the reader to do the small-signal analysis directly on the original circuit diagram, with the MOSFET model used implicitly, we show some of the analysis on the circuit of Fig. 7.55(a). EXERCISES D7.37 Design the bias circuit in Fig.7.55(b) for the CS amplifier of Fig. 7.55(a). Assume the MOSFET is specified to have Vt = 1 V, kn = 4 mA/V2, and VA = 100 V. Neglecting the Early effect, design for ID = 0.5 mA, VS = 3.5 V, and VD = 6 V using a power-supply VDD = 15 V. Specify the values of RS and RD. If a current of 2 μA is used in the voltage divider, specify the values of RG1 and RG2. Give the values of the MOSFET parameters gm and ro at the bias point. Ans. RS = 7 k ; RD = 18 k ; RG1 = 5 M ; RG2 = 2.5 M ; gm = 2 mA/V; ro = 200 k 7.38 For the CS amplifier of Fig. 7.55(a) use the design obtained in Exercise 7.37 to determine Rin, Ro, and the overall voltage gain Gv when Rsig = 100 k and RL = 20 k . Ans. 1.67 M ; 16.5 k ; −17.1 V/V D7.39 As discussed in Section 7.3, beneficial effects can be realized by having an unbypassed resistance Rs in the source lead of the CS amplifier. This can be implemented in the circuit of Fig. 7.55(a) by splitting the resistance RS into two resistances: Rs, which is left unbypassed, and (RS − Rs), across which the bypass capacitor CS is connected. Now, if in order to improve linearity of the amplifier in Exercises 7.37 and 7.38, vgs is to be reduced to half its value, what value should Rs have? What would the amplifier gain Gv become? Recall that when Rs is included it becomes difficult to include ro in the analysis, so neglect it. Ans. Rs = 500 ; Gv = −8.9 V/V 470 Chapter 7 Transistor Amplifiers 7.5.2 A Common-Emitter Amplifier The common-emitter (CE) amplifier is the most widely used of all BJT amplifier configurations. Figure 7.56(a) shows a CE amplifier utilizing the classical biasing arrangement of Fig. 7.48(c), the design of which was considered in Section 7.4. The CE circuit in Fig. 7.54(a) is the BJT counterpart of the CS amplifier of Fig. 7.55(a). It utilizes coupling capacitors CC1 and CC2 and bypass capacitor CE. Here we assume that these capacitors, while blocking dc, behave as perfect short circuits at all signal frequencies of interest. To determine the characteristic parameters of the CE amplifier, we replace the BJT with its hybrid-π model, replace VCC with a short circuit to ground, and replace the coupling and bypass capacitor with short circuits. The resulting small-signal equivalent circuit of the CE amplifier is shown in Fig. 7.56(b). The analysis is straightforward and is given in the Rsig vsig ϩ Ϫ VCC (0 V) RB1 CC1 RC CC2 vc ϩ ϩ vp = vi 0V – vi RB2 RE CE Ϫ ϩ RL vo Ϫ Rin Ro (a) Figure 7.56 (a) A common-emitter amplifier using the classical biasing arrangement of Fig. 7.52(a). (b) Equivalent circuit and analysis. 7.5 Discrete-Circuit Amplifiers 471 figure, thus Rin = RB1 RB2 rπ (7.151) which indicates that to keep Rin relatively high, RB1 and RB2 should be selected large (typically in the range of tens or hundreds of kilohms). This requirement conflicts with the need to keep RB1 and RB2 low so as to minimize the dependence of the dc current IC on the transistor β. We discussed this design trade-off in Section 7.4. The voltage gain Gv is given by Gv = − Rin Rin + Rsig gm(RC RL ro) (7.152) Note that we have taken ro into account because it is easy to do so. However, as already mentioned, the effect of this parameter on discrete-circuit amplifier performance is usually small. EXERCISES D7.40 Design the bias circuit of the CE amplifier of Fig. 7.56(a) to obtain IE = 0.5 mA and VC = +6 V. Design for a dc voltage at the base of 5 V and a current through RB2 of 50 μA. Let VCC = +15 V, β = 100, and VBE 0.7 V. Specify the values of RB1, RB2, RE, and RC. Also give the values of the BJT small-signal parameters gm, rπ , and ro at the bias point. (For the calculation of ro, let VA = 100 V.) Ans. RB1 = 182 k ; RB2 = 100 k ; RE = 8.6 k ; RC = 18 k ; gm = 20 mA/V, rπ = 5 k , ro = 200 k 7.41 For the amplifier designed in Exercise 7.40, find Rin, Ro, and Gv when Rsig = 10 k and RL = 20 k . Ans. Rin = 4.64 k ; Ro = 16.51 k ; Gv = −57.3 V/V 7.5.3 A Common-Emitter Amplifier with an Emitter Resistance Re As discussed in Section 7.3.4, it is beneficial to include a small resistance in the transistor emitter lead. This can be implemented in the circuit of Fig. 7.56(a) by splitting the emitter bias resistance RE into two components: an unbypassed resistance Re, and a resistance (RE − Re) across which the bypass capacitor CE is connected. The resulting circuit is shown in Fig. 7.57(a) and its small-signal model is shown in Fig. 7.57(b). In the latter we utilize the T model of the BJT because it results in much simpler analysis (recall that this is always the case when a resistance is connected in series with the emitter). Also note that we have not included ro, for doing so would complicate the analysis significantly. This burden would not be justified given that ro has little effect on the performance of discrete-circuit amplifiers. 472 Chapter 7 Transistor Amplifiers Rsig CC1 vsig ϩ Ϫ VCC RB1 RC CC2 Re RB2 (RE – Re) CE ϩ RL vo Ϫ (a) Figure 7.57 (a) A common-emitter amplifier with an unbiased emitter resistance Re. (b) The amplifier small-signal model and analysis. Analysis of the circuit in Fig. 7.57(b) is straightforward and is shown in the figure. Thus, Rin = RB1 RB2 (β + 1)(re + Re) = RB1 RB2 [rπ + (β + 1)Re] (7.153) 7.5 Discrete-Circuit Amplifiers 473 from which we note that including Re increases Rin because it increases the input resistance looking into the base by adding a component (β + 1)Re to rπ . The overall voltage gain Gv is Gv = − Rin Rin + Rsig × α Total resistance in collector Total resistance in emitter = − α Rin RC RL Rin + Rsig re + Re (7.154) EXERCISE 7.42 For the amplifier designed in Exercise 7.40 and analyzed in Exercise 7.41, let it be required to raise Rin to 10 k . What is the required value of Re, and what does the overall voltage gain Gv become? Ans. Re = 67.7 ; Gv = −39.8 V/V 7.5.4 A Common-Base (CB) Amplifier Figure 7.58(a) shows a CB amplifier designed using the biasing arrangement of Fig. 7.53. Note that the availability of two power supplies, VCC and −VEE, enables us to connect the base directly to ground, obviating the need for a large bypass capacitor to establish a signal ground at the base. The small-signal equivalent circuit of the CB amplifier is shown in Fig 7.58(b). As expected, we have utilized the T model of the BJT and have not included ro. Including ro would complicate the analysis significantly without making much difference to the results in the case of discrete-circuit amplifiers. From the circuit in Fig. 7.58(b) we find Rin = re RE re 1/gm which as expected can be very small, causing vi to be a small fraction of vsig, vi = vsig Rin Rin + Rsig Now, ie = − vi re and vo = − αie(RC RL) 474 Chapter 7 Transistor Amplifiers VCC (0 V) RC CC2 vo aie vo RL re Ro Rsig ii CC1 ie vsig vi RE Rin VEE (a) C aie B ie = –vi /re re Rsig ii E vo = – ie(RC ʈ RL) RC RL Ro = RC vsig vi RE Rin = re ʈ RE (b) Figure 7.58 (a) A common-base amplifier using the structure of Fig. 7.53 with RB omitted (since the base is grounded). (b) Equivalent circuit obtained by replacing the transistor with its T model. Thus, the overall voltage gain is given by Gv = α Rin Rin + Rsig RC RL re = Rin Rin + Rsig gm(RC RL ) (7.155) 7.5 Discrete-Circuit Amplifiers 475 EXERCISE D7.43 Design the CB amplifier of Fig. 7.58(a) to provide an input resistance Rin that matches the source resistance of a cable with a characteristic resistance of 50 . Assume that RE re. The available power supplies are ±5 V and RL = 8 k . Design for a dc collector voltage VC = +1 V. Specify the values of RC and RE. What overall voltage gain is obtained? If vsig is a sine wave with a peak amplitude of 10 mV, what is the peak amplitude of the output voltage? Let α 1. Ans. RC = 8 k ; RE = 8.6 k ; 40 V/V; 0.4 V 7.5.5 An Emitter Follower Figure 7.59(a) shows an emitter follower designed using the bias arrangement of Fig. 7.53 and two power supplies, VCC and −VEE. The bias resistance RB affects the input resistance of the follower and should be chosen as large as possible while limiting the dc voltage drop across it to a small fraction of VEE; otherwise the dependence of the bias current IC on β can become unacceptably large. Figure 7.59(b) shows the small-signal equivalent circuit of the emitter follower. Here, as expected, we have replaced the BJT with its T model and included ro (since this can be done very simply). The input resistance of the emitter follower can be seen to be Rin = RB Rib (7.156) where Rib, the input resistance looking into the base, can be obtained by using the resistance-reflection rule. Toward that end, note that ro appears in parallel with RE and RL (which is why it can be easily taken into account). Thus, Rib = (β + 1)[re + (RE ro RL)] (7.157) The overall voltage gain can be determined by tracking the signal transmission from source to load, and Thus, vi = vsig Rin Rin + Rsig vo = vi re RE ro + (RE RL ro RL) Gv ≡ vo vsig = Rin Rin + Rsig (RE ro RL) re + (RE ro RL) (7.158) (7.159) (7.160) 476 Chapter 7 Transistor Amplifiers Rsig vsig CC1 RB VCC CC2 RE RL vo Rin VEE Ro, Rout (a) C i Rsig ib (1 a)i b 1 B ai ro i re vsig E RB vi RE RL vo Rin Rib (b) Ro, Rout Figure 7.59 (a) An emitter-follower circuit. (b) Small-signal equivalent circuit of the emitter follower with the transistor replaced by its T model. Note that ro is included because it is easy to do so. Normally, its effect on performance is small. Finally, the output resistance Rout can be obtained by short-circuiting vsig and looking back into the output terminal, excluding RL, as Rout = ro RE re + RB β Rsig +1 (7.161) Note that we have used the inverse resistance-reflection rule, namely, dividing the total resistance in the base, (RB Rsig), by (β + 1). 7.5 Discrete-Circuit Amplifiers 477 EXERCISE D7.44 Design the emitter follower of Fig. 7.59(a) to operate at a dc emitter current IE = 1 mA. Allow a dc voltage drop across RB of 1 V. The available power supplies are ±5 V, β = 100, VBE = 0.7 V, and VA = 100 V. Specify the values required for RB and RE. Now if Rsig = 50 k and RL = 1 k , find Rin, vi/vsig, vo/vi, Gv , and Rout. (Note: In performing the bias design, neglect the Early effect.) Ans. RB = 100 k ; RE = 3.3 k ; 44.3 k ; 0.469 V/V; 0.968 V/V; 0.454 V/V; 320 7.5.6 The Amplifier Frequency Response Thus far, we have assumed that the gain of transistor amplifiers is constant independent of the frequency of the input signal. This would imply that transistor amplifiers have infinite bandwidth, which of course is not true. To illustrate, we show in Fig. 7.60 a sketch of the magnitude of the gain of a common-emitter or a CS amplifier such as those shown in Figs. 7.56 and 7.55, respectively, versus frequency. Observe that there is indeed a wide frequency range over which the gain remains almost constant. This obviously is the useful frequency range of Vo Vsig (dB) Low-frequency band • Gain falls off due to the effects of CC1, CC2, and CE Midband • All capacitances can be neglected 3 dB 20 log AM (dB) High-frequency band • Gain falls off due to the internal capacitive effects in the BJT and the MOSFET fL fH f (Hz) (log scale) Figure 7.60 Sketch of the magnitude of the gain of a CE (Fig. 7.56) or CS (Fig. 7.55) amplifier versus frequency. The graph delineates the three frequency bands relevant to frequency-response determination. 478 Chapter 7 Transistor Amplifiers operation for the particular amplifier. Thus far, we have been assuming that our amplifiers are operating in this frequency band, called the midband. Figure 7.60 indicates that at lower frequencies, the magnitude of amplifier gain falls off. This is because the coupling and bypass capacitors no longer have low impedances. Recall that we assumed that their impedances were small enough to act as short circuits. Although this can be true at midband frequencies, as the frequency of the input signal is lowered, the reactance 1/jωC of each of these capacitors becomes significant, and it can be shown that this results in the overall voltage gain of the amplifier decreasing. Figure 7.60 indicates also that the gain of the amplifier falls off at the high-frequency end. This is due to the internal capacitive effects in the BJT and the MOSFET. In Chapter 10 we shall study the internal capacitive effects of both transistor types and will augment their hybrid-π models with capacitances that model these effects. We will undertake a detailed study of the frequency response of transistor amplifiers in Chapter 10. For the time being, however, it is important for the reader to realize that for every transistor amplifier, there is a finite band over which the gain is almost constant. The boundaries of this useful frequency band, or midband, are the two frequencies fL and fH at which the gain drops by a certain number of decibels (usually 3 dB) below its value at midband. As indicated in Fig. 7.60, the amplifier bandwidth, or 3-dB bandwidth, is defined as the difference between the lower (fL) and upper or higher (fH) 3-dB frequencies: BW = fH − fL and since usually fL fH, BW fH A figure of merit for the amplifier is its gain–bandwidth product, defined as GB = |AM|BW where |AM| is the magnitude of the amplifier gain in the midband. It will be seen in Chapter 10 that in amplifier design it is usually possible to trade off gain for bandwidth. One way to accomplish this, for instance, is by including resistance Re in the emitter of the CE amplifier. Summary The essence of the use of the MOSFET (the BJT) as an amplifier is that when the transistor is operated in the active region, vGS controls iD (vBE controls iC) in the manner of a voltage-controlled current source. When the device is dc biased in the active region, and the signal vgs (vbe) is kept small, the operation becomes almost linear, with id = gmvgs (ic = gmvbe). The most fundamental parameter in characterizing the small-signal linear operation of a transistor is the transconductance gm. For a MOSFET, gm = μnCox(W/L)VOV = 2μnCox(W/L)ID = 2ID/VOV ; and for the BJT, gm = IC/VT . A systematic procedure for the analysis of a transistor amplifier circuit is presented in Table 7.1. Tables 7.2 and 7.3 present the small-signal models for the MOSFET and the BJT, respectively. When a resistance is connected in series with the source (or emitter), the T model is the most convenient to use. The three basic configurations of MOS and BJT amplifiers are presented in Fig. 7.33. Their characteristic parameter values are provided in Table 7.4 (for the MOS case) and in Table 7.5 (for the BJT case). The CS amplifier, which has (ideally) infinite input resistance and a reasonably high gain but a rather high output resistance and a limited high-frequency response (more on the latter point in Chapter 10), is used to obtain most of the gain in a cascade amplifier. Similar remarks apply to the CE amplifier, except that it has a relatively low input resistance (rπ = β/gm) arising from the finite base current of the BJT (finite β). Its voltage gain, however, can be larger than that of the CS amplifier because of the higher values of gm obtained with BJTs. Adding a resistance Rs in the source of a CS amplifier (a resistance Re in the emitter of a CE amplifier) can lead to beneficial effects including the following: raising the input resistance of the CE amplifier, increasing linearity, and extending the useful amplifier bandwidth, at the expense of reducing the gain, all by a factor equal to (1 + gmRs) [(1 + gmRe) for the BJT case]. Summary 479 The CG (CB) amplifier has a low input resistance and thus, used alone, it has limited and specialized applications. However, its excellent high-frequency response makes it attractive in combination with the CS (CE) amplifier (Chapters 8 and 10). The source follower has (ideally) infinite input resistance, a voltage gain lower than but close to unity, and a low output resistance. It is employed as a voltage buffer and as the output stage of a multistage amplifier. Similar remarks apply to the emitter follower except that its input resistance, though large, is finite. Specifically, the emitter follower multiplies the total resistance in the emitter by (β + 1) before presenting it to the signal source. The resistance-reflection rule is a powerful tool in the analysis of BJT amplifier circuits: All resistances in the emitter circuit including the emitter resistance re can be reflected to the base side by multiplying them by (β + 1). Conversely, we can reflect all resistances in the base circuit to the emitter side by dividing them by (β + 1). In the analysis and design of discrete-circuit amplifiers, it is rarely necessary to take the transistor output resistance ro into account. In some situations, however, ro can be easily taken into account; specifically in the CS (CE) amplifier and in the source (emitter) follower. In IC amplifiers, ro must always be taken into account. A key step in the design of transistor amplifiers is to bias the transistor to operate at an appropriate point in the active region. A good bias design ensures that the parameters of the operating point (ID, VOV , and VDS for the MOSFET; IC and VCE for the BJT) are predictable and stable and do not vary by large amounts when the transistor is replaced by another of the same type. The bias methods studied in this chapter are suited for discrete-circuit amplifiers only because they utilize large coupling and bypass capacitors. Discrete-circuit amplifiers predominantly employ BJTs. The opposite is true for IC amplifiers, where the device of choice is the MOSFET. PROBLEMS Computer Simulation Problems Problems identified by the Multisim/PSpice icon are intended to demonstrate the value of using SPICE simulation to verify hand analysis and design, and to investigate important issues such as allowable signal swing and amplifier nonlinear distortion. Instructions to assist in setting up PSpice and Multisim simulations for all the indicated problems can be found in the corresponding files on the website. Note that if a particular parameter value is not specified in the problem statement, you are to make a reasonable assumption. Q on the transfer characteristic. Also, find the value of ID and of the incremental gain Av at the bias point. (c) For the situation in (b), and disregarding the distortion caused by the MOSFET’s square-law characteristic, what is the largest amplitude of a sine-wave voltage signal that can be applied at the input while the transistor remains in saturation? What is the amplitude of the output voltage signal that results? What gain value does the combination of these amplitudes imply? By what percentage is this gain value different from the incremental gain value calculated above? Why is there a difference? Section 7.1: Basic Principles 7.1 For the MOS amplifier of Fig. 7.2(a) with VDD = 5 V, Vt = 0.5 V, kn = 10 mA/V2, and RD = 20 k , determine the coordinates of the active-region segment (AB) of the VTC [Fig. 7.2(b)]. D 7.2 For the MOS amplifier of Fig. 7.2(a) with VDD = 5 V and kn = 5 mA/V2, it is required to have the end point of the VTC, point B, at VDS = 0.5 V. What value of RD is required? If the transistor is replaced with another having twice the value of the transconductance parameter kn, what new value of RD is needed? D 7.3 It is required to bias the MOS amplifier of Fig. 7.3 at point Q for which VOV = 0.2 V and VDS = 1 V. Find the required value of RD when VDD = 5 V, Vt = 0.5 V, and kn = 10 mA/V2. Also specify the coordinates of the VTC end point B. What is the small-signal voltage gain of this amplifier? Assuming linear operation, what is the maximum allowable negative signal swing at the output? What is the corresponding peak input signal? 7.4 The MOS amplifier of Fig. 7.4(a), when operated with VDD = 2 V, is found to have a maximum small-signal voltage gain magnitude of 14 V/V. Find VOV and VDS for bias point Q at which a voltage gain of −12 V/V is obtained. 7.5 Consider the amplifier of Fig. 7.4(a) for the case VDD = 5 V, RD = 24 k , kn(W/L) = 1 mA/V2, and Vt = 1 V. (a) Find the coordinates of the two end points of the saturation-region segment of the amplifier transfer characteristic, that is, points A and B on the sketch of Fig. 7.4(b). (b) If the amplifier is biased to operate with an overdrive voltage VOV of 0.5 V, find the coordinates of the bias point 7.6 Various measurements are made on an NMOS amplifier for which the drain resistor RD is 20 k . First, dc measurements show the voltage across the drain resistor, VRD, to be 1.5 V and the gate-to-source bias voltage to be 0.7 V. Then, ac measurements with small signals show the voltage gain to be –10 V/V. What is the value of Vt for this transistor? If the process transconductance parameter kn is 200 μA/V2, what is the MOSFET’s W/L? *7.7 The expression for the incremental voltage gain Av given in Eq. (7.16) can be written in as Av 2 =− VDD − VDS VOV where VDS is the bias voltage at the drain. This expression indicates that for given values of VDD and VOV , the gain magnitude can be increased by biasing the transistor at a lower VDS. This, however, reduces the allowable output signal swing in the negative direction. Assuming linear operation around the bias point, show that the largest possible negative output signal peak vˆo that is achievable while the transistor remains saturated is vˆo = VDS − VOV 1+ 1 Av For VDD = 5 V and VOV = 0.5 V, provide a table of values for Av , vˆo, and the corresponding vˆi for VDS = 1 V, 1.5 V, 2 V, and 2.5 V. If kn(W/L) = 1 mA/V2, find ID and RD for the design for which VDS = 1 V. D *7.8 Design the MOS amplifier of Fig. 7.4(a) to obtain maximum gain while allowing for an output voltage swing of at least ±0.5 V. Let VDD = 5 V, and utilize an overdrive = Multisim/PSpice; * = difficult problem; ** = more difficult; *** = very challenging; D = design problem CHAPTER 7 PROBLEMS Problems 481 voltage of approximately 0.2 V. (a) Specify VDS at the bias point. (b) What is the gain achieved? What is the signal amplitude vˆgs that results in the 0.5-V signal amplitude at the output? (c) If the dc bias current in the drain is to be 100 μA, what value of RD is needed? (d) If kn = 200 μA/V2, what W/L ratio is required for the MOSFET? *7.9 Figure P7.9 shows an amplifier in which the load resistor RD has been replaced with another NMOS transistor Q2 connected as a two-terminal device. Note that because vDG of Q2 is zero, it will be operating in saturation at all times, even when vI = 0 and iD2 = iD1 = 0. Note also that the two transistors conduct equal drain currents. Using iD1 = iD2, show that for the range of vI over which Q1 is operating in saturation, that is, for Vt1 ≤ vI ≤ vO + Vt1 the output voltage will be given by vO = VDD − Vt + (W/L)1 (W/L)2 Vt − (W/L)1 (W/L)2 v I where we have assumed Vt1 = Vt2 = Vt. Thus the circuit functions as a linear amplifier, even for large input signals. For (W/L)1 = (50 μm/0.5 μm) and (W/L)2 = (5 μm/0.5 μm), find the voltage gain. VDD iD2 Q2 vO iD1 vI Q1 7.10 A BJT amplifier circuit such as that in Fig. 7.6 is operated with VCC = + 5 V and is biased at VCE = +1 V. Find the voltage gain, the maximum allowed output negative swing without the transistor entering saturation, and the corresponding maximum input signal permitted. 7.11 For the amplifier circuit in Fig. 7.6 with VCC = + 5 V and RC = 1 k , find VCE and the voltage gain at the following dc collector bias currents: 0.5 mA, 1 mA, 2.5 mA, 4 mA, and 4.5 mA. For each, give the maximum possible positive- and negative-output signal swing as determined by the need to keep the transistor in the acti