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G.722.2 AMR-WB

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INTERNATIONAL TELECOMMUNICATION UNION ITU-T TELECOMMUNICATION STANDARDIZATION SECTOR OF ITU G.722.2 (07/2003) SERIES G: TRANSMISSION SYSTEMS AND MEDIA, DIGITAL SYSTEMS AND NETWORKS Digital terminal equipments – Coding of analogue signals by methods other than PCM Wideband coding of speech at around 16 kbit/s using Adaptive Multi-Rate Wideband (AMR-WB) ITU-T Recommendation G.722.2 ITU-T G-SERIES RECOMMENDATIONS TRANSMISSION SYSTEMS AND MEDIA, DIGITAL SYSTEMS AND NETWORKS INTERNATIONAL TELEPHONE CONNECTIONS AND CIRCUITS GENERAL CHARACTERISTICS COMMON TO ALL ANALOGUE CARRIERTRANSMISSION SYSTEMS INDIVIDUAL CHARACTERISTICS OF INTERNATIONAL CARRIER TELEPHONE SYSTEMS ON METALLIC LINES GENERAL CHARACTERISTICS OF INTERNATIONAL CARRIER TELEPHONE SYSTEMS ON RADIO-RELAY OR SATELLITE LINKS AND INTERCONNECTION WITH METALLIC LINES COORDINATION OF RADIOTELEPHONY AND LINE TELEPHONY TESTING EQUIPMENTS TRANSMISSION MEDIA CHARACTERISTICS DIGITAL TERMINAL EQUIPMENTS General Coding of analogue signals by pulse code modulation Coding of analogue signals by methods other than PCM Principal characteristics of primary multiplex equipment Principal characteristics of second order multiplex equipment Principal characteristics of higher order multiplex equipment Principal characteristics of transcoder and digital multiplication equipment Operations, administration and maintenance features of transmission equipment Principal characteristics of multiplexing equipment for the synchronous digital hierarchy Other terminal equipment DIGITAL NETWORKS DIGITAL SECTIONS AND DIGITAL LINE SYSTEM QUALITY OF SERVICE AND PERFORMANCE – GENERIC AND USER-RELATED ASPECTS TRANSMISSION MEDIA CHARACTERISTICS DIGITAL TERMINAL EQUIPMENTS DIGITAL NETWORKS G.100–G.199 G.200–G.299 G.300–G.399 G.400–G.449 G.450–G.499 G.500–G.599 G.600–G.699 G.700–G.799 G.700–G.709 G.710–G.719 G.720–G.729 G.730–G.739 G.740–G.749 G.750–G.759 G.760–G.769 G.770–G.779 G.780–G.789 G.790–G.799 G.800–G.899 G.900–G.999 G.1000–G.1999 G.6000–G.6999 G.7000–G.7999 G.8000–G.8999 For further details, please refer to the list of ITU-T Recommendations. ITU-T Recommendation G.722.2 Wideband coding of speech at around 16 kbit/s using Adaptive Multi-Rate Wideband (AMR-WB) Summary This Recommendation describes the high quality Adaptive Multi-Rate Wideband (AMR-WB) encoder and decoder that is primarily intended for 7 kHz bandwidth speech signals. AMR-WB operates at a multitude of bit rates ranging from 6.6 kbit/s to 23.85 kbit/s. The bit rate may be changed at any 20-ms frame boundary. Annex C includes an integrated C source code software package which contains the implementation of the G.722.2 encoder and decoder and its Annexes A and B and Appendix I. A set of digital test vectors for developers is provided in Annex D. These test vectors are a verification tool providing an indication of success in implementing this codec. G.722.2 AMR-WB is the same codec as the 3GPP AMR-WB. The corresponding 3GPP specifications are TS 26.190 for the speech codec and TS 26.194 for the Voice Activity Detector. Source ITU-T Recommendation G.722.2 was approved on 29 July 2003 by ITU-T Study Group 16 (2001-2004) under the ITU-T Recommendation A.8 procedure. ITU-T Rec. G.722.2 (07/2003) i FOREWORD The International Telecommunication Union (ITU) is the United Nations specialized agency in the field of telecommunications. The ITU Telecommunication Standardization Sector (ITU-T) is a permanent organ of ITU. ITU-T is responsible for studying technical, operating and tariff questions and issuing Recommendations on them with a view to standardizing telecommunications on a worldwide basis. The World Telecommunication Standardization Assembly (WTSA), which meets every four years, establishes the topics for study by the ITU-T study groups which, in turn, produce Recommendations on these topics. The approval of ITU-T Recommendations is covered by the procedure laid down in WTSA Resolution 1. In some areas of information technology which fall within ITU-T's purview, the necessary standards are prepared on a collaborative basis with ISO and IEC. NOTE In this Recommendation, the expression "Administration" is used for conciseness to indicate both a telecommunication administration and a recognized operating agency. Compliance with this Recommendation is voluntary. However, the Recommendation may contain certain mandatory provisions (to ensure e.g. interoperability or applicability) and compliance with the Recommendation is achieved when all of these mandatory provisions are met. The words "shall" or some other obligatory language such as "must" and the negative equivalents are used to express requirements. The use of such words does not suggest that compliance with the Recommendation is required of any party. INTELLECTUAL PROPERTY RIGHTS ITU draws attention to the possibility that the practice or implementation of this Recommendation may involve the use of a claimed Intellectual Property Right. ITU takes no position concerning the evidence, validity or applicability of claimed Intellectual Property Rights, whether asserted by ITU members or others outside of the Recommendation development process. As of the date of approval of this Recommendation, ITU had received notice of intellectual property, protected by patents, which may be required to implement this Recommendation. However, implementors are cautioned that this may not represent the latest information and are therefore strongly urged to consult the TSB patent database.  ITU 2004 All rights reserved. No part of this publication may be reproduced, by any means whatsoever, without the prior written permission of ITU. ii ITU-T Rec. G.722.2 (07/2003) CONTENTS Page 1 Scope ............................................................................................................................ 1 2 Normative references.................................................................................................... 2 3 Definitions, symbols and abbreviations........................................................................ 2 3.1 Definitions ...................................................................................................... 2 3.2 Symbols .......................................................................................................... 3 3.3 Abbreviations ................................................................................................. 7 4 Outline description........................................................................................................ 8 4.1 Functional description of audio parts ............................................................. 8 4.2 Preparation of speech samples........................................................................ 8 4.3 Principles of the adaptive multi-rate wideband speech encoder .................... 8 4.4 Principles of the adaptive multi-rate speech decoder ..................................... 13 4.5 Sequence and subjective importance of encoded parameters......................... 14 5 Functional description of the encoder........................................................................... 14 5.1 Preprocessing.................................................................................................. 14 5.2 Linear prediction analysis and quantization ................................................... 15 5.3 Perceptual weighting ...................................................................................... 20 5.4 Open-loop pitch analysis ................................................................................ 20 5.5 Impulse response computation ....................................................................... 22 5.6 Target signal computation .............................................................................. 22 5.7 Adaptive codebook......................................................................................... 23 5.8 Algebraic codebook........................................................................................ 25 5.9 Quantization of the adaptive and fixed codebook gains................................. 36 5.10 Memory update............................................................................................... 37 5.11 High-band gain generation ............................................................................. 37 6 Functional description of the decoder........................................................................... 37 6.1 Decoding and speech synthesis ...................................................................... 38 6.2 High-pass filtering, up-scaling and interpolation ........................................... 40 6.3 High frequency band ...................................................................................... 41 7 Detailed bit allocation of the adaptive multi-rate wideband codec .............................. 43 8 Homing sequences ........................................................................................................ 51 8.1 Functional description .................................................................................... 51 8.2 Definitions ...................................................................................................... 52 8.3 Encoder homing.............................................................................................. 52 8.4 Decoder homing ............................................................................................. 52 ITU-T Rec. G.722.2 (07/2003) iii 9 Voice Activity Detector (VAD).................................................................................... 9.1 VAD symbols ................................................................................................. 9.2 Functional description .................................................................................... 10 Mandatory AMR-WB codec modes for the speech telephony service in 3GPP .......... BIBLIOGRAPHY .................................................................................................................... Page 53 53 55 62 64 iv ITU-T Rec. G.722.2 (07/2003) ITU-T Recommendation G.722.2 Wideband coding of speech at around 16 kbit/s using Adaptive Multi-Rate Wideband (AMR-WB) 1 Scope This Recommendation describes the detailed mapping from input blocks of 320 speech samples in 16-bit uniform PCM format to encoded blocks of 132, 177, 253, 285, 317, 365, 397, 461 and 477 bits and from encoded blocks of 132, 177, 253, 285, 317, 365, 397, 461 and 477 bits to output blocks of 320 reconstructed speech samples. The sampling rate is 16 000 samples/s leading to a bit rate for the encoded bit stream of 6.60, 8.85, 12.65, 14.25, 15.85, 18.25, 19.85, 23.05 or 23.85 kbit/s. The coding scheme for the multi-rate coding modes is the so-called Algebraic Code Excited Linear Prediction Coder, hereafter referred to as ACELP. The multi-rate wideband ACELP coder is referred to as AMR-WB. The codec described in this Recommendation also utilizes an integrated Voice Activity Detector (VAD). The foreseen applications for this Recommendation are the following: Voice over IP (VoIP) and Internet applications, Mobile Communications, PSTN applications, ISDN wideband telephony, ISDN videotelephony and video-conferencing. In addition to the algorithm specified in the main body of this Recommendation, Annexes A and B and Appendix I provide supplemental functionalities allowing interoperability with GSM and 3GPP wireless systems. These functionalities have originally been developed for these systems, but their use is not limited to mobile applications. Annexes D and E describe test vectors and frame structure respectively. These annexes may be implemented independently of this main body specification according to the different requirements of systems deploying the AMR-WB algorithm: • Annex A describes comfort noise aspects for use of the AMR-WB algorithm in source-controlled rate operation. The implementation of Annex A is essential for interoperability with GSM and 3GPP wireless systems. • Annex B describes the source-controlled rate operation for the AMR-WB algorithm. The implementation of Annex B is essential for interoperability with GSM and 3GPP wireless systems. • Annex D describes the digital test sequences, which are a verification tool providing an indication of success in implementing the AMR-WB codec. • Annex E describes the recommended frame structure for use with the different modes of operation for the AMR-WB algorithm. • Appendix I describes an example solution for error concealment of erroneous or lost AMR-WB frames. For better usability, the ANSI-C code with the low-level description of all these functionalities have been grouped into a single annex, Annex C. Should there be any discrepancy between the descriptions in any of the different parts of this Recommendation and the implementation of such descriptions in Annex C, the descriptions in Annex C shall prevail. In clause 8 a specific reset procedure, called codec homing, is described. This is a useful feature for bringing the codec into a known initial state (e.g., for testing purposes). Clause 9 specifies the Voice Activity Detector (VAD) used in this codec as well as in the source controlled rate operation (DTX) specified in Annex B. Clause 10 provides information on minimum requirements for support of AMR-WB modes in the 3GPP speech telephony service. ITU-T Rec. G.722.2 (07/2003) 1 2 Normative references The following ITU-T Recommendations and other references contain provisions which, through reference in this text, constitute provisions of this Recommendation. At the time of publication, the editions indicated were valid. All Recommendations and other references are subject to revision; users of this Recommendation are therefore encouraged to investigate the possibility of applying the most recent edition of the Recommendations and other references listed below. A list of the currently valid ITU-T Recommendations is regularly published. The reference to a document within this Recommendation does not give it, as a stand-alone document, the status of a Recommendation. [1] ITU-T Recommendation G.722 (1988), 7 kHz audio-coding within 64 kbit/s. [2] RFC 3267 (2002), Real-Time Transport Protocol (RTP) Payload Format and File Storage Format for the Adaptive Multi-Rate (AMR) and Adaptive Multi-Rate Wideband (AMR-WB) Audio Codecs. 3 Definitions, symbols and abbreviations 3.1 Definitions This Recommendation defines the following terms: 3.1.1 adaptive codebook: The adaptive codebook contains excitation vectors that are adapted for every subframe. The adaptive codebook is derived from the long-term filter state. The lag value can be viewed as an index into the adaptive codebook. 3.1.2 algebraic codebook: A fixed codebook where algebraic code is used to populate the excitation vectors (innovation vectors). The excitation contains a small number of non-zero pulses with predefined interlaced sets of potential positions. The amplitudes and positions of the pulses of the kth excitation codevector can be derived from its index k through a rule requiring no or minimal physical storage, in contrast with stochastic codebooks, whereby the path from the index to the associated codevector involves look-up tables. 3.1.3 anti-sparseness processing: An adaptive post-processing procedure applied to the fixed codebook vector in order to reduce perceptual artifacts from a sparse fixed codebook vector. 3.1.4 closed-loop pitch analysis: This is the adaptive codebook search, i.e. a process of estimating the pitch (lag) value from the weighted input speech and the long-term filter state. In the closed-loop search, the lag is searched using error minimization loop (analysis-by-synthesis). In the adaptive multi-rate wideband codec, closed-loop pitch search is performed for every subframe. 3.1.5 direct form coefficients: One of the formats for storing the short-term filter parameters. In the adaptive multi-rate wideband codec, all filters which are used to modify speech samples use direct form coefficients. 3.1.6 fixed codebook: The fixed codebook contains excitation vectors for speech synthesis filters. The contents of the codebook are non-adaptive (i.e., fixed). In the adaptive multi-rate wideband codec, the fixed codebook is implemented using an algebraic codebook. 3.1.7 fractional lags: A set of lag values having subsample resolution. In the adaptive multirate wideband codec a sub-sample resolution of 1/4th or 1/2nd of a sample is used. 3.1.8 frame: A time interval equal to 20 ms (320 samples at a 16-kHz sampling rate). 3.1.9 immittance spectral frequencies: (see Immittance Spectral Pair) 3.1.10 immittance spectral pair: Transformation of LPC parameters. Immittance Spectral Pairs are obtained by decomposing the inverse filter transfer function A(z) to a set of two transfer functions, one having even symmetry and the other having odd symmetry. The Immittance Spectral 2 ITU-T Rec. G.722.2 (07/2003) Pairs (also called Immittance Spectral Frequencies) are the roots of these polynomials on the z-unit circle. 3.1.11 integer lags: A set of lag values having whole sample resolution. 3.1.12 interpolating filter: An FIR filter used to produce an estimate of subsample resolution samples, given an input sampled with integer sample resolution. In this implementation, the interpolating filter has low-pass filter characteristics. Thus the adaptive codebook consists of the low-pass filtered interpolated past excitation. 3.1.13 inverse filter: This filter removes the short-term correlation from the speech signal. The filter models an inverse frequency response of the vocal tract. 3.1.14 lag: The long-term filter delay. This is typically the true pitch period, or its multiple or submultiple. 3.1.15 LP analysis window: For each frame, the short-term filter coefficients are computed using the high-pass filtered speech samples within the analysis window. In the adaptive multi-rate wideband codec, the length of the analysis window is always 384 samples. For all the modes, a single asymmetric window is used to generate a single set of LP coefficients. The 5-ms look-ahead is used in the analysis 3.1.16 LP coefficients: Linear Prediction (LP) coefficients (also referred to as Linear Predictive Coding (LPC) coefficients) is a generic descriptive term for the short-term filter coefficients. 3.1.17 mode: When used alone, refers to the source codec mode, i.e., to one of the source codecs employed in the AMR-WB codec 3.1.18 open-loop pitch search: A process of estimating the near optimal lag directly from the weighted speech input. This is done to simplify the pitch analysis and confine the closed-loop pitch search to a small number of lags around the open-loop estimated lags. In the adaptive multi-rate wideband codec, an open-loop pitch search is performed in every other subframe. 3.1.19 residual: The output signal resulting from an inverse filtering operation. 3.1.20 short-term synthesis filter: This filter introduces short-term correlation into the excitation signal which models the impulse response of the vocal tract. 3.1.21 perceptual weighting filter: This filter is employed in the analysis-by-synthesis search of the codebooks. The filter exploits the noise masking properties of the formants (vocal tract resonances) by weighting the error less in regions near the formant frequencies and more in regions away from them. 3.1.22 subframe: A time interval equal to 5 ms (80 samples at 16-kHz sampling rate). 3.1.23 vector quantization: A method of grouping several parameters into a vector and quantizing them simultaneously. 3.1.24 zero input response: The output of a filter due to past inputs, i.e., due to the present state of the filter, given that an input of zeros is applied. 3.1.25 zero state response: The output of a filter due to the present input, given that no past inputs have been applied, i.e., given that the state information in the filter is all zeros. 3.2 Symbols This Recommendation uses the following symbols: A(z) The inverse filter with unquantized coefficients Â(z) The inverse filter with quantized coefficients ITU-T Rec. G.722.2 (07/2003) 3 H(z) = 1 Â(z) ai aˆ i m W (z) γ1 T β Hh1(z) w(n) L1 L2 r(k) wlag(i) f0 fs r′(k) E(i) ki a (ji ) F1' (z ) F2' (z) F1(z) F2(z) qi q qˆ i(n) ωi Tm(x) f1(i), f2(i) f1′(i), f2′(i) f(i) The speech synthesis filter with quantized coefficients The unquantized linear prediction parameters (direct form coefficients) The quantified linear prediction parameters The order of the LP model The perceptual weighting filter (unquantized coefficients) The perceptual weighting factor The integer pitch lag nearest to the closed-loop fractional pitch lag of the subframe The adaptive prefilter coefficient (the quantified pitch gain) Preprocessing high-pass filter LP analysis window Length of the first part of the LP analysis window w(n) Length of the second part of the LP analysis window w(n) The autocorrelations of the windowed speech s′(n) Lag window for the autocorrelations (60-Hz bandwidth expansion) The bandwidth expansion in Hz The sampling frequency in Hz The modified (bandwidth expanded) autocorrelations The prediction error in the ith iteration of the Levinson algorithm The ith reflection coefficient The jth direct form coefficient in the ith iteration of the Levinson algorithm Symmetric ISF polynomial Antisymmetric ISF polynomial Polynomial F1'(z) Polynomial F2' (z) with roots z = 1 and z = –1 eliminated The immittance spectral pairs (ISPs) in the cosine domain An ISP vector in the cosine domain The quantified ISP vector at the ith subframe of the frame n The immittance spectral frequencies (ISFs) A mth order Chebyshev polynomial The coefficients of the polynomials F1(z) and F2(z) The coefficients of the polynomials F1'(z) and F2' (z) The coefficients of either F1(z) or F2(z) 4 ITU-T Rec. G.722.2 (07/2003) C(x) x λk fi f t = [ f1 f2 L f16 ] z(n) r(n) p(n) rˆ(n −1) rˆ ik di h(n) H(z)W(z) T1 s′(n) sw(n) sˆ(n) x(n) x2 (n), x t 2 resLP(n) c(n) v(n) y(n) = v(n)*h(n) yk(n) u(n) û′(n) Top tmin tmax R(k) R(k)t Ak Ck EDk Sum polynomial of the Chebyshev polynomials Cosine of angular frequency ω Recursion coefficients for the Chebyshev polynomial evaluation The immittance spectral frequencies (ISFs) in Hz The vector representation of the ISFs in Hz The mean-removed ISF vector at frame n The ISF prediction residual vector at frame n The predicted ISF vector at frame n The quantified residual vector at the past frame The quantified ISF subvector i at quantization index k The distance between the immittance spectral frequencies fi+1 and fi–1 The impulse response of the weighted synthesis filter The weighted synthesis filter The integer nearest to the fractional pitch lag of the previous (1st or 3rd) subframe The windowed speech signal The weighted speech signal Reconstructed speech signal The target signal for adaptive codebook search The target signal for algebraic codebook search The LP residual signal The fixed codebook vector The adaptive codebook vector The filtered adaptive codebook vector The past filtered excitation The excitation signal The gain-scaled emphasized excitation signal The best open-loop lag Minimum lag search value Maximum lag search value Correlation term to be maximized in the adaptive codebook search The interpolated value of R(k) for the integer delay k and fraction t Correlation term to be maximized in the algebraic codebook search at index k The correlation in the numerator of Ak at index k The energy in the denominator of Ak at index k ITU-T Rec. G.722.2 (07/2003) 5 d = Ht x2 H Φ = HtH d(n) φ(i, j) ck C mi ϑi Np ED resLTP(n) b(n) sb(n) d ′ (n) φ′(i, j) zt, z(n) E(n) E E~(n) [b1 b2 b3 b4] Rˆ(k ) EI R(n) EQ e(n) ew(n) η gc gc′ gˆ c gp gˆ p The correlation between the target signal x2(n) and the impulse response h(n), i.e., backward filtered target The lower triangular Toepliz convolution matrix with diagonal h(0) and lower diagonals h(1),..., h(63) The matrix of correlations of h(n) The elements of the vector d The elements of the symmetric matrix Φ The innovation vector The correlation in the numerator of Ak The position of the ith pulse The amplitude of the ith pulse The number of pulses in the fixed codebook excitation The energy in the denominator of Ak The normalized long-term prediction residual The signal used for presetting the signs in algebraic codebook search The sign signal for the algebraic codebook search Sign extended backward filtered target The modified elements of the matrix Φ, including sign information The fixed codebook vector convolved with h(n) The mean-removed innovation energy (in dB) The mean of the innovation energy The predicted energy The MA prediction coefficients The quantified prediction error at subframe k The mean innovation energy The prediction error of the fixed-codebook gain quantization The quantization error of the fixed-codebook gain quantization The states of the synthesis filter 1/ Aˆ(z) The perceptually weighted error of the analysis-by-synthesis search The gain scaling factor for the emphasized excitation The fixed-codebook gain The predicted fixed-codebook gain The quantified fixed codebook gain The adaptive codebook gain The quantified adaptive codebook gain 6 ITU-T Rec. G.722.2 (07/2003) γ gc = gc gc' γˆ gc γ sc A correction factor between the gain gc and the estimated one gc' The optimum value for γ gc Gain scaling factor 3.3 Abbreviations This Recommendation uses the following abbreviations: ACELP Algebraic Code-Excited Linear Prediction AGC Adaptive Gain Control AMR Adaptive Multi-rate AMR-WB Adaptive Multi-rate Wideband CELP Code Excited Linear Prediction FIR Finite Impulse Response GERAN GSM EDGE Radio Access Network GMSK Gaussian Modified Shift Keying ISF Immittance Spectral Frequency ISP Immittance Spectral Pair ISPP Interleaved Single-Pulse Permutation LP Linear Prediction LPC Linear Predictive Coding LTP Long-Term Predictor (or Long-Term Prediction) MA Moving Average O-TCH/F Octal TCH/Full rate: a 68.4-kbit/s gross bit-rate radio channel of GERAN8PSK O-TCH/H Octal-TCH/Half rate: a 32.4-kbit/s gross bit-rate radio channel of GERAN8PSK S-MSVQ Split-MultiStage Vector Quantization TCH Traffic CHannel (dedicated radio channel for speech or data) TCH/F TCH/Full rate: a 22.8-kbit/s gross bit-rate radio channel of GERAN-GMSK TCH/H TCH/Half rate: a 11.4-kbit/s gross bit-rate radio channel of GERAN-GMSK TFO Tandem Free Operation (Tandem Free is achieved using in-band signalling after call set-up) TrFO Transcoder Free Operation (Tandem Free is achieved using out-of-band signalling before call set-up; the transcoders are in principle not in the communication path) UTRAN UMTS Terrestrial Radio Access Network WB Wideband 8PSK 8 Phase Shift Keying ITU-T Rec. G.722.2 (07/2003) 7 4 Outline description This Recommendation is structured as follows: • Clause 4.1 contains a functional description of the audio parts including the A/D and D/A functions. • Clause 4.2 describes input format for the AMR-WB encoder and the output format for the AMR-WB decoder. • Clauses 4.3 and 4.4 present a simplified description of the principles of the AMR-WB codec encoding and decoding process respectively. • In clause 4.5, the sequence and subjective importance of encoded parameters are given. • Clause 5 presents the functional description of the AMR-WB codec encoding, whereas clause 6 describes the decoding procedures. • In clause 7, the detailed bit allocation of the AMR-WB codec is tabulated. • Clause 8 describes the homing operation. 4.1 Functional description of audio parts The analogue-to-digital and digital-to-analogue conversion will, in principle, comprise the following elements: 1) Analogue to uniform digital PCM [1] – microphone; – input level adjustment device; – input anti-aliasing filter; – sample-hold device sampling at 16 kHz; – analogue-to-uniform digital conversion to 14-bit representation. The uniform format shall be represented in two's complement. 2) Uniform digital PCM to analogue [1] – conversion from 14-bit/16-kHz uniform PCM to analogue; – a hold device; – reconstruction filter including x/sin(x) correction; – output level adjustment device; – earphone or loudspeaker. In the terminal equipment, the A/D function may be achieved – by direct conversion to 14-bit uniform PCM format; For the D/A operation, the inverse operations take place. 4.2 Preparation of speech samples The encoder is fed with data comprising samples with a resolution of 14 bits left justified in a 16-bit word. The decoder outputs data in the same format. Outside the speech codec, further processing must be applied if the traffic data occurs in a different representation. 4.3 Principles of the adaptive multi-rate wideband speech encoder The AMR-WB codec consists of nine source codecs with bit rates of 23.85, 23.05, 19.85, 18.25, 15.85, 14.25, 12.65, 8.85 and 6.60 kbit/s. The codec is based on the code excited linear prediction (CELP) coding model. The input signal is pre-emphasized using the filter Hpre-emph(z) = 1−µz−1. The CELP model is then applied to the pre- 8 ITU-T Rec. G.722.2 (07/2003) emphasized signal. A 16th-order linear prediction (LP), or short-term, synthesis filter is used which is given by: H(z) = 1 Â(z) = 1 ∑ 1+ m i=1 âi z −i (1) where âi,i = 1,…,m are the (quantized) linear prediction (LP) parameters, and m = 16 is the predictor order. The long-term, or pitch, synthesis filter is usually given by: 1 B(z ) = 1 − 1 gp z −T (2) where T is the pitch delay and gp is the pitch gain. The pitch synthesis filter is implemented using the so-called adaptive codebook approach. The CELP speech synthesis model is shown in Figure 1. In this model, the excitation signal at the input of the short-term LP synthesis filter is constructed by adding two excitation vectors from adaptive and fixed (innovative) codebooks. The speech is synthesized by feeding the two properly chosen vectors from these codebooks through the short-term synthesis filter. The optimum excitation sequence in a codebook is chosen using an analysis-by-synthesis search procedure in which the error between the original and synthesized speech is minimized according to a perceptually weighted distortion measure. adaptive codebook fixed codebook v(n) + u(n) c(n) s^(n) LP synthesis postprocessing Figure 1/G.722.2 – Simplified block diagram of the CELP synthesis model The perceptual weighting filter used in the analysis-by-synthesis search technique is given by: W (z) = A(z γ1)Hde-emph (z) (3) where A(z) is the unquantized LP filter, H de-emph = 1 1− 0.68z−1 , and γ1 = 0.92 is the perceptual weighting factor. The weighting filter uses the unquantized LP parameters. The encoder performs the analysis of the LPC, LTP and fixed codebook parameters at 12.8-kHz sampling rate. The coder operates on speech frames of 20 ms. At each frame, the speech signal is analysed to extract the parameters of the CELP model (LP filter coefficients, adaptive and fixed codebooks' indices and gains). In addition to these parameters, high-band gain indices are computed in 23.85-kbit/s mode. These parameters are encoded and transmitted. At the decoder, these parameters are decoded and speech is synthesized by filtering the reconstructed excitation signal through the LP synthesis filter. ITU-T Rec. G.722.2 (07/2003) 9 s(n) TO x(n) x2(n) h(n) A(Z) Â(Z) Figure 2/G.722.2 – Detailed block diagram of the ACELP encoder 10 ITU-T Rec. G.722.2 (07/2003) The signal flow at the encoder is shown in Figure 2. After decimation, high-pass and pre-emphasis filtering is performed. LP analysis is performed once per frame. The set of LP parameters is converted to immittance spectrum pairs (ISP) and vector quantized using split-multistage vector quantization (S-MSVQ). The speech frame is divided into 4 subframes of 5 ms each (64 samples at 12.8-kHz sampling rate). The adaptive and fixed codebook parameters are transmitted every subframe. The quantized and unquantized LP parameters or their interpolated versions are used depending on the subframe. An open-loop pitch lag is estimated in every other subframe or once per frame based on the perceptually weighted speech signal. The following operations are then repeated for each subframe: • The target signal x(n) is computed by filtering the LP residual through the weighted synthesis filter W(z)H(z) with the initial states of the filters having been updated by filtering the error between LP residual and excitation (this is equivalent to the common approach of subtracting the zero input response of the weighted synthesis filter from the weighted speech signal). • The impulse response h(n) of the weighted synthesis filter is computed. • Closed-loop pitch analysis is then performed (to find the pitch lag and gain), using the target x(n) and impulse response h(n), by searching around the open-loop pitch lag. Fractional pitch with 1/4th or 1/2nd of a sample resolution (depending on the mode and the pitch lag value) is used. The interpolating filter in fractional pitch search has low-pass frequency response. Further, there are two potential low-pass characteristics in the adaptive codebook and this information is encoded with 1 bit. • The target signal x(n) is updated by removing the adaptive codebook contribution (filtered adaptive codevector), and this new target, x2(n), is used in the fixed algebraic codebook search (to find the optimum innovation). • The gains of the adaptive and fixed codebook are vector quantified with 6 or 7 bits (with moving average (MA) prediction applied to the fixed codebook gain). • Finally, the filter memories are updated (using the determined excitation signal) for finding the target signal in the next subframe. The bit allocation of the AMR-WB codec modes is shown in Table 1. In each 20-ms speech frame, 132, 177, 253, 285, 317, 365, 397, 461 and 477 bits are produced, corresponding to a bit rate of 6.60, 8.85, 12.65, 14.25, 15.85, 18.25, 19.85, 23.05 or 23.85 kbit/s. More detailed bit allocation among the codec parameters is given in Tables 12a-12i. Note that the most significant bits (MSB) are always sent first. Table 1/G.722.2 – Bit allocation of the AMR-WB coding algorithm for 20-ms frame Mode 23.85 kbit/s Parameter VAD-flag ISP LTP-filtering Pitch delay Algebraic code Codebook gain HB-energy Total 1st 2nd 3rd 4th subframe subframe subframe subframe 1 1 1 1 9 6 9 6 88 88 88 88 7 7 7 7 4 4 4 4 Total per frame 1 46 4 30 352 28 16 477 ITU-T Rec. G.722.2 (07/2003) 11 Table 1/G.722.2 – Bit allocation of the AMR-WB coding algorithm for 20-ms frame Mode 23.05 kbit/s 19.85 kbit/s 18.25 kbit/s 15.85 kbit/s 14.25 kbit/s Parameter VAD-flag ISP LTP-filtering Pitch delay Algebraic code Gains Total VAD-flag ISP LTP-filtering Pitch delay Algebraic code Codebook gain Total VAD-flag ISP LTP-filtering Pitch delay Algebraic code Gains Total VAD-flag ISP LTP-filtering Pitch delay Algebraic code Gains Total VAD-flag ISP LTP-filtering Pitch delay Algebraic code Gains Total 1st 2nd 3rd 4th subframe subframe subframe subframe 1 1 1 1 9 6 9 6 88 88 88 88 7 7 7 7 1 1 1 1 9 6 9 6 72 72 72 72 7 7 7 7 1 1 1 1 9 6 9 6 64 64 64 64 7 7 7 7 1 1 1 1 9 6 9 6 52 52 52 52 7 7 7 7 1 1 1 1 9 6 9 6 44 44 44 44 7 7 7 7 Total per frame 1 46 4 30 352 28 461 1 46 4 30 288 28 397 1 46 4 30 256 28 365 1 46 4 30 208 28 317 1 46 4 30 176 28 285 12 ITU-T Rec. G.722.2 (07/2003) Table 1/G.722.2 – Bit allocation of the AMR-WB coding algorithm for 20-ms frame Mode 12.65 kbit/s 8.85 kbit/s 6.60 kbit/s Parameter VAD-flag ISP LTP-filtering Pitch delay Algebraic code Gains Total VAD-flag ISP Pitch delay Algebraic code Gains Total VAD-flag ISP Pitch delay Algebraic code Gains Total 1st 2nd 3rd 4th subframe subframe subframe subframe 1 1 1 1 9 6 9 6 36 36 36 36 7 7 7 7 8 5 8 5 20 20 20 20 6 6 6 6 8 5 5 5 12 12 12 12 6 6 6 6 Total per frame 1 46 4 30 144 28 253 1 46 26 80 24 177 1 36 23 48 24 132 4.4 Principles of the adaptive multi-rate speech decoder The signal flow at the decoder is shown in Figure 3. At the decoder, the transmitted indices are extracted from the received bitstream. The indices are decoded to obtain the coder parameters at each transmission frame. These parameters are the ISP vector, the 4 fractional pitch lags, the 4 LTP filtering parameters, the 4 innovative codevectors, and the 4 sets of vector quantized pitch and innovative gains. In 23.85-kbit/s mode, also high-band gain index is decoded. The ISP vector is converted to the LP filter coefficients and interpolated to obtain LP filters at each subframe. Then, at each 64-sample subframe: • the excitation is constructed by adding the adaptive and innovative codevectors scaled by their respective gains; • the 12.8-kHz speech is reconstructed by filtering the excitation through the LP synthesis filter; • the reconstructed speech is de-emphasized. Finally, the reconstructed speech is upsampled to 16 kHz and high-band speech signal is added to the frequency band from 6 kHz to 7 kHz. ITU-T Rec. G.722.2 (07/2003) 13 ^s(n) Figure 3/G.722.2 – Detailed block diagram of the ACELP decoder 4.5 Sequence and subjective importance of encoded parameters The encoder will produce the output information in a unique sequence and format, and the decoder must receive the same information in the same way. In Tables 12a-12i, the sequence of output bits and the bit allocation for each parameter is shown. The different parameters of the encoded speech and their individual bits have unequal importance with respect to subjective quality. 5 Functional description of the encoder In this clause, the different functions of the encoder represented in Figure 2 are described. 5.1 Preprocessing The encoder performs the analysis of the LPC, LTP and fixed codebook parameters at 12.8-kHz sampling rate. Therefore, the input signal has to be decimated from 16 kHz to 12.8 kHz. The decimation is performed by first upsampling by 4, then filtering the output through low-pass FIR filter Hdecim(z) that has the cut-off frequency at 6.4 kHz. Then, the signal is downsampled by 5. The filtering delay is compensated by adding zeros into the end of the input vector. After the decimation, two preprocessing functions are applied to the signal prior to the encoding process: high-pass filtering and pre-emphasizing (and signal down-scaling). (Down-scaling consists of dividing the input by a factor of 2 to reduce the possibility of overflows in the fixed-point implementation.) The high-pass filter serves as a precaution against undesired low frequency components. A filter at a cut-off frequency of 50 Hz is used, and it is given by: H h1 (z ) = 0.989502 −1.979004z−1 + 0.989502z−2 1−1.978882z−1 + 0.9799126z−2 (4) 14 ITU-T Rec. G.722.2 (07/2003) (Both down-scaling and high-pass filtering are combined by dividing the coefficients at the numerator of Hh1(z) by 2.) In the pre-emphasis, a first order high-pass filter is used to emphasize higher frequencies, and it is given by: Hpre-emph (z) = 1− 0.68z−1 (5) 5.2 Linear prediction analysis and quantization Short-term prediction, or LP, analysis is performed once per speech frame using the autocorrelation approach with 30-ms asymmetric windows. An overhead of 5 ms is used in the autocorrelation computation. The frame structure is depicted below. windowing frame n–1 windowing frame n frame n (4 × 5 ms) The autocorrelations of windowed speech are converted to the LP coefficients using the LevinsonDurbin algorithm. Then the LP coefficients are transformed to the ISP domain for quantization and interpolation purposes. The interpolated quantized and unquantized filters are converted back to the LP filter coefficients (to construct the synthesis and weighting filters at each subframe). 5.2.1 Windowing and autocorrelation computation LP analysis is performed once per frame using an asymmetric window. The window has its weight concentrated at the fourth subframe and it consists of two parts: the first part is a half of a Hamming window and the second part is a quarter of a Hamming-cosine function cycle. The window is given by: w(n) = 0.54 − 0.46 cos 2πn 2L1 −1 , n = 0,..., L1 −1 (6) = cos 2π(n − L1) 4L2 −1 , n = L1,.., L1 + L2 −1 where the values L1 = 256 and L2 = 128 are used. The autocorrelations of the windowed speech s′(n), n = 0,…,383 are computed by: 383 r(k ) = ∑ s′(n)s′(n − k ), k = 0,...,16 (7) n =k and a 60-Hz bandwidth expansion is used by lag windowing the autocorrelations using the window [3]: wlag (i ) =  exp−  1 2  2πf0i fs 2  ,  i = 1,...,16 (8) where f0 = 60 Hz is the bandwidth expansion and fs = 12 800 Hz is the sampling frequency. Further, r(0) is multiplied by the white noise correction factor 1.0001 which is equivalent to adding a noise floor at –40 dB. ITU-T Rec. G.722.2 (07/2003) 15 5.2.2 Levinson-Durbin algorithm The modified autocorrelations r′(0) = 1.0001r(0) and r′(k ) = r(k )wlag (k ), k = 1,...,16, are used to obtain the LP filter coefficients ak,k = 1,…,16 by solving the set of equations. 16 ∑ ak r′( i − k ) = −r ′(i), i = 1,...,16 (9) k =1 The set of equations in (9) is solved using the Levinson-Durbin algorithm [3]. This algorithm uses the following recursion: E(0) = r′(0) For i = 1 to 16 do [ ] ∑ ki = − r′(i) + ( i−1 j=1 aij−1r′ i − j) / E(i −1) ai(i) = ki For j = 1 to i −1 do a(ji) = a (i−1) j + ki ai(−i −j1) ( ) E(i) = 1− ki2 E(i −1) The final solution is given as aj = a (16) j , j = 1,...,16. The LP filter coefficients are converted to the ISP representation [5] for quantization and interpolation purposes. The conversions to the ISP domain and back to the LP filter domain are described in 5.2.3 and 5.2.4. 5.2.3 LP to ISP conversion The LP filter coefficients ak, k = 1,…,16, are converted to the ISP representation for quantization and interpolation purposes. For a 16th-order LP filter, the ISPs are defined as the roots of the sum and difference polynomials: ( ) f1′(z) = A(z)+ z−16 A z−1 (10) and: ( ) f2′(z) = A(z)− z−16 A z−1 (11) respectively. (The polynomials f1′(z) and f2′(z) are symmetric and antisymmetric, respectively.) It can be proven that all roots of these polynomials are on the unit circle and they alternate each other [6]. f2′(z) has two roots at z = 1 (ω = 0) and z = –1 (ω = π). To eliminate these two roots, we define the new polynomials: f1(z) = f1′(z) (12) and: ( ) f2(z) = f2′(z) 1− z−2 (13) ( ) Polynomials f1(z) and f2(z) have 8 and 7 conjugate roots on the unit circle e± jωi , respectively. Therefore, the polynomials can be written as: ( ) F1(z) = (1+ a[16]) ∏ 1− 2qi z−1 + z−2 (14) i=0,2,...,14 16 ITU-T Rec. G.722.2 (07/2003) and ( ) F2(z) = (1+ a[16]) ∏ 1− 2qi z−1 + z−2 (15) i=1,3,...,13 where qi = cos(ωi) with ωi being the immittance spectral frequencies (ISF) and a [16] is the last predictor coefficient. ISFs satisfy the ordering property 0 < ω1 < ω2 < ... < ω16 < π. We refer to qi as the ISPs in the cosine domain. Since both polynomials f1(z) and f2(z) are symmetric, only the first 8 and 7 coefficients of each polynomial, respectively, and the last predictor coefficient need to be computed. The coefficients of these polynomials are found by the recursive relations: for i = 0 to 7 f1(i) = ai + am−i f2 (i) = ai − am−i + f2 (i − 2) (16) f1(8) = 2a8 where m = 16 is the predictor order, and f2 (− 2) = f2(−1) = 0. The ISPs are found by evaluating the polynomials F1(z) and F2(z) at 100 points equally spaced between 0 and π and checking for sign changes. A sign change signifies the existence of a root and the sign change interval is then divided 4 times to better track the root. The Chebyshev polynomials are used to evaluate F1(z) and F2(z) [7]. In this method the roots are found directly in the cosine domain {qi}. The polynomials F1(z) and F2(z) evaluated at z = e jω can be written as: F1(ω) = 2e− j8ωC1(x) and F2 (ω) = 2e− j7ωC2 (x) (17) with: 7 6 C1(x) = ∑ f1(i)T8−i (x)+ f1 (8) 2 and C2(x) = ∑ f2(i)T8−i (x)+ f2 (7) 2 (18) i=0 i=0 where Tm = cos(mω) is the mth order Chebyshev polynomial, f(i) are the coefficients of either F1(z) or F2(z), computed using the equations in (16). The polynomial C(x) is evaluated at a certain value of x = cos(ω) using the recursive relation: for k = n f −1 down to 1 ( ) bk = 2xbk+1 − bk+2 + f n f − k end ( ) C(x) = xb1 − b2 + f n f 2 where nf = 8 in case of C1(x) and nf = 7 in case of C2(x), with initial values bnf = f(0) and bnf+1 = 0. The details of the Chebyshev polynomial evaluation method are found in [7]. 5.2.4 ISP to LP conversion Once the ISPs are quantized and interpolated, they are converted back to the LP coefficient domain {ak}. The conversion to the LP domain is done as follows. The coefficients of F1(z) and F2(z) are found by expanding Equations (14) and (15) knowing the quantized and interpolated ISPs qi, i = 0,…,m – 1, where m = 16. The following recursive relation is used to compute f1(z): ITU-T Rec. G.722.2 (07/2003) 17 for i = 2 to m 2 f1(i) = −2q2i−2 f1(i −1)+ 2 f1(i − 2) for j = i −1 down to 2 f1( j) = f1( j) – 2q2i–2 f1( j −1)+ f1( j − 2) end f1(1) = f1(1) – 2q2i–2 end with initial values f1(0) = 1 and f1(1) = –2q0. The coefficients f2(i) are computed similarly by replacing q2i–2 by q2i–1 and m/2 by m/2–1, and with initial conditions f2(0) = 1 and f2(1) = –2q1. Once the coefficients f1(z) and f2(z) are found, F2(z) is multiplied by 1–z–2, to obtain F2′(z); that is: f2′(i) = f2(i)− f2(i − 2), i = 2,..., m 2 −1 f1′(i) = f1(i) i = 0,..., m 2 (19) Then F1′(z) and F2′(z) are multiplied by 1+qm–1 and 1–qm–1, respectively. That is: f2′(i) = (1− qm−1) f2′(i), f1′(i) = (1+ qm−1) f1′(i) i = 0,..., m 2 −1 i = 0,..., m 2 Finally the LP coefficients are found by: ai = 0.5 f1′(i)+ 0.5 f2′(i), i = 1,..., m 2 −1 0.5 f1′(i)− 0.5 f2′(i), 0.5 f1′(m 2), i = m 2 +1,...m −1 i=m 2 (20) qm−1, i=m This is directly derived from the relation A(z) = (F1′(z)+ F2′(z)) 2 , and considering the fact that F1′(z) and F2′(z) are symmetric and antisymmetric polynomials, respectively. 5.2.5 Quantization of the ISP coefficients The LP filter coefficients are quantized using the ISP representation in the frequency domain; that is: fi = fs 2π arccos(qi ), = fs 4π arccos(qi ), i = 0,...14 i = 15 (21) where fi are the ISFs in Hz [0, 6400] and fs = 12800 is the sampling frequency. The ISF vector is given by ft = [f0,f1,…,f15], with t denoting transpose. A 1st-order MA prediction is applied, and the residual ISF vector is quantified using a combination of split vector quantization (SVQ) and multistage vector quantization (MSVQ). The prediction and quantization are performed as follows. Let z(n) denote the mean-removed ISF vector at frame n. The prediction residual vector r(n) is given by: r(n) = z(n)− p(n) (22) where p(n) is the predicted LSF vector at frame n. First-order MA prediction is used where: p(n) = 1 3 rˆ(n −1) (23) 18 ITU-T Rec. G.722.2 (07/2003) where rˆ(n −1) is the quantized residual vector at the past frame. The ISF residual vector r is quantized using split-multistage vector quantization S-MSVQ. The vector is split into 2 subvectors r1(n) and r2(n) of dimensions 9 and 7, respectively. The 2 subvectors are quantized in two stages. In the first stage r1(n) is quantized with 8 bits and r2(n) with 8 bits. For 8.85-, 12.65-, 14.25-, 15.85-, 18.25-, 19.85-, 23.05- or 23.85-kbit/s modes, the quantization error vectors ri(2) = r − rˆi ,i = 1,2 are split in the next stage into 3 and 2 subvectors, respectively. The subvectors are quantized using the bit rates described in Table 2. Table 2/G.722.2 – Quantization of ISP vector for the 8.85-, 12.65-, 14.25-, 15.85-, 18.25-, 19.85-, 23.05- or 23.85-kbit/s modes 3) Stage 2 (r(2)1,0 – 2) 6 bits 1) Unquantized 16-element-long ISP vector 2) Stage 1 ( r1 ) 8 bits 3) Stage 2 (r(2)1,3 – 5) 7 bits 3) Stage 2 (r(2)1,6 – 8) 7 bits 2) Stage 1 ( r2 ) 8 bits 3) Stage 2 (r(2) 2,0 – 2) 5 bits 3) Stage 2 (r(2) 2,3 – 6) 5 bits For 6.60-kbit/s mode, the quantization error vectors ri(2) = ri − rˆi ,i = 1,2 are split in the next stage into 2 and 1 subvectors, respectively. The subvectors are quantized using the bit rates described in Table 3. Table 3/G.722.2 – Quantization of ISP vector for the 6.60 kbit/s mode 1) Unquantized 16-element-long ISP vector 2) Stage 1 ( r1 ) 8 bits 3) Stage 2 (r(2)1,0 − 4) 7 bits 3) Stage 2 (r(2)1,5 – 8) 7 bits 2) Stage 1 ( r2 ) 8 bits 3) Stage 2 (r(2) 2,0 – 6) 6 bits A squared error ISP distortion measure is used in the quantization process. In general, for an input ISP or error residual subvector ri,i = 1,2 and a quantized vector at index k, rˆik , the quantization is performed by finding the index k which minimizes: [ ] n 2 ∑ E = ri − rˆik (24) i=m where m and n are the first and last elements of the subvector. 5.2.6 Interpolation of the ISPs The set of quantized (and unquantized) LP parameters is used for the fourth subframe whereas the first, second, and third subframes use a linear interpolation of the parameters in the adjacent frames. The interpolation is performed on the ISPs in the q domain. Let qˆ (n) 4 be the ISP vector at the 4th ITU-T Rec. G.722.2 (07/2003) 19 subframe of the frame, and qˆ (4n−1) the ISP vector at the 4th subframe of the past frame n–1. The interpolated ISP vectors at the 1st, 2nd, and 3rd subframes are given by: ˆˆˆqqq 13(((2nnn))) = = = 000...02545ˆq (4ˆˆqq n((44−nn1−−)11))+++000.8..49ˆq 56(4nˆˆq)q (4(4nn)) The same formula is used for interpolation of the unquantized ISPs. The interpolated ISP vectors are used to compute a different LP filter at each subframe (both quantized and unquantized) using the ISP to LP conversion method described in 5.2.4. 5.3 Perceptual weighting The traditional perceptual weighting filter W (z) = A(z γ1) A(z γ2 ) has inherent limitations in modelling the formant structure and the required spectral tilt concurrently. The spectral tilt is more pronounced in wideband signals due to the wide dynamic range between low and high frequencies. A solution to this problem is to introduce the pre-emphasis filter at the input, compute the LP filter A(z) based on the pre-emphasized speech s(n), and use a modified filter W(z) by fixing its denominator. This structure substantially decouples the formant weighting from the tilt. A weighting filter of the form W (z) = A(z γ1)Hde-emph (z) is used, where H de-emph = 1 1− β1z−1 and β1 = 0.68. Because A(z) is computed based on the pre-emphasized speech signal s(n), the tilt of the filter 1/A(z/γ1) is less pronounced compared to the case when A(z) is computed based on the original speech. Since de-emphasis is performed at the decoder end, it can be shown that the quantization error spectrum is shaped by a filter having a transfer function W–1(z)Hde-emph(z)=1/A(z/γ1). Thus, the spectrum of the quantization error is shaped by a filter whose transfer function is 1/A(z/γ1), with A(z) computed based on the pre-emphasized speech signal. 5.4 Open-loop pitch analysis Depending on the mode, open-loop pitch analysis is performed once per frame (each 20 ms) or twice per frame (each 10 ms) to find two estimates of the pitch lag in each frame. This is done in order to simplify the pitch analysis and confine the closed loop pitch search to a small number of lags around the open-loop estimated lags. Open-loop pitch estimation is based on the weighted speech signal sw(n) which is obtained by filtering the input speech signal through the weighting filter W (z) = A(z )γ1 Hde-emph (z) , where H de-emph = 1 1− β1z−1 and β1 = 0.68. That is, in a subframe of size L, the weighted speech is given by: 16 sw(n) = s(n)+ ∑ aiγ1is(n − i)+ β1sw(n −1), n = 0,..., L −1 (25) i=1 The open-loop pitch analysis is performed to a signal decimated by two. The decimated signal is obtained by filtering sw(n) through a fourth-order FIR filter Hdecim2(z) and then downsampling the output by two to obtain the signal swd(n). 20 ITU-T Rec. G.722.2 (07/2003) 5.4.1 6.60-kbit/s mode Open-loop pitch analysis is performed once per frame (every 20 ms) to find an estimate of the pitch lag in each frame. The open-loop pitch analysis is performed as follows. First, the correlation of decimated weighted speech is determined for each pitch lag value d by: 128 C(d ) = ∑ Swd (n)swd (n − d )w(d ), d = 17,...,115 (26) n=0 where w(d) is a weighting function. The estimated pitch lag is the delay that maximizes the weighted correlation function C(d). The weighting emphasises lower pitch lag values reducing the likelihood of selecting a multiple of the correct delay. The weighting function consists of two parts: a low pitch lag emphasis function, wl(d), and a previous frame lag neighbouring emphasis function, wn(d): w(d ) = wl (d )wn (d ) (27) The low pitch lag emphasis function is given by: wl (d ) = cw(d ) (28) where cw(d) is defined by a table in the fixed-point computational description. The previous frame lag neighbouring emphasis function depends on the pitch lag of previous speech frames: wn (d ) = ( cw   Told − d 1.0, + 98), v > 0.8 otherwise (29) where Told is the median filtered pitch lag of 5 previous voiced speech half-frames and v is an adaptive parameter. If the frame is classified as voiced by having the open-loop gain g > 0.6, then the v-value is set to 1.0 for the next frame. Otherwise, the v-value is updated by v = 0.9v. The open loop gain is given by: 127 ∑ swd (n)swd (n − dmax ) g = n=0 (30) 127 127 ∑ sw2d (n)∑ sw2d (n − dmax ) n=0 n=0 where dmax is the pitch delay that maximizes C(d). The median filter is updated only during voiced speech frames. The weighting depends on the reliability of the old pitch lags. If previous frames have contained unvoiced speech or silence, the weighting is attenuated through the parameter v. 5.4.2 8.85-, 12.65-, 14.25-, 15.85-, 18.25-, 19.85-, 23.05- and 23.85-kbit/s modes Open-loop pitch analysis is performed twice per frame (every 10 ms) to find two estimates of the pitch lag in each frame. The open-loop pitch analysis is performed as follows. First, the correlation of decimated weighted speech is determined for each pitch lag value d by: 63 C(d ) = ∑ swd (n)swd (n − d )w(d ), d = 17,K,115 (31) n=0 where w(d) is a weighting function. The estimated pitch lag is the delay that maximizes the weighted correlation function C(d). The weighting emphasises lower pitch-lag values reducing the likelihood of selecting a multiple of the correct delay. The weighting function consists of two parts: ITU-T Rec. G.722.2 (07/2003) 21 a low pitch lag emphasis function, wl(d), and a previous frame lag neighbouring emphasis function, wn(d): w(d ) = wl (d )wn (d ) (32) The low pitch lag emphasis function is given by: wl (d ) = cw(d ) (33) where cw(d) is defined by a table in the fixed point computational description. The previous frame lag neighbouring emphasis function depends on the pitch lag of previous speech frames: wn (d ) = ( cw   Told − d 1.0, + 98), v > 0.8, otherwise (34) where Told is the median filtered pitch lag of 5 previous voiced speech half-frames and v is an adaptive parameter. If the frame is classified as voiced by having the open-loop gain g > 0.6, then the v-value is set to 1.0 for the next frame. Otherwise, the v-value is updated by v = 0.9v. The open loop gain is given by: 63 ∑ swd (n)swd (n − dmax ) g = n=0 (35) 63 63 ∑ sw2d (n)∑ sw2d (n − dmax ) n=0 n=0 where dmax is the pitch delay that maximizes C(d). The median filter is updated only during voiced speech frames. The weighting depends on the reliability of the old pitch lags. If previous frames have contained unvoiced speech or silence, the weighting is attenuated through the parameter v. 5.5 Impulse response computation The impulse response, h(n), of the weighted synthesis filter H (z)W (z) = A(z γ1)Hde-emph (z) Aˆ(z) is computed each subframe. This impulse response is needed for the search of adaptive and fixed codebooks. The impulse response h(n) is computed by filtering the vector of coefficients of the filter A(z/γ1) extended by zeros through the two filters 1 Aˆ(z)and Hde-emph (z) . 5.6 Target signal computation The target signal for adaptive codebook search is usually computed by subtracting the zero-input response of the weighted synthesis filter H (z)W (z) = A(z γ1)Hde-emph (z) Aˆ(z) from the weighted speech signal sw(n). This is performed on a subframe basis. An equivalent procedure for computing the target signal, which is used in this codec, is the filtering of the LP residual signal r(n) through the combination of synthesis filter 1 Aˆ(z) and the weighting filter A(z γ1)Hde-emph (z) . After determining the excitation for the subframe, the initial states of these filters are updated by filtering the difference between the LP residual and excitation. The memory update of these filters is explained in 5.10. The residual signal r(n) which is needed for finding the target vector is also used in the adaptive codebook search to extend the past excitation buffer. This simplifies the adaptive codebook search procedure for delays less than the subframe size of 64 as will be explained in 5.7. The LP residual is given by: 22 ITU-T Rec. G.722.2 (07/2003) 16 r(n) = s(n) = ∑ aˆis(n − i), n = 0,...,63 (36) i=1 5.7 Adaptive codebook Adaptive codebook search is performed on a subframe basis. It consists of performing closed loop pitch search, and then computing the adaptive codevector by interpolating the past excitation at the selected fractional pitch lag. The adaptive codebook parameters (or pitch parameters) are the delay and gain of the pitch filter. In the search stage, the excitation is extended by the LP residual to simplify the closed-loop search. In 12.65-, 14.25-, 15.85-, 18.25-, 19.85-, 23.05- or 23.85-kbit/s modes, in the first and third subframes, a fractional pitch delay is used with resolutions 1/4 in the range [34, 127¾], resolutions 1/2 in the range [128, 159 1 ], and integers only in the range [160, 231]. For the second and fourth 2 subframes, a pitch resolution of 1/4 is always used in the range [T1–8, T1+7¾], where T1 is nearest integer to the fractional pitch lag of the previous (1st or 3rd) subframe. In 8.85-kbit/s mode, in the first and third subframes, a fractional pitch delay is used with resolutions 1/2 in the range [34, 91½], and integers only in the range [92, 231]. For the second and fourth subframes, a pitch resolution of 1/2 is always used in the range [T1–8, T1+7½], where T1 is nearest integer to the fractional pitch lag of the previous (1st or 3rd) subframe. In 6.60-kbit/s mode, in the first subframe, a fractional pitch delay is used with resolutions 1/2 in the range [34,91½], and integers only in the range [92, 231]. For the second, third and fourth subframes, a pitch resolution of 1/2 is always used in the range [T1–8, T1+7½], where T1 is nearest integer to the fractional pitch lag of the first subframe. Closed-loop pitch analysis is performed around the open-loop pitch estimates on a subframe basis. In 8.85-, 12.65-, 14.25-, 15.85-, 18.25-, 19.85-, 23.05- or 23.85-kbit/s modes, in the first (and third) subframe the range Top ± 7, bounded by 34...231, is searched. In 6.60-kbit/s mode, in the first subframe the range Top ± 7, bounded by 34...231, is searched. For all the modes, for the other subframes, closed-loop pitch analysis is performed around the integer pitch selected in the previous subframe, as described above. In 12.65-, 14.25-, 15.85-, 18.25-, 19.85-, 23.05- or 23.85-kbit/s modes, the pitch delay is encoded with 9 bits in the first and third subframes and the relative delay of the other subframes is encoded with 6 bits. In 8.85-kbit/s mode, the pitch delay is encoded with 8 bits in the first and third subframes and the relative delay of the other subframes is encoded with 5 bits. In 6.60 kbit/s mode, the pitch delay is encoded with 8 bits in the first subframe and the relative delay of the other subframes is encoded with 5 bits. The closed loop pitch search is performed by minimizing the mean-square weighted error between the original and synthesized speech. This is achieved by maximizing the term: Tk = ∑63 n=0 x(n)yk (n) ∑63 n=0 yk (n )yk (n) (37) where x(n) is the target signal and yk(n) is the past filtered excitation at delay k (past excitation convolved with h(n)). Note that the search range is limited around the open-loop pitch as explained earlier. The convolution yk(n) is computed for the first delay in the searched range, and for the other delays, it is updated using the recursive relation: yk (n) = yk−1(n −1)+ u(− k )h(n) (38) ITU-T Rec. G.722.2 (07/2003) 23 where u(n), n = –(231+17),…,63, is the excitation buffer. Note that in search stage, the samples u(n),n = 0,...,63, are not known, and they are needed for pitch delays less than 64. To simplify the search, the LP residual is copied to u(n) in order to make the relation in Equation (38) valid for all delays. Once the optimum integer pitch delay is determined, the fractions from – ¾ to ¾ with a step of ¼ around that integer are tested. The fractional pitch search is performed by interpolating the normalized correlation in Equation (37) and searching for its maximum. Once the fractional pitch lag is determined, v'(n) is computed by interpolating the past excitation signal u(n) at the given phase (fraction). (The interpolation is performed using two FIR filters (Hamming windowed sinc functions); one for interpolating the term in Equation (37) with the sinc truncated at ±17 and the other for interpolating the past excitation with the sinc truncated at ±63.) The filters have their cutoff frequency (–3 dB) at 6000 Hz in the oversampled domain, which means that the interpolation filters exhibit low-pass frequency response. Thus, even when the pitch delay is an integer value, the adaptive codebook excitation consists of a low-pass filtered version of the past excitation at the given delay and not a direct copy thereof. Further, for delays smaller than the subframe size, the adaptive codebook excitation is completed based on the low-pass filtered interpolated past excitation and not by repeating the past excitation. In order to enhance the pitch prediction performance in wideband signals, a frequency-dependent pitch predictor is used. This is important in wideband signals since the periodicity does not necessarily extend over the whole spectrum. In this algorithm, there are two signal paths associated to respective sets of pitch codebook parameters, wherein each signal path comprises a pitch prediction error calculating device for calculating a pitch prediction error of a pitch codevector from a pitch codebook search device. One of these two paths comprises a low-pass filter for filtering the pitch codevector and the pitch prediction error is calculated for these two signal paths. The signal path having the lowest calculated pitch prediction error is selected, along with the associated pitch gain. The low-pass filter used in the second path is in the form BLP(z) = 0.18z + 0.64 + 0.18z–1. Note that 1 bit is used to encode the chosen path. Thus, for 8.85-, 12.65-, 14.25-, 15.85-, 18.25-, 19.85-, 23.05- or 23.85-kbit/s modes, there are two possibilities to generate the adaptive codebook v(n), v(n) = v′(n) in the first path, or 1 v(n) = ∑ bLP (i +1)v′(n + i) in the second path, where bLP = [0.18,0.64,0.18]. The path which results i=1 in minimum energy of the target signal x2(n) defined in Equation (40) is selected for the filtered adaptive codebook vector. For 6.60-kbit/s mode, v(n) is always: 1 v(n) = ∑bLP (i +1)v′(n + i) i=1 The adaptive codebook gain is then found by: gp = ∑63 n=0 x(n)y(n) , ∑63 n=0 y(n)y(n ) bounded by 0 ≤ g p ≤ 1.2 (39) where y(n) = v(n)*h(n) is the filtered adaptive codebook vector (zero-state response of H(z)W(z) to vi(n)). To insure stability, the adaptive codebook gain gp is bounded by 0.95, if the adaptive codebook gains of the previous subframes have been small and the LP filters of the previous subframes have been close to being unstable. 24 ITU-T Rec. G.722.2 (07/2003) 5.8 Algebraic codebook 5.8.1 Codebook structure The codebook structure is based on interleaved single-pulse permutation (ISPP) design. The 64 positions in the codevector are divided into 4 tracks of interleaved positions, with 16 positions in each track. The different codebooks at the different rates are constructed by placing a certain number of signed pulses in the tracks (from 1 to 6 pulses per track). The codebook index, or codeword, represents the pulse positions and signs in each track. Thus, no codebook storage is needed, since the excitation vector at the decoder can be constructed through the information contained in the index itself (no look-up tables). An important feature of the used codebook is that it is a dynamic codebook consisting of an algebraic codebook followed by an adaptive prefilter F(z) which enhances special spectral components in order to improve the synthesis speech quality. A prefilter relevant to wideband signals is used whereby F(z) consists of two parts: a periodicity enhancement part 1/(1 – 0.85z–T) and a tilt part (1 – β1 z–1), where T is the integer part of the pitch lag and β1 is related to the voicing of the previous subframe and is bounded by [0.0,0.5]. The codebook search is performed in the algebraic domain by combining the filter F(z) with the weighed synthesis filter prior to the codebook search. Thus, the impulse response h(n) must be modified to include the prefilter F(z). That is, h(n) ← h(n)* f (n). The codebook structures of different bit rates are given below. 5.8.1.1 23.85- and 23.05-kbit/s mode In this codebook, the innovation vector contains 24 non-zero pulses. All pulses can have the amplitudes +1 or –1. The 64 positions in a subframe are divided into 4 tracks, where each track contains six pulses, as shown in Table 4. Table 4/G.722.2 – Potential positions of individual pulses in the algebraic codebook, 23.85 and 23.05 kbit/s Track 1 2 3 4 Pulse i0, i4, i8, i12, i16, i20 i1, i5, i9, i13, i17, i21 i2, i6, i10, i14, i18, i22 i3, i7, i11, i15, i19, i23 Positions 0, 4, 8, 12, 16, 20, 24, 28, 32, 36, 40, 44, 48, 52, 56, 60 1, 5, 9, 13, 17, 21, 25, 29, 33, 37, 41, 45, 49, 53, 57, 61 2, 6, 10, 14, 18, 22, 26, 30, 34, 38, 42, 46, 50, 54, 58, 62 3, 7, 11, 15, 19, 23, 27, 31, 35, 39, 43, 47, 51, 55, 59, 63 The six pulses in one track are encoded with 22 bits. This gives a total of 88 bits (22+22+22+22) for the algebraic code. 5.8.1.2 19.85-kbit/s mode In this codebook, the innovation vector contains 18 non-zero pulses. All pulses can have the amplitudes +1 or –1. The 64 positions in a subframe are divided into 4 tracks, where each of the first two tracks contains five pulses and each of the other tracks contains four pulses, as shown in Table 5. ITU-T Rec. G.722.2 (07/2003) 25 Table 5/G.722.2 – Potential positions of individual pulses in the algebraic codebook, 19.85 kbit/s Track 1 2 3 4 Pulse i0, i4, i8, i12, i16 i1, i5, i9, i13, i17 i2, i6, i10, i14 i3, i7, i11, i15 Positions 0, 4, 8, 12, 16, 20, 24, 28, 32, 36, 40, 44, 48, 52, 56, 60 1, 5, 9, 13, 17, 21, 25, 29, 33, 37, 41, 45, 49, 53, 57, 61 2, 6, 10, 14, 18, 22, 26, 30, 34, 38, 42, 46, 50, 54, 58, 62 3, 7, 11, 15, 19, 23, 27, 31, 35, 39, 43, 47, 51, 55, 59, 63 The five pulses in one track are encoded with 20 bits. The four pulses in one track are encoded with 16 bits. This gives a total of 72 bits (20+20+16+16) for the algebraic code. 5.8.1.3 18.25-kbit/s mode In this codebook, the innovation vector contains 16 non-zero pulses. All pulses can have the amplitudes +1 or –1. The 64 positions in a subframe are divided into 4 tracks, where each track contains four pulses, as shown in Table 6. Table 6/G.722.2 – Potential positions of individual pulses in the algebraic codebook, 18.25 kbit/s Track 1 2 3 4 Pulse i0, i4, i8, i12 i1, i5, i9, i13 i2, i6, i10, i14 i3, i7, i11, i15 Positions 0, 4, 8, 12, 16, 20, 24, 28, 32, 36, 40, 44, 48, 52, 56, 60 1, 5, 9, 13, 17, 21, 25, 29, 33, 37, 41, 45, 49, 53, 57, 61 2, 6, 10, 14, 18, 22, 26, 30, 34, 38, 42, 46, 50, 54, 58, 62 3, 7, 11, 15, 19, 23, 27, 31, 35, 39, 43, 47, 51, 55, 59, 63 The four pulses in one track are encoded with 16 bits. This gives a total of 64 bits (16+16+16+16) for the algebraic code. 5.8.1.4 15.85-kbit/s mode In this codebook, the innovation vector contains 12 non-zero pulses. All pulses can have the amplitudes +1 or –1. The 64 positions in a subframe are divided into 4 tracks, where each track contains three pulses, as shown in Table 7. Table 7/G.722.2 – Potential positions of individual pulses in the algebraic codebook, 15.85 kbit/s Track 1 2 3 4 Pulse i0, i4, i8 i1, i5, i9 i2, i6, i10 i3, i7, i11 Positions 0, 4, 8, 12, 16, 20, 24, 28, 32, 36, 40, 44, 48, 52, 56, 60 1, 5, 9, 13, 17, 21, 25, 29, 33, 37, 41, 45, 49, 53, 57, 61 2, 6, 10, 14, 18, 22, 26, 30, 34, 38, 42, 46, 50, 54, 58, 62 3, 7, 11, 15, 19, 23, 27, 31, 35, 39, 43, 47, 51, 55, 59, 63 The three pulses in one track are encoded with 13 bits. This gives a total of 52 bits (13+13+13+13) for the algebraic code. 26 ITU-T Rec. G.722.2 (07/2003) 5.8.1.5 14.25-kbit/s mode In this codebook, the innovation vector contains 10 non-zero pulses. All pulses can have the amplitudes +1 or –1. The 64 positions in a subframe are divided into 4 tracks, where each track contains two or three pulses, as shown in Table 8. Table 8/G.722.2 – Potential positions of individual pulses in the algebraic codebook, 14.25 kbit/s Track 1 2 3 4 Pulse i0, i4, i8 i1, i5, i9 i2, i6 i3, i7 Positions 0, 4, 8, 12, 16, 20, 24, 28, 32, 36, 40, 44, 48, 52, 56, 60 1, 5, 9, 13, 17, 21, 25, 29, 33, 37, 41, 45, 49, 53, 57, 61 2, 6, 10, 14, 18, 22, 26, 30, 34, 38, 42, 46, 50, 54, 58, 62 3, 7, 11, 15, 19, 23, 27, 31, 35, 39, 43, 47, 51, 55, 59, 63 Each two pulse positions in one track are encoded with 8 bits (4 bits for the position of every pulse), and the sign of the first pulse in the track is encoded with 1 bit. The three pulses in one track are encoded with 13 bits. This gives a total of 44 bits (13+13+9+9) for the algebraic code. 5.8.1.6 12.65-kbit/s mode In this codebook, the innovation vector contains 8 non-zero pulses. All pulses can have the amplitudes +1 or –1. The 64 positions in a subframe are divided into 4 tracks, where each track contains two pulses, as shown in Table 9. Table 9/G.722.2 – Potential positions of individual pulses in the algebraic codebook, 12.65 kbit/s Track 1 2 3 4 Pulse i0, i4 i1, i5 i2, i6 i3, i7 Positions 0, 4, 8, 12, 16, 20, 24, 28, 32, 36, 40, 44, 48, 52, 56, 60 1, 5, 9, 13, 17, 21, 25, 29, 33, 37, 41, 45, 49, 53, 57, 61 2, 6, 10, 14, 18, 22, 26, 30, 34, 38, 42, 46, 50, 54, 58, 62 3, 7, 11, 15, 19, 23, 27, 31, 35, 39, 43, 47, 51, 55, 59, 63 Each two pulse positions in one track are encoded with 8 bits (total of 32 bits, 4 bits for the position of every pulse), and the sign of the first pulse in the track is encoded with 1 bit (total of 4 bits). This gives a total of 36 bits for the algebraic code. 5.8.1.7 8.85-kbit/s mode In this codebook, the innovation vector contains 4 non-zero pulses. All pulses can have the amplitudes +1 or –1. The 64 positions in a subframe are divided into 4 tracks, where each track contains one pulse, as shown in Table 10. ITU-T Rec. G.722.2 (07/2003) 27 Table 10/G.722.2 – Potential positions of individual pulses in the algebraic codebook, 8.85 kbit/s Track 1 2 3 4 Pulse i0 i1 i2 i3 Positions 0, 4, 8, 12, 16, 20, 24, 28, 32, 36, 40, 44, 48, 52, 56, 60 1, 5, 9, 13, 17, 21, 25, 29, 33, 37, 41, 45, 49, 53, 57, 61 2, 6, 10, 14, 18, 22, 26, 30, 34, 38, 42, 46, 50, 54, 58, 62 3, 7, 11, 15, 19, 23, 27, 31, 35, 39, 43, 47, 51, 55, 59, 63 Each pulse position in one track is encoded with 4 bits and the sign of the pulse in the track is encoded with 1 bit. This gives a total of 20 bits for the algebraic code. 5.8.1.8 6.60-kbit/s mode In this codebook, the innovation vector contains 2 non-zero pulses. All pulses can have the amplitudes +1 or –1. The 64 positions in a subframe are divided into 2 tracks, where each track contains one pulse, as shown in Table 11. Table 11/G.722.2 – Potential positions of individual pulses in the algebraic codebook, 6.60 kbit/s Track 1 2 Pulse i0 i1 Positions 0, 2, 4, 6, 8, 10, 12, 14, 16, 18, 20, 22, 24, 26, 28, 30, 32, 34, 36, 38, 40, 42, 44, 46, 48, 50, 52, 54, 56, 58, 60, 62 1, 3, 5, 7, 9, 11, 13, 15, 17, 19, 21, 23, 25, 27, 29, 31, 33, 35, 37, 39, 41, 43, 45, 47, 49, 51, 53, 55, 57, 59, 61, 63 Each pulse position in one track is encoded with 5 bits and the sign of the pulse in the track is encoded with 1 bit. This gives a total of 12 bits for the algebraic code. 5.8.2 Pulse indexing In 5.8.1, the number of bits needed to encode a number of pulses in a track was given. In this clause, the procedures used for encoding from 1 to 6 pulses per track will be described. The description will be given for the case of 4 tracks per subframe, with 16 positions per track and pulse spacing of 4 (which is the case for all modes except the 6.6 kbit/s mode). Encoding 1 signed pulse per track The pulse position index is encoded with 4 bits and the sign index with 1 bit. The position index is given by the pulse position in the subframe divided by the pulse spacing (integer division). The division remainder gives the track index. For example, a pulse at position 31 has a position index of 31/4 = 7 and it belongs to the track with index 3 (4th track). The sign index here is set to 0 for positive signs and 1 for negative signs. The index of the signed pulse is given by: I1p = p + s × 2M where p is the position index, s is the sign index, and M = 4 is the number of bits per track. 28 ITU-T Rec. G.722.2 (07/2003) Encoding 2 signed pulses per track In case of two pulses per track of K = 2M potential positions (here M = 4), each pulse needs 1 bit for the sign and M bits for the position, which gives a total of 2M + 2 bits. However, some redundancy exists due to the unimportance of the pulse ordering. For example, placing the first pulse at position p and the second pulse at position q is equivalent to placing the first pulse at position q and the second pulse at position p. One bit can be saved by encoding only one sign and deducing the second sign from the ordering of the positions in the index. Here the index is given by: I2 p = p1 + p0 × 2M + s × 22M where s is the sign index of the pulse at position index p0. If the two signs are equal then the smaller position is set to p0 and the larger position is set to p1. On the other hand, if the two signs are not equal, then the larger position is set to p0 and the smaller position is set to p1. At the decoder, the sign of the pulse at position p0 is readily available. The second sign is deduced from the pulse ordering. If p0 is larger than p1 then the sign of the pulse at position p1 is opposite to that at position p0. If this is not the case, then the two signs are set equal. Encoding 3 signed pulses per track In case of three pulses per track, similar logic can be used as in the case of two pulses. For a track with 2M positions, 3M + 1 bits are needed instead of 3M + 3 bits. A simple way of indexing the pulses is to divide the track positions in two sections (or halves) and identify a section that contains at least two pulses. The number of positions in the section is K/2 = 2M/2 = 2M–1, which can be represented with M–1 bits. The two pulses in the section containing at least two pulses are encoded with the procedure for encoding 2 signed pulses which requires 2(M–1) + 1 bits and the remaining pulse which can be anywhere in the track (in either section) is encoded with the M + 1 bits. Finally, the index of the section that contains the two pulses is encoded with 1 bit. Thus the total number of required bits is 2(M–1) + 1 + M + 1 + 1 = 3M + 1. A simple way of checking if two pulses are positioned in the same section is done by checking whether the most significant bits (MSB) of their position indices are equal or not. Note that a MSB of 0 means that the position belongs to the lower half of the track (0-7) and MSB of 1 means it belongs to the upper half (8-15). If the two pulses belong to the upper half, they need to be shifted to the range (0-7) before encoding them using 2×3+1 bits. This can be done by masking the M–1 least significant bits (LSB) with a mask consisting of M–1 ones (which corresponds to the number 7 in this case). The index of the 3 signed pulses is given by: I3 p = I2 p + k × 22M −1 + I1P × 22M where I2p is the index of the two pulses in the same section, k is the section index (0 or 1), and I1p is the index of the third pulse in the track. Encoding 4 signed pulses per track The 4 signed pulses in a track of length K = 2M can be encoded using 4M bits. Similar to the case of 3 pulses, the K positions in the track are divided into 2 sections (two halves) where each section contains K/2 = 8 positions. Here we denote the sections as Section A with positions 0 to K/2–1 and Section B with positions K/2 to K–1. Each section can contain from 0 to 4 pulses. The table below shows the 5 cases representing the possible number of pulses in each section: ITU-T Rec. G.722.2 (07/2003) 29 Case 0 1 2 3 4 Pulses in Section A 0 1 2 3 4 Pulses in Section B 4 3 2 1 0 Bits needed 4M–3 4M–2 4M–2 4M–2 4M–3 In cases 0 or 4, the 4 pulses in a section of length K/2 = 2M–1 can be encoded using 4(M–1)+1 = 4M−3 bits (this will be explained later on). In cases 1 or 3, the 1 pulse in a section of length K/2 = 2M–1 can be encoded with M–1+1 = M bits and the 3 pulses in the other section can be encoded with 3(M–1)+1 = 3M–2 bits. This gives a total of M+3M–2 = 4M–2 bits. In case 2, the pulses in a section of length K/2 = 2M–1 can be encoded with 2(M–1)+1 = 2M–1 bits. Thus for both sections, 2(2M–1) = 4M–2 bits are required. Now the case index can be encoded with 2 bits (4 possible cases) assuming cases 0 and 4 are combined. Then for cases 1, 2, or 3, the number of needed bits is 4M–2. This gives a total of 4M−2+ 2 = 4M bits. For cases 0 or 4, one bit is needed for identifying either case, and 4M–3 bits are needed for encoding the 4 pulses in the section. Adding the 2 bits needed for the general case, this gives a total of 1+4M–3+2= 4M bits. The index of the 4 signed pulses is given by: I4 p = IAB + k × 24M −2 where k is the case index (2 bits), and IAB is the index of the pulses in both sections for each individual case. For cases 0 and 4, IAB is given by: IAB_0,4 = I4 p_section + j × 24M −3 where j is a 1-bit index identifying the section with 4 pulses and I4p_section is the index of the 4 pulses in that section (which requires 4M–3 bits). For case 1, IAB is given by: IAB_1 = I3 p_B + I1p_A × 23(M −1)+1 where I3p_B is the index of the 3 pulses in Section B (3(M–1)+1 bits) and I1p_A is the index of the pulse in Section A ((M–1)+1 bits). For case 2, IAB is given by: IAB_2 = I2 p_B + I2 p_A × 22(M −1)+1 where I2p_B is the index of the 2 pulses in Section B (2(M–1)+1 bits) and I2p_A is the index of the two pulses in Section A (2(M–1)+1 bits). Finally, for case 3, IAB is given by: IAB_3 = I1p_B + I3 p_A × 2M where I1p_B is the index of the pulse in Section B ((M–1)+1 bits) and I3p_A is the index of the 3 pulses in Section A (3(M–1)+1 bits). 30 ITU-T Rec. G.722.2 (07/2003) For cases 0 and 4, it was mentioned that the 4 pulses in one section are encoded using 4(M–1)+1 bits. This is done by further dividing the section into 2 subsections of length K/4 = 2M–2 (= 4 in this case); identifying a subsection that contains at least 2 pulses; coding the 2 pulses in that subsection using 2(M–2)+1 = 2M–3 bits; coding the index of the subsection that contains at least 2 pulses using 1 bit; and coding the remaining 2 pulses, assuming that they can be anywhere in the section, using 2(M–1)+1 = 2M–1 bits. This gives a total of (2M–3)+(1)+(2M–1) = 4M–3 bits. Encoding 5 signed pulses per track The 5 signed pulses in a track of length K = 2M can be encoded using 5M bits. Similar to the case of 4 pulses, the K positions in the track are divided into 2 sections A and B. Each section can contain from 0 to 5 pulses. A simple approach to encode the 5 pulses is to identify a section that contains at least 3 pulses and to encode the 3 pulses in that section using 3(M–1)+1 = 3M–2 bits, and to encode the remaining 2 pulses in the whole track using 2M+1 bits. This gives 5M–1 bits. An extra bit is needed to identify the section that contains at least 3 pulses. Thus a total of 5M bits are needed to encode the 5 signed pulses. The index of the 5 signed pulses is given by: I5 p = I2 p + I3 p × 22M + k × 25M −1 where k is the index of the section that contains at least 3 pulses, I3p is the index of the 3 pulses in that section (3(M–1)+1 bits), and I2p is the index of the remaining 2 pulses in the track (2M+1 bits). Encoding 6 signed pulses per track The 6 signed pulses in a track of length K = 2M are encoded using 6M–2 bits. Similar to the case of 5 pulses, the K positions in the track are divided into 2 sections A and B. Each section can contain from 0 to 6 pulses. The table below shows the 7 cases representing the possible number of pulses in each section: Case 0 1 2 3 4 5 6 Pulses in Section A 0 1 2 3 4 5 6 Pulses in Section B 6 5 4 3 2 1 0 Bits needed 6M–5 6M–5 6M–5 6M–4 6M–5 6M–5 6M–5 Note that cases 0 and 6 are similar except that the 6 pulses are in different section. Similarly, cases 1 and 5 as well as cases 2 and 4 differ only in the section that contains more pulses. Therefore these cases can be coupled and an extra bit can be assigned to identify the section that contains more pulses. Since these cases initially need 6M–5 bits, the coupled cases need 6M–4 bits taking into account the section bit. Thus, we have now 4 states of coupled cases, that is (0,6), (1,5), (2,4), and (3),with 2 extra bits needed for the state. This gives a total of 6M–4+2 = 6M–2 bits for the 6 signed pulses. In cases 0 and 6, 1 bit is needed to identify the section which contains 6 pulses. Five pulses in that section are encoded using 5(M–1) bits (since the pulses are confined to that section), and the remaining pulse is encoded using (M–1)+1 bits. Thus a total of 1+5(M–1)+M = 6M–4 bits are needed for this coupled case. Extra 2 bits are needed to encode the state of the coupled case, giving a total of 6M–2 bits. For this coupled case, the index of the 6 pulses is given by: ITU-T Rec. G.722.2 (07/2003) 31 I6 p = I1p + I5 p × 2M + j × 26M –5 + k × 26M −4 where k is the index of the coupled case (2 bits), j is the index of the section containing 6 pulses (1 bit), I5p is the index of 5 pulses in that section (5(M–1) bits), and I1p is the index of the remaining pulse in that section ((M–1)+1 bits). In cases 1 and 5, 1 bit is needed to identify the section which contains 5 pulses. The 5 pulses in that section are encoded using 5(M–1) bits and the pulse in the other section is encoded using (M–1)+1 bits. For this coupled case, the index of the 6 pulses is given by: I6 p = I1p + I5 p × 2M + j × 26M –5 + k × 26M −4 where k is the index of the coupled case (2 bits), j is the index of the section containing 5 pulses (1 bit), I5p is the index of the 5 pulses in that section (5(M–1) bits), and I1p is the index of the pulse in the other section ((M–1)+1 bits). In cases 2 or 4, 1 bit is needed to identify the section which contains 4 pulses. The 4 pulses in that section are encoded using 4(M–1) bits and the 2 pulses in the other section are encoded using 2(M–1)+1 bits. For this coupled case, the index of the 6 pulses is given by: I6 p = I2 p + I4 p × 22(M –1)+1 + j × 26M –5 + k × 26M −4 where k is the index of the coupled case (2 bits), j is the index of the section containing 4 pulses (1 bit), I4p is the index of 4 pulses in that section (4(M–1) bits), and I2p is the index of the 2 pulses in the other section (2(M–1)+1 bits). In case 3, the 3 pulses in each section are encoded using 3(M–1)+1 bits in each section. For this case, the index of the 6 pulses is given by: I6 p = I3 pB + I3 pA × 23(M –1)+1 + k × 26M –4 (w3h(Mere_1k)+i1s the index of bits), and I3pA the coupled case (2 bits), I3pB is the is the index of the 3 pulses in Section A index of 3 pulses (3(M–1)+1 bits). in Section B 5.8.3 Codebook search The algebraic codebook is searched by minimizing the mean square error between the weighted input speech and the weighted synthesis speech. The target signal used in the closed-loop pitch search is updated by subtracting the adaptive codebook contribution. That is: x2(n) = x(n)− g p y(n), n = 0,...,63 (40) where y(n) = v(n)*h(n) is the filtered adaptive codebook vector and gp is the unquantized adaptive codebook gain. The matrix H is defined as the lower triangular Toeplitz convolution matrix with diagonal h(0) and lower diagonals h(1),…,h(63), and d = Htx2 is the correlation between the target signal x2(n) and the impulse response h(n) (also known as the backward filtered target vector), and Φ = HtH is the matrix of correlations of h(n). The elements of the vector d are computed by: 63 d (n) = ∑ x2(i)h(i − n), n = 0,K,63 (41) i=n 32 ITU-T Rec. G.722.2 (07/2003) and the elements of the symmetric matrix Φ are computed by: 63 φ(i, j) = ∑ h(n − i)h(n − j), i = 0,...,63 j = i,...,63 (42) n= j If ck is the algebraic codevector at index k, then the algebraic codebook is searched by maximizing the search criterion: ( ) ( ) Qk = xt2Hck 2 ckt Ht Hck = dtck 2 ctk Φck = (Rk )2 Ek (43) The vector d and the matrix Φ are usually computed prior to the codebook search. The algebraic structure of the codebooks allows for very fast search procedures since the innovation vector ck contains only a few non-zero pulses. The correlation in the numerator of Equation (43) is given by: Np –1 C = ∑ aid(mi ) (44) i=0 where mi is the position of the ith pulse, ai is its amplitude, and Np is the number of pulses. The energy in the denominator of Equation (43) is given by: ∑ ∑ ∑ ( ) Np–1 Np–2 Np–1 E = φ(mi , mi )+ 2 aia jφ mi , m j (45) i=0 i=0 j=i+1 To simplify the search procedure, the pulse amplitudes are predetermined based on a certain reference signal b(n). In this so-called signal-selected pulse amplitude approach, the sign of a pulse at position i is set equal to the sign of the reference signal at that position. Here, the reference signal b(n) is given by: b(n) = Ed Er rLTP (n)+ αd (n) (46) where Ed = dtd is the energy of the signal d(n) and Er = rLt TPrLTP is the energy of the signal rLTP(n) which is the residual signal after long-term prediction. The scaling factor α controls the amount of dependence of the reference signal on d(n), and it is lowered as the bit rate is increased. Here α = 2 for 6.6 and 8.85 modes; α = 1 for 12.65, 14.25, and 15.85 modes; α = 0.8 for 18.25 mode; α = 0.75 for 19.85 mode; and α = 0.5 for 23.05 and 23.85 modes. To simplify the search, the signal d(n) and matrix Φ are modified to incorporate the pre-selected signs. Let sb(n) denote the vector containing the signs of b(n). The modified signal d′(n) is given by: d′(n) = sb (n)d(n) n = 0,..., N −1 and the modified autocorrelation matrix Φ′ is given by: φ′(i, j) = sb (i)sb ( j)φ(i, j), i = 0,..., N –1; j = i,..., N –1 The correlation at the numerator of the search criterion Qk is now given by: Np –1 R = ∑ d′(i) i=0 ITU-T Rec. G.722.2 (07/2003) 33 and the energy at the denominator of the search criterion Qk is given by: ∑ ∑ ∑ ( ) Np–1 Np–2 Np–1 E = φ′(mi , mi )+ 2 φ′ mi , m j i=0 i=0 j=i+1 The goal of the search now is to determine the codevector with the best set of Np pulse positions assuming amplitudes of the pulses have been selected as described above. The basic selection criterion is the maximization of the above-mentioned ratio Qk. In order to reduce the search complexity, a fast search procedure known as depth-first tree search procedure is used, whereby the pulse positions are determined Nm pulses at a time. More precisely, the Np available pulses are partitioned into M non-empty subsets of Nm pulses respectively such that N1+N2...+Nm...+NM = Np. A particular choice of positions for the first J = N1+N2...+Nm–1 pulses considered is called a level-m path or a path of length J. The basic criterion for a path of J pulse positions is the ratio Qk(J) when only the J relevant pulses are considered. The search begins with subset #1 and proceeds with subsequent subsets according to a tree structure whereby subset m is searched at the mth level of the tree. The purpose of the search at level 1 is to consider the N1 pulses of subset #1 and their valid positions in order to determine one, or a number of, candidate path(s) of length N1 which are the tree nodes at level l. The path at each terminating node of level m–1 is extended to length N1+N2...+Nm at level m by considering Nm new pulses and their valid positions. One, or a number of, candidate extended path(s) are determined to constitute level-m nodes. The best codevector corresponds to that path of length Np which maximizes the criterion Qk(Np) with respect to all level-M nodes. A special form of the depth-first tree search procedure is used here, in which two pulses are searched at a time, that is, Nm = 2, and these 2 pulses belong to two consecutive tracks. Further, instead of assuming that the matrix Φ is precomputed and stored, which requires a memory of N × N words (64 × 64 = 4k words), a memory-efficient approach is used which reduces the memory requirement. In this approach, the search procedure is performed in such a way that only a part of the needed elements of the correlation matrix are precomputed and stored. This part corresponds to the correlations of the impulse response corresponding to potential pulse positions in consecutive tracks, as well as the correlations corresponding to φ(j,j), j = 0,…,N–1 (that is the elements of the main diagonal of matrix Φ). In order to reduce complexity, while testing possible combinations of two pulses, a limited number of potential positions of the first pulse are tested. Further, in case of a large number of pulses, some pulses in the higher levels of the search tree are fixed. In order to guess intelligently which potential pulse positions are considered for the first pulse, or in order to fix some pulse positions, a "pulse-position likelihood-estimate vector" b is used, which is based on speech-related signals. The pth component b(p) of this estimate vector b characterizes the probability of a pulse occupying position p (p = 0, 1,... N–1) in the best codevector we are searching for. Here the estimate vector b is the same vector used for preselecting the amplitudes and given in Equation (46). The search procedures for all bit rate modes are similar. Two pulses are searched at a time, and these two pulses always correspond to consecutive tracks. That is, the two searched pulses are in tracks T0–T1, T1–T2, T2–T3, or T3–T0. Before searching the positions, the sign of a pulse at a potential position n sets the sign of b(n) at that position. Then, the modified signal d'(n) is computed as described above by including the predetermined signs. For the first 2 pulses (1st tree level), the correlation at the numerator of the search criterion is given by: R = d′(m0 )+ d′(m1) 34 ITU-T Rec. G.722.2 (07/2003) and the energy at the denominator of the search criterion Qk is given by: E = φ′(m0, m0 )+ φ′(m1, m1)+ 2φ′(m0, m1) where the correlations φ′(mi , m j ) has been modified to include the preselected signs at positions mi and mj. For subsequent levels, the numerator and denominator are updated by adding the contribution of two new pulses. Assuming that two new pulses at a certain tree level with positions mk and mk+1 from two consecutive tracks are searched, then the updated value of R is given by: R = R + d '(mk )+ d′(mk+1) (47) and the updated energy is given by: E = E + φ′(mk , mk )+ φ′(mk+1, mk+1)+ 2φ′(mk , mk+1)+ 2Rhv (mk )+ 2Rhv (mk+1) (48) where Rhv(m) is the correlation between the impulse response h(n) and a vector vh(n) containing the addition of delayed versions of impulse response at the previously determined positions. That is: k −1 vh (n) = ∑ h(n − mi ) i=0 and: N −1 Rhv (m) = ∑ h(n)vh (n − m) n=m At each tree level, the values of Rhv(m) are computed online for all possible positions in each of the two tracks being tested. It can be seen from Equation (48) that only the correlations φ′(mk ,mk+1) corresponding to pulse positions in two consecutive tracks need to be stored (4 × 16 × 16 words), along with the correlations φ′(mk , mk )corresponding to the diagonal of the matrix Φ (64 words). Thus the memory requirement in the present algebraic structure is 1088 words instead of 64 × 64 = 4096 words. The search procedures at the different bit rates modes are similar. The difference is in the number of pulses, and accordingly, the number of levels in the tree search. In order to keep a comparable search complexity across the different codebooks, the number of tested positions is kept similar. The search in the 12.65-kbit/s mode will be described as an example. In this mode, 2 pulses are placed in each track giving a total of 8 pulses per subframe of length 64. Two pulses are searched at a time, and these two pulses always correspond to consecutive tracks. That is, the two searched pulses are in tracks T0–T1, T1–T2, T2–T3, or T3–T0. The tree has 4 levels in this case. At the first level, pulse P0 is assigned to track T0 and pulse P1 to track T1. In this level, no search is performed and the two pulse positions are set to the maximum of b(n) in each track. In the second level, pulse P2 is assigned to track T2 and pulse P3 to track T3. Four positions for pulse P2 are tested against all 16 positions of pulse P3. The 4 tested positions of P2 are determined based on the maxima of b(n) in the track. In the third level, pulse P4 is assigned to track T1 and pulse P5 to track T2. Eight positions for pulse P4 are tested against all 16 positions of pulse P5. Similar to the previous search level, the 8 tested positions of P4 are determined based on the maxima of b(n) in the track. In the fourth level, pulse P6 is assigned to track T3 and pulse P7 to track T0. Eight positions for pulse P6 are tested against all 16 positions of pulse P7. Thus the total number of tested combination is 4 × 16 + 8 × 16 + 8 × 16 = 320. The whole process is repeated 4 times (4 iterations) by assigning the pulses to different tracks. For example, in the 2nd iteration, pulses P0 to P7 are assigned to tracks T1, T2, T3, T0, T2, T3, T0, and T1, respectively. Thus the total number of tested position combinations is 4 × 320 = 1280. ITU-T Rec. G.722.2 (07/2003) 35 As another search example, in the 15.85-kbit/s mode, 3 pulses are placed in each track giving a total of 12 pulses. There are 6 levels in the tree search whereby two pulses are searched in each level. In the first two levels, 4 pulses are set to the maxima of b(n). In the subsequent 4 levels, the number of tested combinations are 4 × 16, 6 × 16, 8 × 16, and 8 × 16, respectively. Four iterations are used giving a total of 4 × 26 × 16 = 1664 combinations. 5.9 Quantization of the adaptive and fixed codebook gains The adaptive codebook gain (pitch gain) and the fixed (algebraic) codebook gain are vector quantized using a 6-bit codebook for modes 8.85 and 6.60 kbit/s and using a 7-bit codebook for all the other modes. The fixed codebook gain quantization is performed using MA prediction with fixed coefficients. The 4th-order MA prediction is performed on the innovation energy as follows. Let E(n) be the mean-removed innovation energy (in dB) at subframe n, and given by: ∑ E(n) = 10 log  1 N gc2 N −1 i=0 c2(i) − E (49) where N = 64 is the subframe size, c(i) is the fixed codebook excitation, and E = 30 dB is the mean of the innovation energy. The predicted energy is given by: E~(n) = 4 ∑ bi Rˆ (n − i) (50) i=1 where [b1 b2 b3 b4] = [0.5,0.4,0.3,0.2] are the MA prediction coefficients, and Rˆ(k) is the quantized energy prediction error at subframe k. The predicted energy is used to compute a predicted fixedcodebook gain gc′ as in Equation (49) (by substituting E(n) by E~(n) and gc by gc′ ). This is done as follows. First, the mean innovation energy is found by: ∑ Ei = 10 log  1 N N −1 i=0 c 2 (i ) (51) and then the predicted gain gc′ is found by: gc′ = 100.05(E~(n)+E −Ei ) (52) A correction factor between the gain gc and the estimated one gc′ is given by: γ = gc gc′ (53) Note that the prediction error is given by: R(n) = E(n)− E~(n) = 20log (γ) (54) The pitch gain, gp, and correction factor γ are jointly vector quantized using a 6-bit codebook for modes 8.85 and 6.60 kbit/s, and 7-bit codebook for other modes. The gain codebook search is performed by minimizing the mean-square of the weighted error between original and reconstructed speech which is given: E = xt x + g 2 p yt y + gc2 zt z − 2gp xt y − 2gcxt z + 2gp gc yt z (55) where x is the target vector, y is the filtered adaptive codebook vector, and z is the filtered fixed codebook vector. (Each gain vector in the codebook also has an element representing the quantized energy prediction error.) The quantized energy prediction error associated with the chosen gains is 36 ITU-T Rec. G.722.2 (07/2003) used to update Rˆ(n). In the search, only the 64 codevectors that are closest to the unquantized pitch gain, gp, are taken into account. 5.10 Memory update An update of the states of the synthesis and weighting filters is needed in order to compute the target signal in the next subframe. After the two gains have been quantized, the excitation signal, u(n), in the present subframe is found by: u(n) = gˆp v(n)+ gˆcc(n), n = 0,K,63 (56) where gˆp and gˆc are the quantized adaptive and fixed codebook gains, respectively, vi(n) the adaptive codebook vector (interpolated past excitation), and c(n) is the fixed codebook vector (algebraic code including pitch sharpening). The states of the filters can be updated by filtering the signal r(n)−u(n) (difference between residual and excitation) through the filters 1/ Aˆ(z) and A(z γ1)Hde-emph (z) for the 64-sample subframe and saving the states of the filters. This would require 3 filterings. A simpler approach which requires only one filtering is as follows. The local synthesis speech, sˆ(n) , is computed by filtering the excitation signal through 1/ Aˆ(z) . The output of the filter due to the input r(n)−u(n) is equivalent to e(n) = s(n)− sˆ(n). So the states of the synthesis filter 1/ Aˆ(z) are given by e(n), n = 48,…,63. Updating the states of the filter A(z γ1)Hde-emph (z) can be done by filtering the error signal e(n) through this filter to find the perceptually weighted error ew(n). However, the signal ew(n) can be equivalently found by: ew(n) = x(n)− gˆp y(n) = gˆcz(n) (57) Since the signals x(n), y(n), and z(n) are available, the states of the weighting filter are updated by computing ew(n) as in Equation (57) for n = 48,...,63. This saves two filterings. 5.11 High-band gain generation In order to compute the high-band gain for 23.85-kbit/s mode, 16-kHz input speech is filtered through a band-pass FIR filter HHB(z) which has the passband from 6.4 to 7 kHz. The high-band gain gHB is obtained by: 63 ∑ (sHB (i ))2 ∑ g HB = i=0 63 (sHB2 (i))2 (58) i=0 where sHB(i) is band-pass filtered input speech and sHB2(i) is high-band speech synthesis obtained from high-band excitation uHB2(i) filtered through high-band synthesis filter AHB(z) described in 6.3.2.2. 6 Functional description of the decoder The function of the decoder consists of decoding the transmitted parameters (LP parameters, adaptive codebook vector, adaptive codebook gain, fixed codebook vector, fixed codebook gain and high-band gain) and performing synthesis to obtain the reconstructed speech. The reconstructed speech is then postprocessed and upsampled (and upscaled). Finally high-band signal is generated to the frequency band from 6 to 7 kHz. The signal flow at the decoder is shown in Figure 3. ITU-T Rec. G.722.2 (07/2003) 37 6.1 Decoding and speech synthesis The decoding process is performed in the following order: Decoding of LP filter parameters: The received indices of ISP quantization are used to reconstruct the quantized ISP vector. The interpolation described in 5.2.6 is performed to obtain 4 interpolated ISP vectors (corresponding to 4 subframes). For each subframe, the interpolated ISP vector is converted to LP filter coefficient domain ak, which is used for synthesizing the reconstructed speech in the subframe. The following steps are repeated for each subframe: 1) Decoding of the adaptive codebook vector: The received pitch index (adaptive codebook index) is used to find the integer and fractional parts of the pitch lag. The adaptive codebook vector v(n) is found by interpolating the past excitation u(n) (at the pitch delay) using the FIR filter described in 5.6. The received adaptive filter index is used to find out whether the filtered adaptive codebook is v1(n) = v(n) or v2(n) = 0.18v(n) + 0.64v(n – 1) + 0.18v(n – 2). 2) Decoding of the innovative vector: The received algebraic codebook index is used to extract the positions and amplitudes (signs) of the excitation pulses and to find the algebraic codevector c(n). If the integer part of the pitch lag is less than the subframe size 64, the pitch sharpening procedure is applied which translates into modifying c(n) by filtering it through the adaptive prefilter F(z) which consists of two parts: a periodicity enhancement part 1/(1–0.85z−T) and a tilt part (1 – β1 z−1), where T is the integer part of the pitch lag and β1(n) is related to the voicing of the previous subframe and is bounded by [0.0,0.5]. 3) Decoding of the adaptive and innovative codebook gains: The received index gives the fixed codebook gain correction factor γˆ . The estimated fixed codebook gain g'c is found as described in 5.8. First, the predicted energy for every subframe n is found by: E~(n) = 4 ∑ bi Rˆ (n − i ) (59) i=1 and then the mean innovation energy is found by: ∑ E i = 10 log  1 N N −1 c i=0 2 (i ) (60) The predicted gain gc′ is found by: gc′ = 100.05(E~(n)+E −Ei ) (61) The quantized fixed codebook gain is given by: gˆc = γˆgc′ (62) 4) Computing the reconstructed speech: The following steps are for n = 0,..., 63. The total excitation is constructed by: u(n) = gˆp v(n)+ gˆcc(n) (63) Before the speech synthesis, a post-processing of excitation elements is performed. 5) Anti-sparseness processing (6.60- and 8.85-kbit/s modes): An adaptive anti-sparseness post-processing procedure is applied to the fixed codebook vector c(n) in order to reduce perceptual artifacts arising from the sparseness of the algebraic fixed codebook vectors with only a few non-zero samples per subframe. The anti-sparseness processing consists of circular convolution of the fixed codebook vector with an impulse response. Three prestored impulse responses are used and a number impNr = 0,1,2 is set to select one of them. 38 ITU-T Rec. G.722.2 (07/2003) A value of 2 corresponds to no modification, a value of 1 corresponds to medium modification, while a value of 0 corresponds to strong modification. The selection of the impulse response is performed adaptively from the adaptive and fixed codebook gains. The following procedure is employed: if gˆp < 0.6 then impNr = 0; else if gˆp < 0.9 then impNr = 1; else impNr = 2; Detect onset by comparing the fixed codebook gain to the previous fixed codebook gain. If the current value is more than three times the previous value, an onset is detected. If not onset and impNr = 0, the median filtered value of the current and the previous 4 adaptive codebook gains are computed. If this value is less than 0.6, impNr = 0. If not onset, the impNr value is restricted to increase by one step from the previous subframe. If an onset is declared, the impNr value is increased by one if it is less than 2. In case of 8.85-kbit/s mode, the impNr value is increased by one. 6) Noise enhancer: A non-linear gain smoothing technique is applied to the fixed codebook gain gˆc in order to enhance excitation in noise. Based on the stability and voicing of the speech segment, the gain of the fixed codebook is smoothed in order to reduce fluctuation in the energy of the excitation in case of stationary signals. This improves the performance in case of stationary background noise. The voicing factor is given by λ = 0.5(1–rv) with rv = (Ev – Ec)/(Ev + Ec), where Ev and Ec are the energies of the scaled pitch codevector and scaled innovation codevector, respectively. Note that since the value of rv is between –1 and 1, the value of λ is between 0 and 1. Note that the factor λ is related to the amount of unvoicing with a value of 0 for purely voiced segments and a value of 1 for purely unvoiced segments. A stability factor θ is computed based on a distance measure between the adjacent LP filters. Here, the factor θ is related to the ISP distance measure and it is bounded by 0 ≤ θ ≤ 1, with larger values of θ corresponding to more stable signals. Finally, a gain smoothing factor Sm is given by: Sm = λθ (64) The value of Sm approaches 1 for unvoiced and stable signals, which is the case of stationary background noise signals. For purely voiced signals or for unstable signals, the value of Sm approaches 0. An initial modified gain g0 is computed by comparing the fixed codebook gain gˆc to a threshold given by the initial modified gain from the previous subframe, g–1. If gˆc is larger or equal to g–1, then g0 is computed by decrementing gˆc by 1.5 dB bounded by g0 ≥ g–1. If gˆc is smaller than g–1, then g0 is computed by incrementing gˆc by 1.5 dB bounded by g0 ≤ g–1. Finally, the gain is updated with the value of the smoothed gain as follows: gˆc = θg0 + (1+ θ)gˆc (65) ITU-T Rec. G.722.2 (07/2003) 39 7) Pitch enhancer: A pitch enhancer procedure modifies the total excitation u(n) by filtering the fixed codebook excitation through an innovation filter whose frequency response emphasizes the higher frequencies more than lower frequencies, and whose coefficients are related to the periodicity in the signal. A filter of the form: Finno (z) = −cpe z +1− cpe z−1 (66) where cpe = 0.125(1 − rv), with rv = (Ev – Ec)/(Ev + Ec) as described above. The filtered fixed codevector is given by: c′(n) = c(n)− cpe (c (n +1)+ c (n −1)) (67) and the updated excitation is given by: u(n) = gˆpv (n)+ gˆc c'(n) (68) The above procedure can be done in one step by updating the excitation as follows: u(n) = u(n)− gˆccpe (c (n +1)+ c (n −1)) (69) 8) Post-processing of excitation elements (6.60- and 8.85-kbit/s modes): A post-processing of excitation elements procedure is applied to the total excitation u(n) by emphasizing the contribution of the adaptive codebook vector: uˆ(n ) = u(n) + u(n), 0.25βgˆp v (n), gˆp > 0.5 gˆp ≤ 0.5 (70) Adaptive gain control (AGC) is used to compensate for the gain difference between the non-emphasized excitation u(n) and emphasized excitation û(n). The gain scaling factor η for the emphasized excitation is computed by:  ∑ η =   ∑   63 n=0 63 n=0 u uˆ 2 2 (n) (n) , gˆ p > 0.5 1.0, gˆp ≤ 0.5 (71) The gain-scaled emphasized excitation signal uˆ′(n) is given by: uˆ′(n) = uˆ(n)η (72) The reconstructed speech for the subframe of size 64 is given by: 16 sˆ(n) = uˆ(n)− ∑ aˆisˆ(n − i), n = 0,K,63 (73) i=1 where âi are the interpolated LP filter coefficients. The synthesis speech sˆ(n) is then passed through an adaptive postprocessing which is described in the following clause. 6.2 High-pass filtering, up-scaling and interpolation The high-pass filter serves as a precaution against undesired low-frequency components. The signal is filtered through the high-pass filter Hh1(z) and de-emphasis filter Hde-emph(z). Finally, the signal is upsampled to 16 kHz to obtain the lower-band synthesis signal sˆ16k (n) . sˆ16k (n) is produced by first upsampling the lower-band synthesis sˆ12.8k (n) at 12.8 kHz by 5, then filtering the output through Hdecim(z), and finally downsampling it by 4. 40 ITU-T Rec. G.722.2 (07/2003) (Up-scaling consists of multiplying the output from the high-pass filtering by a factor of 2 in order to compensate the down-scaling at the preprocessing stage.) 6.3 High frequency band For the higher frequency band (6.4-7.0 kHz), excitation is generated to model the highest frequencies. The high frequency content is generated by filling the upper part of the spectrum with a white noise properly scaled in the excitation domain, then converted to the speech domain by shaping it with a filter derived from the same LP synthesis filter used for synthesizing the downsampled signal. 6.3.1 Generation of high-band excitation The high-band excitation is obtained by first generating white noise uHB1(n). The power of the highband excitation is set equal to the power of the lower band excitation u2(n) which means that: 63 ∑ u2 2 (k ) ∑ uHB2(n) = uHB1(n) k =0 63 u 2 HB1 (k ) (74) k =0 Finally the high-band excitation is found by: uHB (n) = gˆ HBuHB2 (n) (75) where gˆ HB is a gain factor. In the 23.85-kbit/s mode, gˆ HB is decoded from the received gain index. In 6.60-, 8.85-, 12.65-, 14.25-, 15.85-, 18.25-, 19.85- and 23.05-kbit/s modes, gHB is estimated using voicing information bounded by [0.1,1.0]. First, tilt of synthesis etilt is found: 63 ∑ sˆhp (n)sˆhp (n −1) ∑ ( ) etilt = n=1 63 sˆhp2 n (76) n=0 where sˆhp (n) is high-pass filtered lower band speech synthesis sˆ12.8k (n) with cut-off frequency of 400 Hz. The gˆ HB is then found by: gˆ HB = wSP gSP + (1− wSP )gBG (77) where gSP = 1 – etilt is gain for speech signal, gBG = 1.25gSP is gain for background noise signal, and wSP is a weighting function set to 1, when VAD is ON, and 0 when VAD is OFF. gHB is bounded between [0.1,1.0]. In case of voiced segments where less energy is present at high frequencies, etilt approaches 1 resulting in a lower gain gHB. This reduces the energy of the generated noise in case of voiced segments. ITU-T Rec. G.722.2 (07/2003) 41 6.3.2 LP filter for the high frequency band 6.3.2.1 6.60-kbit/s mode The high-band LP synthesis filter AHB(z) is found by extrapolating the quantized ISF vector f into 20th-order ISF vector fe. First, maximum of the autocorrelation Cmax(i) of ISF vector difference vector f∆ (i) = f (i +1)− f (i),i = 1,...,14 is obtained. Then new 16-kHz ISF vector fe'(i) is computed by: fe′(i) =    f (i −1), fe′(i −1)+ fe′(i − Cmax (i)−1)− fe′(i − Cmax (i)− 2), i = 1,...,15 i = 16,..,19 (78) An approximation of the last element of new ISF vector fe19 is updated based on lower frequency coefficients. New extrapolated ISF vector difference vector fe′∆ (i) is: fe′∆ (i) = ( cscale fe′(i)− fe′(i −1)), i = 16,...,19 (79) where cscale scales fe′∆ (i) so that fe(19) will be equal to fe19. In order to insure stability, fe′∆ (i) is bounded by: fe′∆ (i)+ fe′∆ (i −1) > 500, i = 17,...,19 (80) Finally, the extrapolated ISF vector fe is obtained by:  f (i), i = 1,...,15 fe (i) =   fe′∆ (i)+ fe (i −1), i = 16,...,19 (81)  f (16), i = 20 fe is converted to cosine domain to obtain qe with 16 000-Hz sampling rate. The high-band LP synthesis filter AHB(z) is obtained by converting qe to LP filter as described in 5.2.4 with m = 20. 6.3.2.2 8.85-, 12.65-, 14.25-, 15.85-, 18.25-, 19.85-, 23.05- or 23.85-kbit/s modes The high-band LP synthesis filter AHB(z) is weighted low-band LP synthesis filter: AHB (z) = Aˆ(z 0.8) (82) where Â(z) is the interpolated LP synthesis filter. Â(z) has been computed analysing signal with the sampling rate of 12.8 kHz but it is now used for a 16-kHz signal. Effectively, this means that the frequency response FR16(f) of AHB(z) is obtained by: FR16 ( f )= FR12.8   12.8 16 f   (83) where FR12.8(f) is the frequency response of A(z). This means that the band 5.1-5.6 kHz in the 12.8-kHz domain will be mapped to 6.4-7.0 kHz in the 16-kHz domain. 6.3.3 High-band synthesis uHB(n) is filtered through AHB(z). The output of this high-band synthesis sHB(n) is filtered through a band-pass FIR filter HHB(z) which has the passband from 6 to 7 kHz. Finally, sHB is added to synthesized speech sˆ16k (n) to produce the synthesized output speech signal sˆoutput (n) . 42 ITU-T Rec. G.722.2 (07/2003) 7 Detailed bit allocation of the adaptive multi-rate wideband codec The detailed allocation of the bits in the adaptive multi-rate wideband speech encoder is shown for each mode in Tables 12a-12i. These tables show the order of the bits produced by the speech encoder. Note that the most significant bit (MSB) of each codec parameter is always sent first. Table 12a/G.722.2 – Source encoder output parameters in order of occurrence and bit allocation within the speech frame of 477 bits/20 ms, 23.85-kbit/s mode Bits (MSB-LSB) s1 s2-s9 s10-s17 s18-s23 s24-s30 s31-s37 s38-s42 s43-s47 s48-s56 s57 s58-s68 s69-s79 s80-s90 s91-s101 s102-s112 s113-s123 s124-s134 s135-s145 s146-s152 s153-s156 s157-s162 s163-s262 s263-s371 s372-s477 subframe 1 subframe 2 subframe 3 subframe 4 Description VAD-flag index of 1st ISP subvector index of 2nd ISP subvector index of 3rd ISP subvector index of 4th ISP subvector index of 5th ISP subvector index of 6th ISP subvector index of 7th ISP subvector adaptive codebook index LTP-filtering-flag Codebook Index1 for track 1 Codebook Index1 for track 2 Codebook Index1 for track 3 Codebook Index1 for track 4 Codebook Index2 for track 1 Codebook Index2 for track 2 Codebook Index2 for track 3 Codebook Index2 for track 4 codebook gains High-band energy adaptive codebook index (relative) same description as s57-s156 same description as s48-s156 same description as s157-s262 ITU-T Rec. G.722.2 (07/2003) 43 Table 12b/G.722.2 – Source encoder output parameters in order of occurrence and bit allocation within the speech frame of 461 bits/20 ms, 23.05-kbit/s mode Bits (MSB-LSB) s1 s2-s9 s10-s17 s18-s23 s24-s30 s31-s37 s38-s42 s43-s47 s48-s56 s57 s58-s68 s69-s79 s80-s90 s91-s101 s102-s112 s113-s123 s124-s134 s135-s145 s146-s152 s153-s158 s159-s254 s255-s359 s360-s461 subframe 1 subframe 2 subframe 3 subframe 4 Description VAD-flag index of 1st ISP subvector index of 2nd ISP subvector index of 3rd ISP subvector index of 4th ISP subvector index of 5th ISP subvector index of 6th ISP subvector index of 7th ISP subvector adaptive codebook index LTP-filtering-flag Codebook Index1 for track 1 Codebook Index1 for track 2 Codebook Index1 for track 3 Codebook Index1 for track 4 Codebook Index2 for track 1 Codebook Index2 for track 2 Codebook Index2 for track 3 Codebook Index2 for track 4 codebook gains adaptive codebook index (relative) same description as s57-s152 same description as s48-s152 same description as s153-s254 44 ITU-T Rec. G.722.2 (07/2003) Table 12c/G.722.2 – Source encoder output parameters in order of occurrence and bit allocation within the speech frame of 397 bits/20 ms, 19.85-kbit/s mode Bits (MSB-LSB) s1 s2-s9 s10-s17 s18-s23 s24-s30 s31-s37 s38-s42 s43-s47 s48-s56 s57 s58-s67 s68-s77 s78-s79 s80-s81 s82-s91 s92-s101 s102-s115 s116-s129 s130-s136 s137-s142 s143-s222 s223-s311 s312-s397 subframe 1 subframe 2 subframe 3 subframe 4 Description VAD-flag index of 1st ISP subvector index of 2nd ISP subvector index of 3rd ISP subvector index of 4th ISP subvector index of 5th ISP subvector index of 6th ISP subvector index of 7th ISP subvector adaptive codebook index LTP-filtering-flag Codebook Index1 for track 1 Codebook Index1 for track 2 Pulse Selector for track 3 Pulse Selector for track 4 Codebook index2 for track 1 Codebook index2 for track 2 Codebook index for track 3 Codebook index for track 4 VQ gain adaptive codebook index (relative) same description as s57-s136 same description as s48-s136 same description as s137-s222 ITU-T Rec. G.722.2 (07/2003) 45 Table 12d/G.722.2 – Source encoder output parameters in order of occurrence and bit allocation within the speech frame of 365 bits/20 ms, 18.25-kbit/s mode Bits (MSB-LSB) s1 s2-s9 s10-s17 s18-s23 s24-s30 s31-s37 s38-s42 s43-s47 s48-s56 s57 s58-s59 s60-s61 s62-s63 s64-s65 s66-s79 s80-s93 s94-s107 s108-s121 s122-s128 s129-s134 s135-s206 s207-s287 s288-s365 subframe 1 subframe 2 subframe 3 subframe 4 Description VAD-flag index of 1st ISP subvector index of 2nd ISP subvector index of 3rd ISP subvector index of 4th ISP subvector index of 5th ISP subvector index of 6th ISP subvector index of 7th ISP subvector adaptive codebook index LTP-filtering-flag Pulse Selector for track 1 Pulse Selector for track 2 Pulse Selector for track 3 Pulse Selector for track 4 Codebook index for track 1 Codebook index for track 2 Codebook index for track 3 Codebook index for track 4 VQ gain adaptive codebook index (relative) same description as s57-s128 same description as s48-s128 same description as s129-s206 46 ITU-T Rec. G.722.2 (07/2003) Table 12e/G.722.2 – Source encoder output parameters in order of occurrence and bit allocation within the speech frame of 317 bits/20 ms, 15.85-kbit/s mode Bits (MSB-LSB) s1 s2-s9 s10-s17 s18-s23 s24-s30 s31-s37 s38-s42 s43-s47 s48-s56 s57 s58-s70 s71-s83 s84-s96 s97-s109 s110-s116 s117-s122 s123-s182 s183-s251 s252-s317 subframe 1 subframe 2 subframe 3 subframe 4 Description VAD-flag index of 1st ISP subvector index of 2nd ISP subvector index of 3rd ISP subvector index of 4th ISP subvector index of 5th ISP subvector index of 6th ISP subvector index of 7th ISP subvector adaptive codebook index LTP-filtering-flag Codebook index for track 1 Codebook index for track 2 Codebook index for track 3 Codebook index for track 4 VQ gain adaptive codebook index (relative) same description as s57-s116 same description as s48-s116 same description as s117-s182 ITU-T Rec. G.722.2 (07/2003) 47 Table 12f/G.722.2 – Source encoder output parameters in order of occurrence and bit allocation within the speech frame of 285 bits/20 ms, 14.25-kbit/s mode Bits (MSB-LSB) s1 s2-s9 s10-s17 s18-s23 s24-s30 s31-s37 s38-s42 s43-s47 s48-s56 s57 s58-s70 s71-s83 s84-s92 s93-s101 s102-s108 s109-s114 s115-s166 s167-s227 s228-s285 subframe 1 subframe 2 subframe 3 subframe 4 Description VAD-flag index of 1st ISP subvector index of 2nd ISP subvector index of 3rd ISP subvector index of 4th ISP subvector index of 5th ISP subvector index of 6th ISP subvector index of 7th ISP subvector adaptive codebook index LTP-filtering-flag Codebook index for track 1 Codebook index for track 2 Codebook index for track 3 Codebook index for track 4 VQ gain adaptive codebook index (relative) same description as s57-s108 same description as s48-s108 same description as s109-s166 48 ITU-T Rec. G.722.2 (07/2003) Table 12g/G.722.2 – Source encoder output parameters in order of occurrence and bit allocation within the speech frame of 253 bits/20 ms, 12.65-kbit/s mode Bits (MSB-LSB) s1 s2-s9 s10-s17 s18-s23 s24-s30 s31-s37 s38-s42 s43-s47 s48-s56 s57 s58-s66 s67-s75 s76-s84 s85-s93 s94-s100 s101-s106 s107-s150 s151-s203 s204-s253 subframe 1 subframe 2 subframe 3 subframe 4 Description VAD-flag index of 1st ISP subvector index of 2nd ISP subvector index of 3rd ISP subvector index of 4th ISP subvector index of 5th ISP subvector index of 6th ISP subvector index of 7th ISP subvector adaptive codebook index LTP-filtering-flag Codebook index for track 1 Codebook index for track 2 Codebook index for track 3 Codebook index for track 4 VQ gain adaptive codebook index (relative) same description as s57-s100 same description as s48-s100 same description as s101-s150 ITU-T Rec. G.722.2 (07/2003) 49 Table 12h/G.722.2 – Source encoder output parameters in order of occurrence and bit allocation within the speech frame of 177 bits/20 ms, 8.85-kbit/s mode Bits (MSB-LSB) s1 s2-s9 s10-s17 s18-s23 s24-s30 s31-s37 s38-s42 s43-s47 s48-s55 s56-s60 s61-s65 s66-s70 s71-s75 s76-s81 s82-s86 s87-s112 s113-s146 s147-s177 subframe 1 subframe 2 subframe 3 subframe 4 Description VAD-flag index of 1st ISP subvector index of 2nd ISP subvector index of 3rd ISP subvector index of 4th ISP subvector index of 5th ISP subvector index of 6th ISP subvector index of 7th ISP subvector adaptive codebook index Codebook index for track 1 Codebook index for track 2 Codebook index for track 3 Codebook index for track 4 VQ gain adaptive codebook index (relative) same description as s56-s81 same description as s48-s81 same description as s82-s112 50 ITU-T Rec. G.722.2 (07/2003) Table 12i/G.722.2 – Source encoder output parameters in order of occurrence and bit allocation within the speech frame of 132 bits/20 ms, 6.60-kbit/s mode Bits (MSB-LSB) s1 s2-s9 s10-s17 s18-s24 s25-s31 s32-s37 s38-s45 s46-57 s58-s63 s64-s68 s69-s86 s87-s109 s110-s132 subframe 1 subframe 2 subframe 3 subframe 4 Description VAD-flag index of 1st ISP subvector index of 2nd ISP subvector index of 3rd ISP subvector index of 4th ISP subvector index of 5th ISP subvector adaptive codebook index Codebook Index VQ gain adaptive codebook index (relative) same description as s46-s63 same description as s64-s86 same description as s64-s86 8 Homing sequences 8.1 Functional description The adaptive multi-rate wideband speech codec is described in a bit-exact arithmetic to allow easy type approval as well as general testing of correct operation of the adaptive multi-rate wideband speech codec. The response of the codec to a predefined input sequence can only be foreseen if the internal state variables of the codec are in a predefined state at the beginning of the experiment. Therefore, the codec has to be put in a so-called home state before a bit-exact test can be performed. This is usually done by a reset (a procedure in which the internal state variables of the codec are set to their defined initial values). The codec mode of the speech encoder and speech decoder shall be set to the tested codec mode by external means at reset. To allow a reset of the codec in remote locations, special homing frames have been defined for the encoder and the decoder, thus enabling a codec homing by inband signalling. The codec homing procedure is defined in such a way that in either direction (encoder or decoder), the homing functions are called after the processing of the homing frame. The output corresponding to the first homing frame is therefore dependent on the used codec mode and the codec state when receiving that frame and hence usually not known. The response of the encoder to any further homing frame is by definition the corresponding decoder homing frame for the used codec mode. The response of the decoder to any further homing frame is by definition the encoder homing frame. This procedure allows homing of both the encoder and decoder from either side, if a loop back configuration is implemented, taking proper framing into account. ITU-T Rec. G.722.2 (07/2003) 51 8.2 Definitions 8.2.1 Encoder homing frame: The encoder homing frame consists of 320 identical samples, each 13 bits long, with the least significant bit set to "one" and all other bits set to "zero". When written to 16-bit words with left justification, the samples have a value of 0008 hex. The speech decoder has to produce this frame as a response to the second and any further decoder homing frame if at least two decoder homing frames were input to the decoder consecutively. The encoder homing frame is identical for all codec modes. 8.2.2 Decoder homing frame: There exist nine different decoder homing frames, which correspond to the nine AMR-WB codec modes. Using one of these codec modes, the corresponding decoder homing frame is the natural response of the speech encoder to the second and any further encoder homing frame if at least two encoder homing frames were input to the encoder consecutively. In Annex C, for each decoder homing frame, the parameter values are given. 8.3 Encoder homing Whenever the adaptive multi-rate wideband speech encoder receives at its input an encoder homing frame exactly aligned with its internal speech frame segmentation, the following events take place: Step 1 The speech encoder performs its normal operation including VAD and SCR and produces in accordance with the used codec mode a speech parameter frame at its output which is in general unknown. But if the speech encoder was in its home state at the beginning of that frame, then the resulting speech parameter frame is identical to that decoder homing frame, which corresponds to the used codec mode (this is the way how the decoder homing frames were constructed). Step 2 After successful termination of that operation, the speech encoder provokes the homing functions for all submodules, including VAD and SCR, and sets all state variables into their home state. On the reception of the next input frame, the speech encoder will start from its home state. NOTE – Applying a sequence of N encoder homing frames will cause at least N–1 decoder homing frames at the output of the speech encoder. 8.4 Decoder homing Whenever the speech decoder receives at its input a decoder homing frame, which corresponds to the used codec mode, then the following events take place: Step 1 The speech decoder performs its normal operation and produces a speech frame at its output which is, in general, unknown. But if the speech decoder was in its home state at the beginning of that frame, then the resulting speech frame is replaced by the encoder homing frame. This would not naturally be the case but is forced by this definition here. Step 2 After successful termination of that operation, the speech decoder provokes the homing functions for all submodules, including the comfort noise generator, and sets all state variables into their home state. On the reception of the next input frame, the speech decoder will start from its home state. NOTE 1 – Applying a sequence of N decoder homing frames will cause at least N–1 encoder homing frames at the output of the speech decoder. NOTE 2 – By definition, the first frame of each decoder test sequence must differ from the decoder homing frame at least in one bit position within the parameters for LPC and first subframe. Therefore, if the decoder 52 ITU-T Rec. G.722.2 (07/2003) is in its home state, it is sufficient to check only these parameters to detect a subsequent decoder homing frame. This definition is made to support a delay-optimized implementation. 9 Voice Activity Detector (VAD) The function of the VAD algorithm is to indicate whether each 20-ms frame contains signals that should be transmitted, e.g., speech, music or information tones. The output of the VAD algorithm is a Boolean flag (VAD_flag) indicating presence of such signals. This flag is used in the AMR-WB speech coder and Annex B. 9.1 VAD symbols For the purposes of this clause, the following symbols apply. 9.1.1 VAD variables bckr_est[n] burst_count hang_count level[n] new_speech background noise estimate at the frequency band "n" counts length of a speech burst, used by VAD hangover addition hangover counter, used by VAD hangover addition signal level at the frequency band "n" pointer of the speech encoder, points a buffer containing last received samples of a speech frame noise_level pow_sum s(i) snr_sum speech_level stat_count stat_rat tone_flag vad_thr VAD_flag vadreg estimated noise level input power samples of the input frame measure between input frame and noise estimate estimated speech level stationary counter measure indicating stationary of the input frame flag indicating the presence of a tone VAD threshold Boolean VAD flag intermediate VAD decision 9.1.2 VAD constants ALPHA_UP1 ALPHA_DOWN1 ALPHA_UP2 ALPHA_DOWN2 ALPHA3 ALPHA4 ALPHA5 BURST_HIGH BURST_P1 constant for updating noise estimate constant for updating noise estimate constant for updating noise estimate constant for updating noise estimate constant for updating noise estimate constant for updating average signal level constant for updating average signal level constant for controlling VAD hangover addition constant for controlling VAD hangover addition ITU-T Rec. G.722.2 (07/2003) 53 BURST_SLOPE constant for controlling VAD hangover addition COEFF3 coefficient for the filter bank COEFF5_1 coefficient for the filter bank COEFF5_2 coefficient for the filter bank HANG_HIGH constant for controlling VAD hangover addition HANG_LOW constant for controlling VAD hangover addition HANG_P1 constant for controlling VAD hangover addition HANG_SLOPE constant for controlling VAD hangover addition FRAME_LEN size of a speech frame, 256 samples (20 ms) MIN_SPEECH_LEVEL1 constant for speech estimation MIN_SPEECH_LEVEL2 constant for speech estimation MIN_SPEECH_SNR constant for VAD threshold adaptation NO_P1 constant for VAD threshold adaptation NO_SLOPE constant for VAD threshold adaptation NOISE_MAX maximum value for noise estimate NOISE_MIN minimum value for noise estimate POW_TONE_THR threshold for tone detection SP_ACTIVITY_COUNT constant for speech estimation SP_ALPHA_DOWN constant for speech estimation SP_ALPHA_UP constant for speech estimation SP_CH_MAX constant for VAD threshold adaptation SP_CH_MIN constant for VAD threshold adaptation SP_EST_COUNT constant for speech estimation SP_P1 constant for VAD threshold adaptation SP_SLOPE constant for VAD threshold adaptation STAT_COUNT threshold for stationary detection STAT_THR threshold for stationary detection STAT_THR_LEVEL threshold for stationary detection THR_HIGH constant for VAD threshold adaptation TONE_THR threshold for tone detection VAD_POW_LOW constant for controlling VAD hangover addition 9.1.3 Functions + Addition – Subtraction * Multiplication / Division 54 ITU-T Rec. G.722.2 (07/2003) |x| AND OR b ∑ x(n) n=a MIN(x,y) MAX(x,y) absolute value of x Boolean AND Boolean OR x(a) + x(a + 1) +. .. + x(b – 1) + x (b) = x,   y, x y ≤ < y x =    x, y, x y ≥ > y x 9.2 Functional description The block diagram of the VAD algorithm is depicted in Figure 4. The VAD algorithm uses parameters of the speech encoder to compute the Boolean VAD flag (VAD_flag). This input frame for VAD is sampled at the 12.8-kHz frequency and thus it contains 256 samples. Samples of the input frame (s(i)) are divided into sub-bands and level of the signal (level[n]) in each band is calculated. The normalized open-loop pitch gains are the input for the tone detection function, gains which are calculated by open-loop pitch analysis of the speech encoder. The tone detection function computes a flag (tone_flag) which indicates presence of a signalling tone, voiced speech, or other strongly periodic signal. Background noise level (bckr_est[n]) is estimated in each band based on the VAD decision, signal stationarity and the tone-flag. Intermediate VAD decision is calculated by comparing input SNR (level[n]/bckr_est[n]) to an adaptive threshold. The threshold is adapted based on noise and long-term speech estimates. Finally, the VAD flag is calculated by adding hangover to the intermediate VAD decision. Figure 4/G.722.2 – Simplified block diagram of the VAD algorithm 9.2.1 Filter bank and computation of sub-band levels The input signal is divided into frequency bands using a 12-band filter bank (Figure 5). Cut-off frequencies for the filter bank are shown in Table 13. ITU-T Rec. G.722.2 (07/2003) 55 Table 13/G.722.2 – Cut-off frequencies for the filter bank Band number 1 2 3 4 5 6 7 8 9 10 11 12 Frequencies 0-200 Hz 200-400 Hz 400-600 Hz 600-800 Hz 800-1200 Hz 1200-1600 Hz 1600-2000 Hz 2000-2400 Hz 2400-3200 Hz 3200-4000 Hz 4000-4800 Hz 4800-6400 Hz Input for the filter bank is a speech frame pointed by the new_speech pointer of the G.722.2 speech encoder. Input values for the filter bank are scaled down by one bit. This ensures safe scaling, i.e., saturation cannot occur during calculation of the filter bank. Figure 5/G.722.2 – Filter bank 56 ITU-T Rec. G.722.2 (07/2003) The filter bank consists of 5th- and 3rd-order filter blocks. Each filter block divides the input into high-pass and low-pass parts and decimates the sampling frequency by 2. The 5th-order filter block is calculated as follows: xlp (i) = 0.5× (A1(x(2× i))+ A2(x(2×i +1))) (84a) xhp (i) = 0.5× (A1(x(2× i))− A2(x(2×i +1))) (84b) where: x(i) input signal for a filter block xlp (i) low-pass component xhp (i) high-pass component The 3rd-order filter block is calculated as follows: xlp (i) = 0.5× (x(2×i +1)+ A3(x(2× i))) (85a) xhp (i) = 0.5× (x(2×i +1)− A3(x(2× i))) (85b) The filters A1(), A2(), and A3() are first-order direct form all-pass filters, whose transfer function is given by: A(z ) = 1 C+ +C z × −1 z −1 (86) where C is the filter coefficient. Coefficients for the all-pass filters A1(), A2(), and A3() are COEFF5_1, COEFF5_2, and COEFF3, respectively. Signal level is calculated at the output of the filter bank at each frequency band as follows: ENDn level(n) = ∑ xn (i) (87) i=STARTn where: n index for the frequency band x (i) sample i at the output of the filter bank at frequency band n n − 6, 1 ≤ n ≤ 4 STARTn = − − 12, 24, 5≤n≤8 9 ≤ n ≤ 11 − 48, n =12 7, 1≤ n ≤ 4 ENDn = 15, 31, 5≤n≤8 9 ≤ n ≤11 63, n =12 Negative indices of xn (i) refer to the previous frame. ITU-T Rec. G.722.2 (07/2003) 57 9.2.2 Tone detection The purpose of the tone detection function is to detect information tones, vowel sounds and other periodic signals. The tone detection uses normalized open-loop pitch gains (ol_gain), which are received from the speech encoder. If the pitch gain is higher than the constant TONE_THR, tone is detected and the tone flag is set: if (ol_gain > TONE_THR) tone_flag = 1 The open-loop pitch search and correspondingly the tone flag is computed twice in each frame, except for mode 6.60 kbit/s, where it is computed only once. 9.2.3 VAD decision The block diagram of the VAD decision algorithm is shown in Figure 6. Figure 6/G.722.2 – Simplified block diagram of the VAD decision algorithm Power of the input frame is calculated as follows: FRAME _ LEN frame _ pow = ∑ s(i) × s(i) (88) i=0 where samples s(i) of the input frame are pointed by the new_speech pointer of the speech encoder. Variable pow_sum is sum of the powers of the current and previous frames. If pow_sum is lower than the constant POW_TONE_THR, tone-flag is set to zero. 58 ITU-T Rec. G.722.2 (07/2003) The difference between the signal levels of the input frame and the background noise estimate is calculated as follows: snr _ sum = 12 ∑ n=1 MAX 1.0, level[n] bckr _ est[n] 2 (89) where: level[n] bckr_est[n] signal level at band n level of background noise estimate at band n VAD decision is made by comparing the variable snr_sum to a threshold. The threshold (vad_thr) is adapted to get desired sensitivity depending on estimated speech and background noise levels. Average background noise level is calculated by adding noise estimates at each band except the lowest band: 12 noise _ level = ∑bckr _ est[n] (90) n=2 If SNR is lower than the threshold (MIN_SPEECH_SNR), speech level is increased as follows: If (speech_level/noise_level < MIN_SPEECH_SNR) Speech_level = MIN_SPEECH_SNR × noise_level Logarithmic value for noise estimate is calculated as follows: ilog2_noise_level=log (noise_level) (91) 2 Before logarithmic value from the speech estimate is calculated, MIN_SPEECH_SNR×noise_level is subtracted from the speech level to correct its value in low SNR situations. ilog2_speech_level=log (speech_level–MIN_SPEECH_SNR ×noise_level) (92) 2 Threshold for VAD decision is calculated as follows: Vad_thr=NO_SLOPE×(ilog2_noise_level–NO_P1)+THR_HIGH+ MIN(SP_CH_MAX,MAX(SP_CH_MIN,SP_CH_MIN (93) +SP_SLOPE×(ilog2_speech_level–SP_P1))), where NO_SLOPE, SP_SLOPE, NO_P1, SP_P1, THR_HIGH, SP_CH_MAX and SP_CH_MIN are constants. The variable vadreg indicates intermediate VAD decision and it is calculated as follows: if (snr_sum > vad_thr) vadreg = 1 else vadreg = 0 9.2.3.1 Hangover addition Before the final VAD flag is given, a hangover is added. The hangover addition helps to detect low power endings of speech bursts, which are subjectively important but difficult to detect. ITU-T Rec. G.722.2 (07/2003) 59 VAD_flag is set to "1" if less than hang_len frames with "0" decision have been elapsed since burst_len consecutive "1" decisions have been detected. The variables hang_len and burst_len are computed using vad_thr as follows: hang_len = MAX(HANG_LOW,(HANG_SLOPE ×(vad_thr HANG_P1) + HANG_HIGH)) (94) burst_len = BURST_SLOPE × (vad_thr BURST_P1) + BURST_HIGH (95) The power of the input frame is compared to a threshold (VAD_POW_LOW). If the power is lower, the VAD flag is set to "0" and no hangover is added. The VAD_flag is calculated as follows: Vad_flag = 0; if (pow_sum < VAD_POW_LOW) burst_count = 0 hang_count = 0 else if (vadreg = 1) burst_count = burst_count + 1 if (burst_count ≥ burst_len) hang_count = hang_len VAD_flag = 1 else burst_count = 0 if (hang_count > 0) hang_count = hang_count – 1 VAD_flag = 1 9.2.3.2 Background noise estimation Background noise estimate (bckr_est[n]) is updated using amplitude levels of the previous frame. Thus, the update is delayed by one frame to avoid undetected start of speech bursts to corrupt the noise estimate. The update speed for the current frame is selected using intermediate VAD decisions (vadreg) and stationarity counter (stat_count) as follows: if (vadreg for the last 4 frames has been zero) alpha_up = ALPHA_UP1 alpha_down = ALPHA_DOWN1 else if (stat_count = 0) alpha_up = ALPHA_UP2 alpha_down = ALPHA_DOWN2 else alpha_up = 0 alpha_down = ALPHA3 The variable stat_count indicates stationary and its purpose is explained later in this clause. The variables alpha_up and alpha_down define the update speed for upwards and downwards, respectively. The update speed for each band "n" is selected as follows: if (bckr_est [n] < level [n]) m m–1 alpha[n] = alpha_up else alpha[n] = alpha_down 60 ITU-T Rec. G.722.2 (07/2003) Finally, noise estimate is updated as follows: bckr_estm+1[n]=(1.0 − alpha[n])× bckr_estm[n]+ alpha[n]×levelm−1[n] (96) where: n index of the frequency band m index of the frame Level of the background estimate (bckr_est[n]) is limited between constants NOISE_MIN and NOISE_MAX. If level of background noise increases suddenly, vadreg will be set to "1" and background noise is not normally updated upwards. To recover from this situation, update of the background noise estimate is enabled if the intermediate VAD decision (vadreg) is "1" for long enough time and spectrum is stationary. Stationary (stat_rat) is estimated using following equation: ∑ stat _ rat = 12 n=1 MAX (STAT_THR_LEVEL,MAX(ave _ levelm[n],levelm[n])) MAX (STAT_THR_LEVEL,MIN(ave _ levelm[n],levelm[n])) (97) where: STAT_THR_LEVEL a constant n index of the frequency band m index of the frame ave_level average level of the input signal If the stationary estimate (stat_rat) is higher than a threshold, the stationary counter (stat_count) is set to the initial value defined by constant STAT_COUNT. If the signal is not stationary but speech has been detected (VAD decision is "1"), stat_count is decreased by one in each frame until it is zero. if (5 last tone flags have been one) stat_count = STAT_COUNT else if (8 last internal VAD decisions have been zero) OR (stat_rat > STAT_THR) stat_count = STAT_COUNT else if (vadreg) AND (stat_count ≠ 0) stat_count = stat_count – 1 The average signal levels (ave_level[n]) are calculated as follows: ave_levelm+1[n]=(1.0 − alpha)× ave_levelm[n]+ alpha×levelm[n] (98) The update speed (alpha) for the previous equation is selected as follows: if (stat_count = STAT_COUNT) alpha = 1.0 else if (vadreg = 1) alpha=ALPHA5 else alpha = ALPHA4 ITU-T Rec. G.722.2 (07/2003) 61 9.2.3.3 Speech level estimation First, full-band input level is calculated by summing input levels in each band except the lowest band as follows: 12 in _ level = ∑ level[n] (99) n=2 A frame is assumed to contain speech if its level if high enough (MIN_SPEECH_LEVEL1), and the intermediate VAD flag (vadreg) is set or the input level is higher than the current speech level estimate. Maximum level (sp_max) from SP_EST_COUNT frames is searched. If the SP_ACTIVITY_COUNT number of speech frames is located in within SP_EST_COUNT number of frames, speech level estimate is updated by the maximum signal level (sp_max). The pseudo-code for the speech level estimation is as follows: If (SP_ACTIVITY_COUNT > SP_EST_COUNT – sp_est_cnt + sp_max_cnt) sp_est_cnt = 0 sp_max_cnt = 0 sp_max = 0 sp_est_cnt = sp_est_cnt + 1 if (in_level > MIN_SPEECH_LEVEL1) AND ((vadreg = 1) OR (in_level > speech_level)) sp_max_cnt = sp_max_cnt + 1 sp_max = MAX(sp_max, in_level) if (sp_max_cnt > SP_ACTIVITY_COUNT) if (sp_max > MIN_SPEECH_LEVEL2) if (sp_max > speech_level) speech_level = speech_level + SP_ALPHA_UP × (sp_max – speech_level) else speech_level = speech_level + SP_ALPHA_DOWN × (sp_max – speech_level) sp_max_cnt = 0 sp_max = 0 sp_est_cnt = 0 10 Mandatory AMR-WB codec modes for the speech telephony service in 3GPP This clause should be observed when interoperability with 3GPP standard-based wireless networks is required. To facilitate the implementation of AMR-WB for the circuit-switched speech telephony service in 3GPP systems, the number of mandatory AMR-WB codec modes have been reduced to five. This allows for less complexity for the channel coding in terminals and networks. Following modes are used: 23.85, 15.85, 12.65, 8.85 and 6.60 kbit/s. Based on listening test results for the speech telephony service channels, these five modes are considered sufficient for high-quality speech telephony service. For other services and applications in 3GPP, all the 9 modes remain in use. All 9 source codecs are kept in the 3GPP AMR-WB codec, but the number of modes used for the speech telephony service is just limited to five. In order to improve interoperability, the allowed AMR-WB codec mode configurations within active codec sets (ACS), i.e., which modes can be configured to be used within the mode adaptation 62 ITU-T Rec. G.722.2 (07/2003) at the same time, were further limited. The following three configurations are allowed: Configuration A (6.60, 8.85, 12.65), Configuration B (6.60, 8.85, 12.65, 15.85) and Configuration C (6.60, 8.85, 12.65, 23.85). This improves Tandem Free Operation/Transcoder Free Operation (TFO/TrFO) interoperability in 3GPP in the various speech service transmission channels because all these configurations are compatible. They contain the same three lower modes. This enables immediate establishment of TFO/TrFO (where double transcoding for calls between mobile terminals is avoided). This restriction for allowed mode configurations also simplifies signalling (e.g., in call set-up and handovers) and also drastically simplifies testing. Table 14 summarizes the requirements for support of AMR-WB mode configurations in 3GPP speech telephony service. Table 14/G.722.2 – Minimum requirements for AMR-WB modes in 3GPP speech telephony service Speech telephony service channel Terminal TCH/F Configuration A (6.60, 8.85 and 12.65) O-TCH/H Configuration A (6.60, 8.85 and 12.65) O-TCH/F Configurations A (6.60, 8.85, 12.65), B (6.60, 8.85, 12.65, 15.85) and C (6.60, 8.85, 12.65, 23.85) UTRAN Configurations A (6.60, 8.85, 12.65), B (6.60, 8.85, 12.65, 15.85) and C (6.60, 8.85, 12.65, 23.85) a) Support for configurations B and C is optional. Network Configuration A (6.60, 8.85 and 12.65) Configuration A (6.60, 8.85 and 12.65) Configuration A (6.60, 8.85 and 12.65)a) Configuration A (6.60, 8.85 and 12.65)a) ITU-T Rec. G.722.2 (07/2003) 63 BIBLIOGRAPHY [3] RABINER (L.R.), SCHAFER (R.W.): Digital processing of speech signals, Prentice-Hall Int., 1978. [4] ITAKURA (F.): Line spectral representation of linear predictive coefficients of speech signals, J. Acoust. Soc. Amer., Vol. 57, Supplement No. 1, S35, 1975. [5] BISTRITZ (Y.), PELLERM (S.): Immittance Spectral Pairs (ISP) for speech encoding, Proc. ICASSP'93, pp. II-9 to II-12. [6] PALIWAL (K.K.), ATAL (B.S.): Efficient vector quantization of LPC parameters at 24 bits/frame, IEEE Trans. Speech and Audio Processing, Vol. 1, No. 1, pp. 3-14, 1993. [7] KABAL (P.), RAMACHANDRAN (R.P.): The computation of line spectral frequencies using Chebyshev polynomials, IEEE Trans. on ASSP, Vol. 34, No. 6, pp. 1419-1426, 1986. [8] JÄRVINEN (K.), VAINIO (J.), KAPANEN (P.), HONKANEN (T.), HAAVISTO (P.), SALAMI (R.), LAFLAMME (C.), ADOUL (J.-P.): GSM enhanced full rate speech codec, Proc. ICASSP'97, pp. 771-774. [9] HONKANEN (T.), VAINIO (J.), JÄRVINEN (K.), HAAVISTO (P.), SALAMI (R.), LAFLAMME (C.), ADOUL (J.-P.): Enhanced full rate speech codec for IS-136 digital cellular system, Proc. ICASSP'97, pp. 731-734. [10] HAGEN (R.), EKUDDEN (E.), JOHANSSON (B.), KLEIJN (W.B.): Removal of sparse-excitation artifacts in CELP, Proc. ICASSP'98, pp. I-145 to I-148. [11] 3GPP TS 26.171: AMR Wideband Speech Codec; General description. [12] 3GPP TS 26.190: AMR Wideband Speech Codec; Transcoding functions. [13] 3GPP TS 26.173: AMR Wideband Speech Codec; ANSI-C code. [14] 3GPP TS 26.174: AMR Wideband Speech Codec; Test sequences. [15] 3GPP TS 26.193: AMR Wideband Speech Codec; Source Controlled Rate operation. [16] 3GPP TS 26.194: AMR Wideband Speech Codec; Voice Activity Detector (VAD). [17] 3GPP TS 26.192: AMR Wideband Speech Codec; Comfort Noise Aspects. [18] 3GPP TS 26.191: AMR Wideband Speech Codec; Error Concealment of Lost Frames. [19] 3GPP TS 26.201: AMR Wideband Speech Codec; Frame Structure. [20] 3GPP TR 26.901: AMR Wideband Speech Codec; Feasibility study report. [21] SCHROEDER (M.R.), ATAL (B.S.): Code-Excited Linear Prediction (CELP): High quality speech at very low bit rates, Proc. ICASSP'85, pp. 937-940, 1985. 64 ITU-T Rec. G.722.2 (07/2003) SERIES OF ITU-T RECOMMENDATIONS Series A Series B Series C Series D Series E Series F Series G Series H Series I Series J Series K Series L Series M Series N Series O Series P Series Q Series R Series S Series T Series U Series V Series X Series Y Series Z Organization of the work of ITU-T Means of expression: definitions, symbols, classification General telecommunication statistics General tariff principles Overall network operation, telephone service, service operation and human factors Non-telephone telecommunication services Transmission systems and media, digital systems and networks Audiovisual and multimedia systems Integrated services digital network Cable networks and transmission of television, sound programme and other multimedia signals Protection against interference Construction, installation and protection of cables and other elements of outside plant TMN and network maintenance: international transmission systems, telephone circuits, telegraphy, facsimile and leased circuits Maintenance: international sound programme and television transmission circuits Specifications of measuring equipment Telephone transmission quality, telephone installations, local line networks Switching and signalling Telegraph transmission Telegraph services terminal equipment Terminals for telematic services Telegraph switching Data communication over the telephone network Data networks and open system communications Global information infrastructure, Internet protocol aspects and Next Generation Networks Languages and general software aspects for telecommunication systems *26015* Printed in Switzerland Geneva, 2004

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