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LTC3780数据手册

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    LTC3780数据手册 电源  充电器

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    LTC3780 High Efficiency, Synchronoous, 4-Switch Buck-Boost Controller FEATURES DESCRIPTION n Single Inductor Architecture Allows VIN Above, Below or Equal to VOUT n Wide VIN Range: 4V to 36V Operation n Synchronous Rectification: Up to 98% Efficiency n Current Mode Control n ±1% Output Voltage Accuracy: 0.8V < VOUT < 30V n Phase-Lockable Fixed Frequency: 200kHz to 400kHz n Power Good Output Voltage Monitor n Internal LDO for MOSFET Supply n Quad N-Channel MOSFET Synchronous Drive n VOUT Disconnected from VIN During Shutdown n Adjustable Soft-Start Current Ramping n Foldback Output Current Limiting n Selectable Low Current Modes n Output Overvoltage Protection n Available in 24-Lead SSOP and Exposed Pad (5mm × 5mm) 32-Lead QFN Packages APPLICATIONS n Automotive Systems n Telecom Systems n DC Power Distribution Systems n High Power Battery-Operated Devices n Industrial Control The LTC®3780 is a high performance buck-boost switching regulator controller that operates from input voltages above, below or equal to the output voltage. The constant frequency current mode architecture allows a phase-lockable frequency of up to 400kHz. With a wide 4V to 30V (36V maximum) input and output range and seamless transfers between operating modes, the LTC3780 is ideal for automotive, telecom and battery-powered systems. The operating mode of the controller is determined through the FCB pin. For boost operation, the FCB mode pin can select among Burst Mode® operation, Discontinuous mode and forced continuous mode. During buck operation, the FCB mode pin can select among skip-cycle mode, discontinuous mode and forced continuous mode. Burst Mode operation and skip-cycle mode provide high efficiency operation at light loads while forced continuous mode and discontinuous mode operate at a constant frequency. Fault protection is provided by an output overvoltage comparator and internal foldback current limiting. A power good output pin indicates when the output is within 7.5% of its designed set point. , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. Burst Mode is a registered trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents including 5481178, 6304066, 5929620, 5408150, 6580258, patent pending on current mode architecture and protection TYPICAL APPLICATION VIN 5V TO 32V High Efficiency buck-Boost Converter 22μF + 50V 4.7μF CER A 0.1μF VIN PGOOD INTVCC TG2 TG1 BOOST2 BOOST1 SW2 SW1 B LTC3780 BG2 BG1 2200pF 20k ITH SS 0.1μF PLLIN RUN VOSENSE SGND FCB SENSE+ SENSE– PGND 1μF CER D 0.1μF C ON/OFF 105k 1% 7.5k 1% 0.010Ω 4.7μH 100μF + 16V CER 330μF 16V VOUT 12V 5A 3780 TA01 EFFICIENCY (%) POWER LOSS (W) Efficiency and Power Loss VOUT = 12V, ILOAD = 5A 100 10 9 95 8 90 7 6 85 5 4 80 3 75 2 1 70 05 0 10 15 20 25 30 35 VIN (V) 3780 TA01b 3780fd 1 LTC3780 ABSOLUTE MAXIMUM RATINGS (Note 1) Input Supply Voltage (VIN) ........................ –0.3V to 36V Topside Driver Voltages (BOOST1, boost2)..................................... –0.3V to 42V Switch Voltage (SW1, SW2) ........................ –5V to 36V INTVCC, EXTVCC, RUN, SS, (BOOST – SW1), (BOOST2 – SW2), PGOOD .......................... –0.3V to 7V PLLIN Voltage .......................................... –0.3V to 5.5V PLLFLTR Voltage....................................... –0.3V to 2.7V FCB, STBYMD Voltages........................ –0.3V to INTVCC ITH, VOSENSE Voltages .............................. –0.3V to 2.4V Peak Output Current <10μs (TG1, TG2, BG1, BG2).....3A INTVCC Peak Output Current ................................. 40mA Operating Temperature Range (Note 7) LTC3780E............................................. –40°C to 85°C LTC3780I............................................ –40°C to 125°C Junction Temperature (Note 2) ............................ 125°C Storage Temperature Range................... –65°C to 125°C Lead Temperature (Soldering, 10 sec) SSOP Only ........................................................ 300°C PIN CONFIGURATION PGOOD 1 SS 2 SENSE+ 3 SENSE– 4 ITH 5 VOSENSE 6 SGND 7 RUN 8 FCB 9 PLLFLTR 10 PLLIN 11 STBYMD 12 TOP VIEW 24 BOOST1 23 TG1 22 SW1 21 VIN 20 EXTVCC 19 INTVCC 18 BG1 17 PGND 16 BG2 15 SW2 14 TG2 13 BOOST2 G PACKAGE 24-LEAD PLASTIC SSOP TJMAX = 125°C, θJA = 130°C/W TOP VIEW NC SS PGOOD NC NC BOOST1 TG1 NC SENSE+ 1 SENSE– 2 ITH 3 VOSENSE 4 SGND 5 32 31 30 29 28 27 26 25 24 SW1 23 VIN 22 EXTVCC 21 INTVCC 33 20 BG1 RUN 6 19 PGND FCB 7 18 BG2 PLLFTR 8 17 SW2 9 10 11 12 13 14 15 16 NC PLLIN STBYMD NC NC BOOST2 TG2 NC UH PACKAGE 32-LEAD (5mm s 5mm) PLASTIC QFN TJMAX = 125°C, θJA = 34°C/W EXPOSED PAD (PIN 33) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION LTC3780EG#PBF LTC3780EG#TRPBF 24-Lead Plastic SSOP LTC3780IG#PBF LTC3780IG#TRPBF 24-Lead Plastic SSOP LTC3780EUH#PBF LTC3780EUH#TRPBF 3780 32-Lead (5mm × 5mm) Plastic QFN LTC3780IUH#PBF LTC3780IUH#TRPBF 3780I 32-Lead (5mm × 5mm) Plastic QFN LEAD BASED FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION LTC3780EG LTC3780EG#TR 24-Lead Plastic SSOP LTC3780IG LTC3780IG#TR 24-Lead Plastic SSOP LTC3780EUH LTC3780EUH#TR 3780 32-Lead (5mm × 5mm) Plastic QFN LTC3780IUH LTC3780IUH#TR 3780I 32-Lead (5mm × 5mm) Plastic QFN Consult LTC Marketing for parts specified with wider operating temperature ranges. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 2 TEMPERATURE RANGE –40°C to 85°C –40°C to 125°C –40°C to 85°C –40°C to 125°C TEMPERATURE RANGE –40°C to 85°C –40°C to 125°C –40°C to 85°C –40°C to 125°C 3780fd LTC3780 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX Main Control Loop VOSENSE Feedback Reference Voltage ITH = 1.2V, –40°C ≤ T ≤ 85°C (Note 3) –40°C ≤ T ≤ 125°C l 0.792 0.800 0.808 l 0.792 0.800 0.811 IVOSENSE VLOADREG Feedback Pin Input Current Output Voltage Load Regulation VREF(LINEREG) gm(EA) gm(GBW) IQ VFCB IFCB VBINHIBIT Reference Voltage Line Regulation Error Amplifier Transconductance Error Amplifier GBW Input DC Supply Current Normal Standby Shutdown Supply Current forced continuous Threshold forced continuous Pin Current Burst Inhibit (Constant Frequency) Threshold (Note 3) (Note 3) ΔITH = 1.2V to 0.7V ΔITH = 1.2V to 1.8V VIN = 4V to 30V, ITH = 1.2V (Note 3) ITH = 1.2V, Sink/Source = 3μA (Note 3) (Note 4) VRUN = 0V, VSTBYMD > 2V VRUN = 0V, VSTBYMD = Open VFCB = 0.85V Measured at FCB Pin –5 –50 l 0.1 0.5 l –0.1 –0.5 0.002 0.02 0.32 0.6 2400 1500 55 70 0.76 0.800 0.84 –0.30 –0.18 –0.1 5.3 5.5 UVLO VOVL ISENSE VSTBYMD(START) VSTBYMD(KA) DF MAX, boost Undervoltage Reset Feedback Overvoltage Lockout Sense Pins Total Source Current Start-Up Threshold Keep-Alive Power-On Threshold Maximum Duty Factor VIN Falling Measured at VOSENSE Pin VSENSE– = VSENSE+ = 0V VSTBYMD Rising VSTBYMD Rising, VRUN = 0V % switch C On l 3.8 4 0.84 0.86 0.88 –380 0.4 0.7 1.25 99 DF MAX, buck Maximum Duty Factor % switch A On (in Dropout) 99 VRUN(ON) ISS VSENSE(MAX) RUN Pin On Threshold Soft-Start Charge Current Maximum Current Sense Threshold VSENSE(MIN,BUCK) TG1, TG2 tr TG1, TG2 tf BG1, BG2 tr BG1, BG2 tf TG1/BG1 t1D Minimum Current Sense Threshold TG Rise Time TG Fall Time BG Rise Time BG Fall Time TG1 Off to BG1 On Delay, Switch C On Delay VRUN Rising VRUN = 2V Boost: VOSENSE = VREF – 50mV Buck: VOSENSE = VREF – 50mV Discontinuous Mode CLOAD = 3300pF (Note 5) CLOAD = 3300pF (Note 5) CLOAD = 3300pF (Note 5) CLOAD = 3300pF (Note 5) CLOAD = 3300pF Each Driver 1 0.5 l l –95 1.5 2 1.2 160 185 –130 –150 –6 50 45 45 55 80 BG1/TG1 t2D BG1 Off to TG1 On Delay, CLOAD = 3300pF Each Driver 80 Synchronous switch D On Delay TG2/BG2 t3D TG2 Off to BG2 On Delay, CLOAD = 3300pF Each Driver 80 Synchronous switch B On Delay BG2/TG2 t4D BG2 Off to TG2 On Delay, CLOAD = 3300pF Each Driver 80 Switch A On Delay Mode BG1 Off to BG2 On Delay, CLOAD = 3300pF Each Driver 90 Transition 1 Switch A On Delay UNITS V V nA % % %/V mS MHz μA μA μA V μA V V V μA V V % % V μA mV mV mV ns ns ns ns ns ns ns ns ns 3780fd 3 LTC3780 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Mode Transition 2 BG2 Off to BG1 On Delay, Synchronous switch D On Delay CLOAD = 3300pF Each Driver 90 ns tON(MIN,BOOST) Minimum On-Time for Main switch in boost Operation Switch C (Note 6) 200 ns tON(MIN,BUCK) Minimum On-Time for synchronous switch in buck Operation Switch B (Note 6) 180 ns Internal VCC Regulator VINTVCC Internal VCC Voltage ΔVLDO(LOADREG) Internal VCC Load Regulation VEXTVCC EXTVCC switchover Voltage ΔVEXTVCC(HYS) EXTVCC switchover Hysteresis ΔVEXTVCC EXTVCC switch Drop Voltage Oscillator and Phase-Locked Loop 7V < VIN < 30V, VEXTVCC = 5V ICC = 0mA to 20mA, VEXTVCC = 5V ICC = 20mA, VEXTVCC Rising ICC = 20mA, VEXTVCC = 6V l 5.7 6 6.3 V 0.2 2 % l 5.4 5.7 V 200 mV 150 300 mV fNOM fLOW fHIGH RPLLIN IPLLLPF Nominal Frequency Lowest Frequency Highest Frequency PLLIN Input Resistance Phase Detector Output Current PGOOD Output VPLLFLTR = 1.2V VPLLFLTR = 0V VPLLFLTR = 2.4V fPLLIN < fOSC fPLLIN > fOSC 260 300 330 kHz 170 200 220 kHz 340 400 440 kHz 50 kΩ –15 μA 15 μA ΔVFBH ΔVFBL ΔVFB(HYST) VPGL IPGOOD PGOOD Upper Threshold PGOOD Lower Threshold PGOOD Hysteresis PGOOD Low Voltage PGOOD Leakage Current VOSENSE Rising VOSENSE Falling VOSENSE Returning IPGOOD = 2mA VPGOOD = 5V 5.5 7.5 10 % –5.5 –7.5 –10 % 2.5 % 0.1 0.3 V ±1 μA Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: TJ for the QFN package is calculated from the temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD • 34°C/W) Note 3: The IC is tested in a feedback loop that servos VITH to a specified voltage and measures the resultant VOSENSE. Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: Rise and fall times are measured using 10% and 90% levels. Delay times are measured using 50% levels. Note 6: The minimum on-time condition is specified for an inductor peak-to-peak ripple current ≥ 40% of IMAX (see minimum on-time considerations in the Applications Information section). Note 7: The LTC3780E is guaranteed to meet performance specifications from 0°C to 85°C. Performance over the –40°C to 85°C operating temperature range is assured by design, characterization and correlation with statistical process controls. The LTC3780I is guaranteed over the –40°C to 125°C operating temperature range. 3780fd 4 LTC3780 TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C unless othewise noted. EFFICIENCY (%) SUPPLY CURRENT (μA) Efficiency vs Output Current (Boost Operation) 100 BURST 90 DCM 80 70 CCM 60 50 40 0.01 VIN = 6V VOUT = 12V 0.1 1 10 ILOAD (A) 3780 G01 Supply Current vs Input Voltage 2500 2000 VFCB = 0V 1500 STANDBY 1000 500 SHUTDOWN 0 0 5 10 15 20 25 30 35 INPUT VOLTAGE (V) 3780 G04 INTVCC and EXTVCC switch Voltage vs Temperature 6.05 6.00 INTVCC VOLTAGE 5.95 5.90 5.85 5.80 5.75 5.70 5.65 EXTVCC SWITCHOVER THRESHOLD 5.60 5.55 –50 –25 0 25 50 75 100 125 TEMPERATURE (°C) 3780 G07 EXTVCC SWITCH RESISTANCE (Ω) INTVCC VOLTAGE (V) EFFICIENCY (%) Efficiency vs Output Current 100 BURST 90 80 DCM CCM 70 60 50 40 0.01 VIN = 12V VOUT = 12V 0.1 1 10 ILOAD (A) 3780 G02 Internal 6V LDO Line Regulation 6.5 6.0 5.5 5.0 4.5 4.0 3.5 05 10 15 20 25 30 35 INPUT VOLTAGE (V) 3780 G05 EXTVCC switch Resistance vs Temperature 5 4 3 2 1 0 –50 –25 0 25 50 75 TEMPERATURE (°C) 100 125 3780 G08 NORMALIZED VOUT (%) EXTVCC VOLTAGE DROP (mV) EFFICIENCY (%) Efficiency vs Output Current (Buck Operation) 100 90 SC 80 CCM 70 DCM 60 50 40 0.01 VIN = 18V VOUT = 12V 0.1 1 10 ILOAD (A) 3780 G03 EXTVCC Voltage Drop 120 100 80 60 40 20 0 0 10 20 30 40 50 CURRENT (mA) 3780 G06 Load Regulation 0 –0.1 VIN = 18V –0.2 VIN = 12V –0.3 VIN = 6V –0.4 FCB = 0V VOUT = 12V –0.5 0 1 2 3 4 5 LOAD CURRENT (A) 3780 G09 INTVCC AND EXTVCC SWITCH VOLTAGE (V) 3780fd 5 LTC3780 TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C unless othewise noted. Continuous Current Mode (CCM, VIN = 6V, VOUT = 12V) SW2 10V/DIV SW1 10V/DIV VOUT 100mV/DIV Continuous Current Mode (CCM, VIN = 12V, VOUT = 12V) SW2 10V/DIV SW1 10V/DIV VOUT 100mV/DIV Continuous Current Mode (CCM, VIN = 18V, VOUT = 12V) SW2 10V/DIV SW1 10V/DIV VOUT 100mV/DIV IL 2A/DIV VIN = 6V VOUT = 12V 5μs/DIV 3780 G10 IL 2A/DIV VIN = 12V VOUT = 12V 5μs/DIV 3780 G11 IL 2A/DIV VIN = 18V VOUT = 12V 5μs/DIV 3780 G12 Burst Mode Operation (VIN = 6V, VOUT = 12V) SW2 10V/DIV SW1 10V/DIV VOUT 500mV/DIV IL 2A/DIV VIN = 6V VOUT = 12V 25μs/DIV Burst Mode Operation (VIN = 12V, VOUT = 12V) SW2 10V/DIV SW1 10V/DIV VOUT 200mV/DIV IL 2A/DIV 3780 G13 VIN = 12V VOUT = 12V 10μs/DIV Skip Cycle Mode (VIN = 18V, VOUT = 12V) SW2 10V/DIV SW1 10V/DIV VOUT 100mV/DIV 3780 G14 IL 1A/DIV VIN = 18V VOUT = 12V 2.5μs/DIV 3780 G15 Discontinuous Current Mode (DCM, VIN = 6V, VOUT = 12V) SW2 10V/DIV SW1 10V/DIV Discontinuous Current Mode (DCM, VIN = 12V, VOUT = 12V) SW2 10V/DIV SW1 10V/DIV Discontinuous Current Mode (DCM, VIN = 18V, VOUT = 12V) SW2 10V/DIV SW1 10V/DIV VOUT 100mV/DIV IL 1A/DIV VIN = 6V VOUT = 12V 5μs/DIV VOUT 100mV/DIV 3780 G16 IL 2A/DIV VIN = 12V VOUT = 12V 5μs/DIV VOUT 100mV/DIV 3780 G17 IL 1A/DIV VIN = 18V VOUT = 12V 2.5μs/DIV 3780 G18 3780fd 6 LTC3780 TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C unless othewise noted. FREQUENCY (kHz) ISENSE+ (mV) Oscillator Frequency vs Temperature 450 400 VPLLFLTR = 2.4V 350 VPLLFLTR = 1.2V 300 250 VPLLFLTR = 0V 200 150 100 50 0 –50 –25 0 25 50 75 TEMPERATURE (°C) 100 125 3780 G19 Maximum Current Sense Threshold vs Duty Factor (Boost) 180 160 140 120 100 0 20 40 60 80 100 DUTY FACTOR (%) 3780 G22 Peak Current Threshold vs VITH (Boost) 200 150 100 50 0 –50 –100 0 0.4 0.8 1.2 1.6 VITH (V) 1.8 2.4 3780 G25 ISENSE+ (mV) ISNESE+ (mV) UNDERVOLTAGE RESET (V) Undervoltage Reset vs Temperature 4.0 3.8 3.6 3.4 3.2 3.0 –50 –25 0 25 50 75 TEMPERATURE (°C) 100 125 3780 G20 Maximum Current Sense Threshold vs Duty Factor (Buck) 140 130 120 110 0 20 40 60 80 100 DUTY FACTOR (%) 3780 G23 Valley Current Threshold vs VITH (Buck) 100 50 0 –50 –100 –150 0 0.4 0.8 1.2 1.6 VITH (V) 2.0 2.4 3780 G26 ISENSE+ (mV) MAXIMUM ISNESE+ THRESHOLD (mV) ISENSE+ (mV) Minimum Current Sense Threshold vs Duty Factor (Buck) –20 –40 –60 –80 100 80 60 40 20 0 DUTY FACTOR (%) 3780 G21 Minimum Current Sense Threshold vs Temperature 200 BOOST 150 100 50 0 –50 –100 –150 –50 –25 BUCK 0 25 50 75 TEMPERATURE (°C) 100 125 3780 G24 Current Foldback Limit 200 160 120 BOOST BUCK 80 40 0 0 0.2 0.4 0.6 0.8 VOSENSE (V) 3780 G32 3780fd 7 ISENSE+ (mV) LTC3780 TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C unless othewise noted. Load Step Load Step Load Step VOUT 500mV/DIV VOUT 500mV/DIV VOUT 500mV/DIV IL 5A/DIV IL 5A/DIV IL 5A/DIV VIN = 18V 200μs/DIV VOUT = 12V LOAD STEP: 0A TO 5A CONTINUOUS MODE 3780 G27 Line Transient VIN = 12V 200μs/DIV VOUT = 12V LOAD STEP: 0A TO 5A CONTINUOUS MODE 3780 G28 VIN = 6V 200μs/DIV VOUT = 12V LOAD STEP: 0A TO 5A CONTINUOUS MODE Line Transient VIN 10V/DIV VOUT 500mV/DIV VIN 10V/DIV VOUT 500mV/DIV IL 1A/DIV VOUT = 12V 500μs/DIV ILOAD = 1A VIN STEP: 7V TO 20V CONTINUOUS MODE 3780 G30 IL 1A/DIV VOUT = 12V 500μs/DIV ILOAD = 1A VIN STEP: 20V TO 7V CONTINUOUS MODE 3780 G31 3780 G29 PIN FUNCTIONS (SSOP/QFN) PGOOD (Pin 1/Pin 30): Open-Drain Logic Output. PGOOD is pulled to ground when the output voltage is not within ±7.5% of the regulation point. SS (Pin 2/Pin 31): Soft-start reduces the input power sources’ surge currents by gradually increasing the controller’s current limit. A minimum value of 6.8nF is recommended on this pin. SENSE+ (Pin 3/Pin 1): The (+) Input to the Current Sense and Reverse Current Detect Comparators. The ITH pin voltage and built-in offsets between SENSE– and SENSE+ pins, in conjunction with RSENSE, set the current trip threshold. SENSE– (Pin 4/Pin 2): The (–) Input to the Current Sense and Reverse Current Detect Comparators. ITH (Pin 5/Pin 3): Current Control Threshold and Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. The voltage ranges from 0V to 2.4V. 3780fd 8 LTC3780 PIN FUNCTIONS (SSOP/QFN) VOSENSE (Pin 6/Pin 4): Error Amplifier Feedback Input. This pin connects the error amplifier input to an external resistor divider from VOUT. SGND (Pin 7/Pin 5): Signal Ground. All small-signal components and compensation components should connect to this ground, which should be connected to PGND at a single point. RUN (Pin 8/Pin 6): Run Control Input. Forcing the RUN pin below 1.5V causes the IC to shut down the switching regulator circuitry. There is a 100k resistor between the RUN pin and SGND in the IC. Do not apply >6V to this pin. FCB (Pin 9/Pin 7): Forced continuous Control Input. The voltage applied to this pin sets the operating mode of the controller. When the applied voltage is less than 0.8V, the forced continuous current mode is active. When this pin is allowed to float, the burst mode is active in boost operation and the skip cycle mode is active in buck operation. When the pin is tied to INTVCC, the constant frequency discontinuous current mode is active in buck or boost operation. PLLFLTR (Pin 10/Pin 8): The Phase-Locked Loop’s Lowpass Filter is Tied to This Pin. Alternatively, this pin can be driven with an AC or DC voltage source to vary the frequency of the internal oscillator. PLLIN (Pin 11/Pin 10): External Synchronization Input to Phase Detector. This pin is internally terminated to SGND with 50kΩ. The phase-locked loop will force the rising bottom gate signal of the controller to be synchronized with the rising edge of the PLLIN signal. STBYMD (Pin 12/Pin 11): LDO Control Pin. Determines whether the internal LDO remains active when the controller is shut down. See Operation section for details. If the STBYMD pin is pulled to ground, the SS pin is internally pulled to ground, preventing start-up and thereby providing a single control pin for turning off the controller. To keep the LDO active when RUN is low, for example to power a “wake up” circuit which controls the state of the RUN pin, bypass STBYMD to signal ground with a 0.1μF capacitor, or use a resistor divider from VIN to keep the pin within 2V to 5V. BOOST2, boost1 (Pins 13, 24/Pins 14, 27): Boosted Floating Driver Supply. The (+) terminal of the bootstrap capacitor CA and CB (Figure 11) connects here. The boost2 pin swings from a diode voltage below INTVCC up to VIN + INTVCC. The boost1 pin swings from a diode voltage below INTVCC up to VOUT + INTVCC. TG2, TG1 (Pins 14, 23/Pins 15, 26): Top Gate Drive. Drives the top N-channel MOSFET with a voltage swing equal to INTVCC superimposed on the switch node voltage SW. SW2, SW1 (Pins 15, 22/Pins 17, 24): Switch Node. The (–) terminal of the bootstrap capacitor CA and CB (Figure 11) connects here. The SW2 pin swings from a Schottky diode (external) voltage drop below ground up to VIN. The SW1 pin swings from a Schottky diode (external) voltage drop below ground up to VOUT. BG2, BG1 (Pins 16, 18/Pins 18, 20): Bottom Gate Drive. Drives the gate of the bottom N-channel MOSFET between ground and INTVCC. PGND (Pin 17/Pin 19): Power Ground. Connect this pin closely to the source of the bottom N-channel MOSFET, the (–) terminal of CVCC and the (–) terminal of CIN (Figure 11). INTVCC (Pin 19/Pin 21): Internal 6V Regulator Output. The driver and control circuits are powered from this voltage. Bypass this pin to ground with a minimum of 4.7μF low ESR tantalum or ceramic capacitor. EXTVCC (Pin 20/Pin 22): External VCC Input. When EXTVCC exceeds 5.7V, an internal switch connects this pin to INTVCC and shuts down the internal regulator so that the controller and gate drive power is drawn from EXTVCC. Do not exceed 7V at this pin and ensure that EXTVCC < VIN. VIN (Pin 21/Pin 23): Main Input Supply. Bypass this pin to SGND with an RC filter (1Ω, 0.1μF). Exposed Pad (Pin 33, QFN Only): This pin is SGND and must be soldered to PCB ground. 3780fd 9 LTC3780 BLOCK DIAGRAM STBYMD FCB 1.2V 4(VFB) 1.2μA SS RUN 100k VIN +6V VIN 5.7V + – EXTVCC INTVCC SGND 10 ILIM + – IREV + – ICMP + – INTVCC VIN BOOST2 FCB TG2 BUCK LOGIC SW2 INTVCC BG2 PGND BG1 FCB BOOST LOGIC INTVCC SW1 TG1 BOOST1 OV 0.86V INTVCC VOUT + – + – SLOPE VREF EA VOSENSE VFB SHDN RST 4(VFB) 0.80V ITH RUN/ SS SENSE+ SENSE– PLLIN PHASE DET FIN 50k 6V LDO REG – 0.86V + INTERNAL SUPPLY VOSENSE – PLLFLTR CLK OSCILLATOR PGOOD RLP CLP + 0.74V 3780 BD RSENSE 3780fd LTC3780 OPERATION MAIN CONTROL LOOP The LTC3780 is a current mode controller that provides an output voltage above, equal to or below the input voltage. The LTC proprietary topology and control architecture employs a current-sensing resistor in buck or boost modes. The sensed inductor current is controlled by the voltage on the ITH pin, which is the output of the amplifier EA. The VOSENSE pin receives the voltage feedback signal, which is compared to the internal reference voltage by the EA. The top MOSFET drivers are biased from floating booststrap capacitors CA and CB (Figure 11), which are normally recharged through an external diode when the top MOSFET is turned off. Schottky diodes across the synchronous switch D and synchronous switch B are not required, but provide a lower drop during the dead time. The addition of the Schottky diodes will typically improve peak efficiency by 1% to 2% at 400kHz. The main control loop is shut down by pulling the RUN pin low. When the RUN pin voltage is higher than 1.5V, an internal 1.2μA current source charges soft-start capacitor CSS at the SS pin. The ITH voltage is then clamped to the SS voltage while CSS is slowly charged during start-up. This “soft-start” clamping prevents abrupt current from being drawn from the input power supply. POWER SWITCH CONTROL Figure 1 shows a simplified diagram of how the four power switches are connected to the inductor, VIN, VOUT and GND. Figure 2 shows the regions of operation for the LTC3780 as a function of duty cycle D. The power switches are properly controlled so the transfer between modes is continuous. When VIN approaches VOUT, the buck-Boost region is reached; the mode-to-mode transition time is typically 200ns. Buck Region (VIN > VOUT) Switch D is always on and switch C is always off during this mode. At the start of every cycle, synchronous switch B is turned on first. Inductor current is sensed when synchronous switch B is turned on. After the sensed inductor current falls below the reference voltage, which is proportional to VITH, synchronous switch B is turned off VIN VOUT TG2 A D TG1 SW2 L SW1 BG2 B C BG1 RSENSE 3780 F01 Figure 1. Simplified Diagram of the Output switches 98% DMAX BOOST DMIN BOOST DMAX BUCK 3% DMIN BUCK A ON, B OFF PWM C, D SWITCHES BOOST REGION FOUR SWITCH PWM BUCK/BOOST REGION D ON, C OFF PWM A, B SWITCHES BUCK REGION 3780 F02 Figure 2. Operating Mode vs Duty Cycle 3780fd 11 LTC3780 OPERATION and switch A is turned on for the remainder of the cycle. CLOCK switches A and B will alternate, behaving like a typical synchronous buck regulator. The duty cycle of switch A SWITCH A increases until the maximum duty cycle of the converter SWITCH B in buck mode reaches DMAX_BUCK, given by: SWITCH C DMAX_BUCK = 100% – DBUCK-BOOST SWITCH D where DBUCK-BOOST = duty cycle of the buck-boost switch range: IL 3780 F04a DBUCK-BOOST = (200ns • f) • 100% (4a) buck-Boost Mode (VIN ≥ VOUT) and f is the operating frequency in Hz. Figure 3 shows typical buck mode waveforms. If VIN ap- CLOCK proaches VOUT, the buck-boost region is reached. SWITCH A CLOCK SWITCH A SWITCH B SWITCH C SWITCH D IL Figure 3. buck Mode (VIN > VOUT) 0V HIGH SWITCH B SWITCH C SWITCH D IL 3780 F03 (4b) buck-Boost Mode (VIN ≤ VOUT) Figure 4. buck-Boost Mode 3780 F04b Buck-Boost (VIN ≅ VOUT) When VIN is close to VOUT, the controller is in buck-boost mode. Figure 4 shows typical waveforms in this mode. Every cycle, if the controller starts with switches B and D turned on, switches A and C are then turned on. Finally, switches A and D are turned on for the remainder of the time. If the controller starts with switches A and C turned on, switches B and D are then turned on. Finally, switches A and D are turned on for the remainder of the time. Boost Region (VIN < VOUT) Switch A is always on and synchronous switch B is always off in boost mode. Every cycle, switch C is turned on first. Inductor current is sensed when synchronous switch C is turned on. After the sensed inductor current exceeds the reference voltage which is proportional to VITH, switch C is turned off and synchronous switch D is turned on for the remainder of the cycle. switches C and D will alternate, behaving like a typical synchronous boost regulator. 3780fd 12 LTC3780 OPERATION The duty cycle of switch C decreases until the minimum duty cycle of the converter in boost mode reaches DMIN_BOOST, given by: DMIN_BOOST = DBUCK-BOOST where DBUCK-BOOST is the duty cycle of the buck-boost switch range: DBUCK-BOOST = (200ns • f) • 100% and f is the operating frequency in Hz. Figure 5 shows typical boost mode waveforms. If VIN approaches VOUT, the buck-boost region is reached. CLOCK SWITCH A SWITCH B SWITCH C SWITCH D IL Figure 5. boost Mode (VIN < VOUT) HIGH 0V 3780 F05 LOW CURRENT OPERATION The FCB pin is used to select among three modes for both buck and boost operations by accepting a logic input. Figure 6 shows the different modes. FCB PIN 0V to 0.75V 0.85V to 5V BUCK MODE Force Continuous Mode Skip-Cycle Mode BOOST MODE Force Continuous Mode Burst Mode Operation >5.3V DCM with Constant Freq DCM with Constant Freq Figure 6. Different Operating Modes When the FCB pin voltage is lower than 0.8V, the controller behaves as a continuous, PWM current mode synchronous switching regulator. In boost mode, switch A is always on. switch C and synchronous switch D are alternately turned on to maintain the output voltage independent of direction of inductor current. Every ten cycles, switch A is forced off for about 300ns to allow boost capacitor CA (Figure 13) to recharge. In buck mode, synchronous switch D is always on. switch A and synchronous switch B are alternately turned on to maintain the output voltage independent of direction of inductor current. Every ten cycles, synchronous switch D is forced off for about 300ns to allow CB to recharge. This is the least efficient operating mode at light load, but may be desirable in certain applications. In this mode, the output can source or sink current. When the FCB pin voltage is below VINTVCC – 1V, but greater than 0.8V, the controller enters Burst Mode operation in boost operation or enters skip-cycle mode in buck operation. During boost operation, Burst Mode operation sets a minimum output current level before inhibiting the switch C and turns off synchronous switch D when the inductor current goes negative. This combination of requirements will, at low currents, force the ITH pin below a voltage threshold that will temporarily inhibit turn-on of power switches C and D until the output voltage drops. There is 100mV of hysteresis in the burst comparator tied to the ITH pin. This hysteresis produces output signals to the MOSFETs C and D that turn them on for several cycles, followed by a variable “sleep” interval depending upon the load current. The maximum output voltage ripple is limited to 3% of the nominal DC output voltage as determined by a resistive feedback divider. During buck operation at no load, switch A is turned on for its minimum on-time. This will not occur every clock cycle when the output load 3780fd 13 LTC3780 OPERATION current drops below 1% of the maximum designed load. The body diode of synchronous switch B or the Schottky diode, which is in parallel with switch B, is used to discharge the inductor current; switch B only turns on every ten clock cycles to allow CB to recharge. As load current is applied, switch A turns on every cycle, and its on-time begins to increase. At higher current, switch B turns on briefly after each turn-off of switch A. switches C and D remain off at light load, except to refresh CA (Figure 11) every 10 clock cycles. In Burst Mode operation/skip cycle mode, the output is prevented from sinking current. When the FCB pin voltage is tied to the INTVCC pin, the controller enters constant frequency discontinuous current mode (DCM). For boost operation, synchronous switch D is held off whenever the ITH pin is below a threshold voltage. In every cycle, switch C is used to charge inductor current. After the output voltage is high enough, the controller will enter continuous current buck mode for one cycle to discharge inductor current. In the following cycle, the controller will resume DCM boost operation. For buck operation, constant frequency discontinuous current mode sets a minimum negative inductor current level. synchronous switch B is turned off whenever inductor current is lower than this level. At very light loads, this constant frequency operation is not as efficient as Burst Mode operation or skip-cycle, but does provide lower noise, constant frequency operation. FREQUENCY SYNCHRONIZATION AND FREQUENCY SETUP The phase-locked loop allows the internal oscillator to be synchronized to an external source via the PLLIN pin. The phase detector output at the PLLFLTR pin is also the DC frequency control input of the oscillator. The frequency ranges from 200kHz to 400kHz, corresponding to a DC voltage input from 0V to 2.4V at PLLFLTR. When locked, the PLL aligns the turn on of the top MOSFET to the rising edge of the synchronizing signal. When PLLIN is left open, the PLLFLTR pin goes low, forcing the oscillator to its minimum frequency. INTVCC/EXTVCC Power Power for all power MOSFET drivers and most internal circuitry is derived from the INTVCC pin. When the EXTVCC pin is left open, an internal 6V low dropout linear regulator supplies INTVCC power. If EXTVCC is taken above 5.7V, the 6V regulator is turned off and an internal switch is turned on, connecting EXTVCC to INTVCC. This allows the INTVCC power to be derived from a high efficiency external source. POWER GOOD (PGOOD) PIN The PGOOD pin is connected to an open drain of an internal MOSFET. The MOSFET turns on and pulls the pin low when the output is not within ±7.5% of the nominal output level as determined by the resistive feedback divider. When the output meets the ±7.5% requirement, the MOSFET is turned off and the pin is allowed to be pulled up by an external resistor to a source of up to 7V. 3780fd 14 LTC3780 OPERATION FOLDBACK CURRENT Foldback current limiting is activated when the output voltage falls below 70% of its nominal level, reducing power waste. During start-up, foldback current limiting is disabled. INPUT UNDERVOLTAGE RESET The SS capacitor will be reset if the input voltage is allowed to fall below approximately 4V. The SS capacitor will attempt to charge through a normal soft-start ramp after the input voltage rises above 4V. OUTPUT OVERVOLTAGE PROTECTION An overvoltage comparator guards against transient overshoots (>7.5%) as well as other more serious conditions that may overvoltage the output. In this case, synchronous switch B and synchronous switch D are turned on until the overvoltage condition is cleared or the maximum negative current limit is reached. When inductor current is lower than the maximum negative current limit, synchronous switch B and synchronous switch D are turned off, and switch A and switch C are turned on until the inductor current reaches another negative current limit. If the comparator still detects an overvoltage condition, switch A and switch C are turned off, and synchronous switch B and synchronous switch D are turned on again. SHORT-CIRCUIT PROTECTION AND CURRENT LIMIT Switch A on-time is limited by output voltage. When output voltage is reduced and is lower than its nominal level, switch A on-time will be reduced. In every boost mode cycle, current is limited by a voltage reference, which is proportional to the ITH pin voltage. The maximum sensed current is limited to 160mV. In every buck mode cycle, the maximum sensed current is limited to 130mV. STANDBY MODE PIN The STBYMD pin is a three-state input that controls circuitry within the IC as follows: When the STBYMD pin is held at ground, the SS pin is pulled to ground. When the pin is left open, the internal SS current source charges the SS capacitor, allowing turn-on of the controller and activating necessary internal biasing. When the STBYMD pin is taken above 2V, the internal linear regulator is turned on independent of the state on the RUN and SS pins, providing an output power source for “wake-up” circuitry. Bypass the pin with a small capacitor (0.1μF) to ground if the pin is not connected to a DC potential. 3780fd 15 LTC3780 APPLICATIONS INFORMATION Figure 11 is a basic LTC3780 application circuit. External component selection is driven by the load requirement, and begins with the selection of RSENSE and the inductor value. Next, the power MOSFETs are selected. Finally, CIN and COUT are selected. This circuit can be configured for operation up to an input voltage of 36V. Selection of Operation Frequency The LTC3780 uses a constant frequency architecture and has an internal voltage controlled oscillator. The switching frequency is determined by the internal oscillator capacitor. This internal capacitor is charged by a fixed current plus an additional current that is proportional to the voltage applied to the PLLFLTR pin. The frequency of this oscillator can be varied over a 2-to-1 range. The PLLFLTR pin can be grounded to lower the frequency to 200kHz or tied to 2.4V to yield approximately 400kHz. When PLLIN is left open, the PLLFLTR pin goes low, forcing the oscillator to minimum frequency. A graph for the voltage applied to the PLLFLTR pin vs frequency is given in Figure 7. As the operating frequency is increased the gate charge losses will be higher, reducing efficiency. The maximum switching frequency is approximately 400kHz. OPERATING FREQUENCY (kHz) 450 400 350 300 250 200 150 100 50 0 0 0.5 1 1.5 2 2.5 PLLFLTR PIN VOLTAGE (V) 3780 F07 Figure 7. Frequency vs PLLFLTR Pin Voltage Inductor Selection The operating frequency and inductor selection are inter- related in that higher operating frequencies allow the use of smaller inductor and capacitor values. The inductor value has a direct effect on ripple current. The inductor current ripple ΔIL is typically set to 20% to 40% of the maximum inductor current at boost mode VIN(MIN). For a given ripple the inductance terms in continuous mode are as follows: ( ) LBOOST > VIN(MIN)2 • VOUT – VIN(MIN) • 100 ƒ • IOUT(MAX) • % Ripple • VOUT2 H, ( ) LBUCK > VOUT • VIN(MAX) – VOUT ƒ • IOUT(MAX) • % Ripple • • 100 VIN(MAX) H where: f is operating frequency, Hz % Ripple is allowable inductor current ripple, % VIN(MIN) is minimum input voltage, V VIN(MAX) is maximum input voltage, V VOUT is output voltage, V IOUT(MAX) is maximum output load current For high efficiency, choose an inductor with low core loss, such as ferrite and molypermalloy (from Magnetics, Inc.). Also, the inductor should have low DC resistance to reduce the I2R losses, and must be able to handle the peak inductor current without saturating. To minimize radiated noise, use a toroid, pot core or shielded bobbin inductor. RSENSE Selection and Maximum Output Current RSENSE is chosen based on the required output current. The current comparator threshold sets the peak of the inductor current in boost mode and the maximum inductor valley current in buck mode. In boost mode, the maximum average load current at VIN(MIN) is: IOUT(MAX,BOOST) = ⎛ ⎝⎜ 160mV RSENSE – ΔIL 2 ⎞ ⎠⎟ • VIN(MIN) VOUT 3780fd 16 LTC3780 APPLICATIONS INFORMATION where ΔIL is peak-to-peak inductor ripple current. In buck mode, the maximum average load current is: IOUT(MAX,BUCK) = 130mV RSENSE + ΔIL 2 Figure 8 shows how the load current (IMAXLOAD • RSENSE) varies with input and output voltage The maximum current sensing RSENSE value for the boost mode is: RSENSE(MAX) = 2 • 160mV • VIN(MIN) 2 •IOUT(MAX,BOOST) • VOUT + ΔIL,BOOST • VIN(MIN) The maximum current sensing RSENSE value for the buck mode is: RSENSE(MAX) = 2 • 2 • 130mV IOUT(MAX,BUCK) – ΔIL,BUCK The final RSENSE value should be lower than the calculated RSENSE(MAX) in both the boost and buck modes. A 20% to 30% margin is usually recommended. CIN and COUT Selection In boost mode, input current is continuous. In buck mode, input current is discontinuous. In buck mode, the selection of input capacitor CIN is driven by the need to filter the 160 150 IMAX(LOAD) • RSENSE (mV) 140 130 120 110 100 0.1 1 VIN/VOUT (V) 10 3780 F08 Figure 8. Load Current vs VIN/VOUT input square wave current. Use a low ESR capacitor sized to handle the maximum RMS current. For buck operation, the input RMS current is given by: IRMS ≈ IOUT(MAX) • VOUT VIN • VIN – 1 VOUT This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT(MAX)/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life which makes it advisable to derate the capacitor. In boost mode, the discontinuous current shifts from the input to the output, so COUT must be capable of reducing the output voltage ripple. The effects of ESR (equivalent series resistance) and the bulk capacitance must be considered when choosing the right capacitor for a given output ripple voltage. The steady ripple due to charging and discharging the bulk capacitance is given by: ( ) Ripple (Boost,Cap) = IOUT(MAX) • VOUT – VIN(MIN) V COUT • VOUT • f ( ) Ripple (Buck,Cap) = IOUT(MAX) • VIN(MAX) – VOUT V COUT • VIN(MAX) • f where COUT is the output filter capacitor. The steady ripple due to the voltage drop across the ESR is given by: ΔVBOOST,ESR = IL(MAX,BOOST) • ESR ΔVBUCK,ESR = IL(MAX,BUCK) • ESR Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, special polymer, aluminum electrolytic and ceramic capacitors are all available in surface mount packages. Ceramic capacitors have excellent low ESR characteristics but can have a high voltage coefficient. Capacitors are now available with low ESR and high ripple current ratings, such as OS-CON and POSCAP. 3780fd 17 LTC3780 APPLICATIONS INFORMATION Power MOSFET Selection and Efficiency Considerations The LTC3780 requires four external N-channel power MOSFETs, two for the top switches (switch A and D, shown in Figure 1) and two for the bottom switches (switch B and C shown in Figure 1). Important parameters for the power MOSFETs are the breakdown voltage VBR,DSS, threshold voltage VGS,TH, on-resistance RDS(ON), reverse transfer capacitance CRSS and maximum current IDS(MAX). The drive voltage is set by the 6V INTVCC supply. Consequently, logic-level threshold MOSFETs must be used in LTC3780 applications. If the input voltage is expected to drop below 5V, then the sub-logic threshold MOSFETs should be considered. In order to select the power MOSFETs, the power dissipated by the device must be known. For switch A, the maximum power dissipation happens in boost mode, when it remains on all the time. Its maximum power dissipation at maximum output current is given by: PA,BOOST = ⎛ ⎝⎜ VOUT VIN • IOUT(MAX)⎞⎠⎟ 2 • ρT • RDS(ON) where ρT is a normalization factor (unity at 25°C) accounting for the significant variation in on-resistance with temperature, typically about 0.4%/°C as shown in Figure 9. For a maximum junction temperature of 125°C, using a value ρT = 1.5 is reasonable. 2.0 RT NORMALIZED ON-RESISTANCE (Ω) 1.5 1.0 0.5 0 –50 0 50 100 150 JUNCTION TEMPERATURE (°C) 3780 F09 Figure 9. Normalized RDS(ON) vs Temperature Switch B operates in buck mode as the synchronous rectifier. Its power dissipation at maximum output current is given by: PB,BUCK = VIN – VOUT VIN • IOUT(MAX)2 • ρT • RDS(ON) Switch C operates in boost mode as the control switch. Its power dissipation at maximum current is given by: ( ) PC,BOOST = VOUT – VIN VIN2 VOUT • IOUT(MAX)2 • ρT • RDS(ON) + k • VOUT3 • IOUT(MAX) VIN • CRSS • f where CRSS is usually specified by the MOSFET manufacturers. The constant k, which accounts for the loss caused by reverse recovery current, is inversely proportional to the gate drive current and has an empirical value of 1.7. For switch D, the maximum power dissipation happens in boost mode, when its duty cycle is higher than 50%. Its maximum power dissipation at maximum output current is given by: PD,BOOST = VIN VOUT • ⎛ ⎝⎜ VOUT VIN ⎞2 • IOUT(MAX) ⎠⎟ • ρT • RDS(ON) For the same output voltage and current, switch A has the highest power dissipation and switch B has the lowest power dissipation unless a short occurs at the output. From a known power dissipated in the power MOSFET, its junction temperature can be obtained using the following formula: TJ = TA + P • RTH(JA) The RTH(JA) to be used in the equation normally includes the RTH(JC) for the device plus the thermal resistance from the case to the ambient temperature (RTH(JC)). This value of TJ can then be compared to the original, assumed value used in the iterative calculation process. 3780fd 18 LTC3780 APPLICATIONS INFORMATION Schottky Diode (D1, D2) Selection and Light Load Operation The Schottky diodes D1 and D2 shown in Figure 1 conduct during the dead time between the conduction of the power MOSFET switches. They are intended to prevent the body diode of synchronous switches B and D from turning on and storing charge during the dead time. In particular, D2 significantly reduces reverse recovery current between switch D turn-off and switch C turn-on, which improves converter efficiency and reduces switch C voltage stress. In order for the diode to be effective, the inductance between it and the synchronous switch must be as small as possible, mandating that these components be placed adjacently. In buck mode, when the FCB pin voltage is 0.85 < VFCB < 5V, the converter operates in skip-cycle mode. In this mode, synchronous switch B remains off until the inductor peak current exceeds one-fifth of its maximum peak current. As a result, D1 should be rated for about one-half to one-third of the full load current. In boost mode, when the FCB pin voltage is higher than 5.3V, the converter operates in discontinuous current mode. In this mode, synchronous switch D remains off until the inductor peak current exceeds one-fifth of its maximum peak current. As a result, D2 should be rated for about one-third to one-fourth of the full load current. In buck mode, when the FCB pin voltage is higher than 5.3V, the converter operates in constant frequency discontinuous current mode. In this mode, synchronous switch B remains on until the inductor valley current is lower than the sense voltage representing the minimum negative inductor current level (VSENSE = –5mV). Both switch A and B are off until next clock signal. In boost mode, when the FCB pin voltage is 0.85 < VFCB < 5.3V, the converter operates in Burst Mode operation. In this mode, the controller clamps the peak inductor current to approximately 20% of the maximum inductor current. The output voltage ripple can increase during Burst Mode operation. INTVCC Regulator An internal P-channel low dropout regulator produces 6V at the INTVCC pin from the VIN supply pin. INTVCC powers the drivers and internal circuitry within the LTC3780. The INTVCC pin regulator can supply a peak current of 40mA and must be bypassed to ground with a minimum of 4.7μF tantalum, 10μF special polymer or low ESR type electrolytic capacitor. A 1μF ceramic capacitor placed directly adjacent to the INTVCC and PGND IC pins is highly recommended. Good bypassing is necessary to supply the high transient current required by MOSFET gate drivers. Higher input voltage applications in which large MOSFETs are being driven at high frequencies may cause the maximum junction temperature rating for the LTC3780 to be exceeded. The system supply current is normally dominated by the gate charge current. Additional external loading of the INTVCC also needs to be taken into account for the power dissipation calculations. The total INTVCC current can be supplied by either the 6V internal linear regulator or by the EXTVCC input pin. When the voltage applied to the EXTVCC pin is less than 5.7V, all of the INTVCC current is supplied by the internal 6V linear regulator. Power dissipation for the IC in this case is VIN • IINTVCC, and overall efficiency is lowered. The junction temperature can be estimated by using the equations given in Note 2 of the Electrical Characteristics. For example, a typical application operating in continuous current mode might draw 24mA from a 24V supply when not using the EXTVCC pin: TJ = 70°C + 24mA • 24V • 34°C/W = 90°C Use of the EXTVCC input pin reduces the junction temperature to: TJ = 70°C + 24mA • 6V • 34°C/W = 75°C To prevent maximum junction temperature from being exceeded, the input supply current must be checked operating in continuous mode at maximum VIN. 3780fd 19 LTC3780 APPLICATIONS INFORMATION EXTVCC Connection The LTC3780 contains an internal P-channel MOSFET switch connected between the EXTVCC and INTVCC pins. When the voltage applied to EXTVCC rises above 5.7V, the internal regulator is turned off and a switch connects the EXTVCC pin to the INTVCC pin thereby supplying internal power. The switch remains closed as long as the voltage applied to EXTVCC remains above 5.5V. This allows the MOSFET driver and control power to be derived from the output when (5.7V < VOUT < 7V) and from the internal regulator when the output is out of regulation (start-up, short-circuit). If more current is required through the EXTVCC switch than is specified, an external Schottky diode can be interposed between the EXTVCC and INTVCC pins. Ensure that EXTVCC ≤ VIN. The following list summarizes the three possible connections for EXTVCC: 1. EXTVCC left open (or grounded). This will cause INTVCC to be powered from the internal 6V regulator at the cost of a small efficiency penalty. 2. EXTVCC connected directly to VOUT (5.7V < VOUT < 7V). This is the normal connection for a 6V regulator and provides the highest efficiency. 3. EXTVCC connected to an external supply. If an external supply is available in the 5.5V to 7V range, it may be used to power EXTVCC provided it is compatible with the MOSFET gate drive requirements. Output Voltage The LTC3780 output voltage is set by an external feedback resistive divider carefully placed across the output capacitor. The resultant feedback signal is compared with the internal precision 0.800V voltage reference by the error amplifier. The output voltage is given by the equation: VOUT = 0.8V • ⎛ ⎝⎜ 1+ R2 ⎞ R1⎠⎟ Topside MOSFET Driver Supply (CA, DA, CB, DB) Referring to Figure 11, the external bootstrap capacitors CA and CB connected to the boost1 and boost2 pins supply 20 the gate drive voltage for the topside MOSFET switches A and D. When the top MOSFET switch A turns on, the switch node SW2 rises to VIN and the boost2 pin rises to approximately VIN + INTVCC. When the bottom MOSFET switch B turns on, the switch node SW2 drops to low and the boost capacitor CB is charged through DB from INTVCC. When the top MOSFET switch D turns on, the switch node SW1 rises to VOUT and the boost1 pin rises to approximately VOUT + INTVCC. When the bottom MOSFET switch C turns on, the switch node SW1 drops to low and the boost capacitor CA is charged through DA from INTVCC. The boost capacitors CA and CB need to store about 100 times the gate charge required by the top MOSFET switch A and D. In most applications a 0.1μF to 0.47μF, X5R or X7R dielectric capacitor is adequate. Run Function The RUN pin provides simple ON/OFF control for the LTC3780. Driving the RUN pin above 1.5V permits the controller to start operating. Pulling RUN below 1.5V puts the LTC3780 into low current shutdown. Do not apply more than 6V to the RUN pin. Soft-Start Function Soft-start reduces the input power sources’ surge currents by gradually increasing the controller’s current limit (proportional to an internally buffered and clamped equivalent of VITH). An internal 1.2μA current source charges up the CSS capacitor. As the voltage on SS increases from 0V to 2.4V, the internal current limit rises from 0V/RSENSE to 150mV/RSENSE. The output current limit ramps up slowly, taking 1.5s/μF to reach full current. The output current thus ramps up slowly, eliminating the starting surge current required from the input power supply. TIRMP = 2.4V 1.2μA • CSS = (1.5s/μF) • CSS Do not apply more than 6V to the SS pin. Current foldback is disabled during soft-start until the voltage on CSS reaches 2V. Make sure CSS is large enough when there is loading during start-up. 3780fd LTC3780 APPLICATIONS INFORMATION The Standby Mode (STBYMD) Pin Function The Standby mode (STBYMD) pin provides several choices for start-up and standby operational modes. If the pin is pulled to ground, the SS pin is internally pulled to ground, preventing start-up and thereby providing a single control pin for turning off the controller. If the pin is left open or bypassed to ground with a capacitor, the SS pin is internally provided with a starting current, permitting external control for turning on the controller. If the pin is connected to a voltage greater than 1.25V, the internal regulator (INTVCC) will be on even when the controller is shut down (RUN pin voltage < 1.5V). In this mode, the onboard 6V linear regulator can provide power to keep-alive functions such as a keyboard controller. Fault Conditions: Overvoltage Protection A comparator monitors the output for overvoltage conditions. The comparator (OV) detects overvoltage faults greater than 7.5% above the nominal output voltage. When the condition is sensed, switches A and C are turned off, and switches B and D are turned on until the overvoltage condition is cleared. During an overvoltage condition, a negative current limit (VSENSE = –60mV) is set to limit negative inductor current. When the sensed current inductor current is lower than –60mV, switch A and C are turned on, and switch B and D are turned off until the sensed current is higher than –20mV. If the output is still in overvoltage condition, switch A and C are turned off, and switch B and D are turned on again. Fault Conditions: Current Limit and Current Foldback The maximum inductor current is inherently limited in a current mode controller by the maximum sense voltage. In boost mode, maximum sense voltage and the sense resistance determines the maximum allowed inductor peak current, which is: IL(MAX,BOOST) = 160mV RSENSE In buck mode, maximum sense voltage and the sense resistance determines the maximum allowed inductor valley current, which is: IL(MAX,BUCK) = 130mV RSENSE To further limit current in the event of a short circuit to ground, the LTC3780 includes foldback current limiting. If the output falls by more than 30%, then the maximum sense voltage is progressively lowered to about one third of its full value. Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Although all dissipative elements in circuit produce losses, four main sources account for most of the losses in LTC3780 circuits: 1. DC I2R losses. These arise from the resistances of the MOSFETs, sensing resistor, inductor and PC board traces and cause the efficiency to drop at high output currents. 2. Transition loss. This loss arises from the brief amount of time switch A or switch C spends in the saturated region during switch node transitions. It depends upon the input voltage, load current, driver strength and MOSFET capacitance, among other factors. The loss is significant at input voltages above 20V and can be estimated from: Transition Loss ≈ 1.7A–1 • VIN2 • IOUT • CRSS • f where CRSS is the reverse transfer capacitance. 3780fd 21 LTC3780 APPLICATIONS INFORMATION 3. INTVCC current. This is the sum of the MOSFET driver and control currents. This loss can be reduced by supplying INTVCC current through the EXTVCC pin from a high efficiency source, such as an output derived boost network or alternate supply if available. 4. CIN and COUT loss. The input capacitor has the difficult job of filtering the large RMS input current to the regulator in buck mode. The output capacitor has the more difficult job of filtering the large RMS output current in boost mode. Both CIN and COUT are required to have low ESR to minimize the AC I2R loss and sufficient capacitance to prevent the RMS current from causing additional upstream losses in fuses or batteries. 5. Other losses. Schottky diode D1 and D2 are responsible for conduction losses during dead time and light load conduction periods. Inductor core loss occurs predominately at light loads. Switch C causes reverse recovery current loss in boost mode. When making adjustments to improve efficiency, the input current is the best indicator of changes in efficiency. If you make a change and the input current decreases, then the efficiency has increased. If there is no change in input current, then there is no change in efficiency. Design Example As a design example, assume VIN = 5V to 18V (12V nominal), VOUT = 12V (5%), IOUT(MAX) = 5A and f = 400kHz. Set the PLLFLTR pin at 2.4V for 400kHz operation. The inductance value is chosen first based on a 30% ripple current assumption. In buck mode, the ripple current is: ΔIL,BUCK = VOUT f •L ⎛ • ⎝⎜ 1– VOUT VIN ⎞ ⎠⎟ IRIPPLE,BUCK = ΔIL,BUCK • 100 IOUT % The highest value of ripple current occurs at the maximum input voltage. In boost mode, the ripple current is: ΔIL,BOOST = VIN f •L • ⎛⎝⎜1– VIN VOUT ⎞ ⎠⎟ IRIPPLE,BOOST = ΔIL,BOOST IIN • 100 % The highest value of ripple current occurs at VIN = VOUT/2. A 6.8μH inductor will produce 11% ripple in boost mode (VIN = 6V) and 29% ripple in buck mode (VIN = 18V). The RSENSE resistor value can be calculated by using the maximum current sense voltage specification with some accommodation for tolerances. RSENSE = 2 2 • 160mV • • IOUT(MAX,BOOST) • VOUT VIN(MIN) + ΔIL,BOOST • VIN(MIN) Select an RSENSE of 10mΩ. Output voltage is 12V. Select R1 as 20k. R2 is: R2 = VOUT • R1 – R1 0.8 Select R2 as 280k. Both R1 and R2 should have a tolerance of no more than 1%. Next, choose the MOSFET switches. A suitable choice is the Siliconix Si4840 (RDS(ON) = 0.009Ω (at VGS = 6V), CRSS = 150pF, θJA = 40°C/W). The maximum power dissipation of switch A occurs in boost mode when switch A stays on all the time. Assuming a junction temperature of TJ = 150°C with ρ150°C = 1.5, the power dissipation at VIN = 5V is: PA,BOOST = ⎛ ⎝⎜ 12 5 • 5⎞⎠⎟ 2 • 1.5 • 0.009 = 1.94W 3780fd 22 LTC3780 APPLICATIONS INFORMATION Double-check the TJ in the MOSFET with 70°C ambient temperature: TJ = 70°C + 1.94W • 40°C/W = 147.6°C The maximum power dissipation of switch B occurs in buck mode. Assuming a junction temperature of TJ = 80°C with ρ80°C = 1.2, the power dissipation at VIN = 18V is: PB,BUCK = 18 – 12 18 • 52 • 1.2 • 0.009 = 90mW Double-check the TJ in the MOSFET at 70°C ambient temperature: TJ = 70°C + 0.09W • 40°C/W = 73.6°C The maximum power dissipation of switch C occurs in boost mode. Assuming a junction temperature of TJ = 110°C with ρ110°C = 1.4, the power dissipation at VIN = 5V is: PC,BOOST = (12 – 5) 52 • 12 • 52 • 1.4 • 0.009 + 2 • 123 • 5 • 150p • 400k = 1.27W 5 Double-check the TJ in the MOSFET at 70°C ambient temperature: TJ = 70°C + 1.08W • 40°C/W = 113°C The maximum power dissipation of switch D occurs in boost mode when its duty cycle is higher than 50%. Assuming a junction temperature of TJ = 100°C with ρ100°C = 1.35, the power dissipation at VIN = 5V is: PD,BOOST = 5 12 • ⎛ ⎝⎜ 12 5 • 5⎞⎠⎟ 2 • 1.35 • 0.009 = 0.73W Double-check the TJ in the MOSFET at 70°C ambient temperature: TJ = 70°C + 0.73W • 40°C/W = 99°C CIN is chosen to filter the square current in buck mode. In this mode, the maximum input current peak is: IIN,PEAK(MAX,BUCK) = 5 • ⎛ ⎝⎜ 1+ 29% ⎞ 2 ⎠⎟ = 5.7A A low ESR (10mΩ) capacitor is selected. Input voltage ripple is 57mV (assuming ESR dominate ripple). COUT is chosen to filter the square current in boost mode. In this mode, the maximum output current peak is: IOUT,PEAK(MAX,BOOST) = 12 5 • 5 • ⎛ ⎝⎜ 1+ 11% ⎞ 2 ⎠⎟ = 10.6A A low ESR (5mΩ) capacitor is suggested. This capacitor will limit output voltage ripple to 53mV (assuming ESR dominate ripple). PC Board Layout Checklist The basic PC board layout requires a dedicated ground plane layer. Also, for high current, a multilayer board provides heat sinking for power components. • The ground plane layer should not have any traces and it should be as close as possible to the layer with power MOSFETs. • Place CIN, switch A, switch B and D1 in one compact area. Place COUT, switch C, switch D and D2 in one compact area. One layout example is shown in Figure 10. VIN SW2 SW1 VOUT L D2 QA QD D1 QB QC CIN COUT RSENSE LTC3780 CKT Figure 10. Switches Layout GND 3780 F10 3780fd 23 LTC3780 APPLICATIONS INFORMATION • Use immediate vias to connect the components (including the LTC3780’s SGND and PGND pins) to the ground plane. Use several large vias for each power component. • Use planes for VIN and VOUT to maintain good voltage filtering and to keep power losses low. • Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of power components. Connect the copper areas to any DC net (VIN or GND). • Segregate the signal and power grounds. All small signal components should return to the SGND pin at one point, which is then tied to the PGND pin close to the sources of switch B and switch C. • Place switch B and switch C as close to the controller as possible, keeping the PGND, BG and SW traces short. • Keep the high dV/dT SW1, SW2, boost1, boost2, TG1 and TG2 nodes away from sensitive small-signal nodes. • The path formed by switch A, switch B, D1 and the CIN capacitor should have short leads and PC trace lengths. The path formed by switch C, switch D, D2 and the COUT capacitor also should have short leads and PC trace lengths. • The output capacitor (–) terminals should be connected as close as possible the (–) terminals of the input capacitor. • Connect the top driver boost capacitor CA closely to the boost1 and SW1 pins. Connect the top driver boost capacitor CB closely to the boost2 and SW2 pins. • Connect the input capacitors CIN and output capacitors COUT closely to the power MOSFETs. These capacitors carry the MOSFET AC current in boost and buck mode. • Connect VOSENSE pin resistive dividers to the (+) terminals of COUT and signal ground. A small VOSENSE bypass capacitor may be connected closely to the LTC3780 SGND pin. The R2 connection should not be along the high current or noise paths, such as the input capacitors. • Route SENSE– and SENSE+ leads together with minimum PC trace spacing. Avoid sense lines pass through noisy area, such as switch nodes. The filter capacitor between SENSE+ and SENSE– should be as close as possible to the IC. Ensure accurate current sensing with Kelvin connections at the SENSE resistor. One layout example is shown in Figure 12. • Connect the ITH pin compensation network close to the IC, between ITH and the signal ground pins. The capacitor helps to filter the effects of PCB noise and output voltage ripple voltage from the compensation loop. • Connect the INTVCC bypass capacitor, CVCC, close to the IC, between the INTVCC and the power ground pins. This capacitor carries the MOSFET drivers’ current peaks. An additional 1μF ceramic capacitor placed immediately next to the INTVCC and PGND pins can help improve noise performance substantially. 3780fd 24 APPLICATIONS INFORMATION CSS CC2 C CC1 RC RR R1 R2 RPU 1 24 PGOOD BOOST1 2 SS 23 TG1 LTC3780 3 SENSE+ 22 SW1 4 SENSE– 21 VIN 5 20 ITH EXTVCC 6 19 VOSENSE INTVCC 7 18 SGND BG1 8 RUN 17 PGND 9 FCB 16 BG2 10 15 PLLFLTR SW2 VPULLUP CA DA CF CVCC RSENSE DB 11 14 fIN PLLIN TG2 12 13 STBYMD BOOST2 CB RIN LTC3780 D D2 VOUT COUT C L D1 B A CIN VIN 3780 F11 Figure 11. LTC3780 Layout Diagram PGND RSENSE 13 14 15 16 17 18 19 20 21 22 23 24 12 11 10 9 8 7 6 5 4 3 2 1 C RR SGND 3780 F12 Figure 12. Sense Lines Layout 3780fd 25 LTC3780 PACKAGE DESCRIPTION G Package 24-Lead Plastic SSOP (5.3mm) (Reference LTC DWG # 05-08-1640) 7.8 – 8.2 1.25 p0.12 5.3 – 5.7 7.90 – 8.50* (.311 – .335) 24 23 22 21 20 19 18 17 16 15 14 13 7.40 – 8.20 (.291 – .323) 0.42 p0.03 RECOMMENDED SOLDER PAD LAYOUT 0.65 BSC 5.00 – 5.60** (.197 – .221) 0o – 8o 1 2 3 4 5 6 7 8 9 10 11 12 2.0 (.079) MAX 0.09 – 0.25 (.0035 – .010) 0.55 – 0.95 (.022 – .037) NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS 2. DIMENSIONS ARE IN MILLIMETERS (INCHES) 3. DRAWING NOT TO SCALE *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED .152mm (.006") PER SIDE **DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED .254mm (.010") PER SIDE 0.65 (.0256) BSC 0.22 – 0.38 (.009 – .015) TYP 0.05 (.002) MIN G24 SSOP 0204 3780fd 26 PACKAGE DESCRIPTION UH Package 32-Lead Plastic QFN (5mm × 5mm) (Reference LTC DWG # 05-08-1693) LTC3780 0.70 p0.05 5.50 p0.05 4.10 p0.05 3.50 REF (4 SIDES) 3.45 p 0.05 3.45 p 0.05 PACKAGE OUTLINE 0.25 p 0.05 0.50 BSC RECOMMENDED SOLDER PAD LAYOUT APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 5.00 p 0.10 (4 SIDES) 0.75 p 0.05 PIN 1 TOP MARK (NOTE 6) R = 0.05 TYP 0.00 – 0.05 BOTTOM VIEW—EXPOSED PAD R = 0.115 TYP 31 32 3.50 REF (4-SIDES) 3.45 p 0.10 3.45 p 0.10 PIN 1 NOTCH R = 0.30 TYP OR 0.35 s 45o CHAMFER 0.40 p 0.10 1 2 0.200 REF NOTE: 1. DRAWING PROPOSED TO BE A JEDEC PACKAGE OUTLINE M0-220 VARIATION WHHD-(X) (TO BE APPROVED) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 0.25 p 0.05 0.50 BSC (UH32) QFN 0406 REV D Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 3780fd 27 LTC3780 TYPICAL APPLICATION CSS 0.022μF CC2 47pF CC1 0.01μF RC 100k R1 8.06k, 1% R2 113k, 1% ON/OFF 10k 68pF RPU VPULLUP CA 0.22μF 1 24 PGOOD BOOST1 2 SS 23 TG1 LTC3780 3 SENSE+ 22 SW1 DA BO540W 4 SENSE– 21 VIN 5 20 ITH EXTVCC 6 19 VOSENSE INTVCC 7 SGND 8 RUN 9 FCB 10 PLLFLTR BG1 18 PGND 17 BG2 16 SW2 15 DB BO540W CF 0.1μF CVCC 4.7μF 9mΩ 11 PLLIN TG2 14 2V 12 STBYMD BOOST2 13 CSTBYMD 0.01μF 10Ω 100Ω 100Ω CB 0.22μF 22μF 16V, X7R s3 + D Si7884DP D2 B320A VOUT 12V 5A COUT 330μF 16V C Si7884DP L 4.7μH B D1 Si7884DP B340A A Si7884DP CIN + 22μF 35V 3.3μF 50V, X5R s3 3780 TA02 VIN 5V TO 32V Figure 13. LTC3789 12V/5A, buck-Boost Regulator RELATED PARTS PART NUMBER LTC1871 LT1339 LTC1702A LTC1735 LTC1778 LT1956 LT3010 LT3430/LT3431 LT3433 DESCRIPTION Boost, Flyback, SEPIC Controller High Power Synchronous DC/DC Controller Dual, 2-Phase Synchronous DC/DC Controller Synchronous Step-Down DC/DC Controller No RSENSE™ Synchronous DC/DC Controller Monolithic 1.5A, 500kHz Step-Down Regulator 50mA, 3V to 80V Linear Regulator Monolithic 3A, 200kHz/500kHz Step-Down Regulator Monolithic Step-Up/Step-Down DC/DC Converter LTC3443 Monolithic Buck-Boost Converter LTC3703 100V Synchronous DC/DC Controller LTC3785 2.7V to 10V Synchronous buck-Boost Controller LTC4440 High Voltage, High Side MOSFET Driver No RSENSE is a trademark of Linear Technology Corporation COMMENTS No RSENSE, 2.5V ≤ VIN ≤ 36V, 92% Duty Cycle VIN Up to 60V, Drivers 10,000pF Gate Capacitance, IOUT ≤ 20A 550kHz Operation, No RSENSE, 3V ≤ VIN ≤ 7V, IOUT ≤ 20A 3.5V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤6V, Current Mode, IOUT ≤ 20A 4V ≤ VIN ≤ 36V, Fast Transient Response, Current Mode, IOUT ≤ 20A 5.5V ≤ VIN ≤ 60V, 2.5mA Supply Current, 16-Pin SSOP 1.275V ≤ VOUT ≤ 60V, No Protection Diode Required, 8-Lead MSOP 5.5V ≤ VIN ≤ 60V, 0.1Ω Saturation Switch, 16-Pin SSOP 4V ≤ VIN ≤ 60V, 500mA Switch, Automatic Step-Up/Step-Down, Single Inductor 2.4V ≤ VIN ≤ 5.5V, 96% Efficiency, 600kHz Operation, 2A Switch VIN Up to 100V, 9.3V to 15V Gate Drive Supply 1MHz Operation, No RSENSE, High Efficiency, Burst Mode Operation, IOUT ≤ 10A 100V, 2.4A Pull-Up, 1.6Ω Pulldown, SOT-23, MSOP 28 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com 3780fd LT 1008 REV D • PRINTED IN USA © LINEAR TECHNOLOGY CORPORATION 2005

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