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    iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation  1.0 Features ●● Primary-side feedback eliminates opto-isolators and simplifies design ●● Quasi-resonant operation for highest overall efficiency ●● EZ-EMI ® design to easily meet global EMI standards ●● Up to 130 kHz switching frequency enables small adapter size ●● Very tight output voltage regulation ●● No external compensation components required ●● Complies with CEC/EPA no-load power consumption and average efficiency regulations ●● Built-in output constant-current control with primary-side feedback ●● Low start-up current (10 µA typical) ●● Built-in soft start ●● Built-in short circuit protection and output overvoltage protection ●● Optional AC line under/overvoltage protection ●● PFM operation at light load ●● Current sense resistor short protection ●● Overtemperature Protection 2.0 Description The iW1710 is a high performance AC/DC power supply controller which uses digital control technology to build peak current mode PWM flyback power supplies. The device operates in quasi-resonant mode at heavy load to provide high efficiency along with a number of key built-in protection features while minimizing the external component count, simplifying EMI design and lowering the total bill of material cost. The iW1710 removes the need for secondary feedback circuitry while achieving excellent line and load regulation. It also eliminates the need for loop compensation components while maintaining stability over all operating conditions. Pulse-by-pulse waveform analysis allows for a loop response that is much faster than traditional solutions, resulting in improved dynamic load response. The built-in current limit function enables optimized transformer design in universal off-line applications over a wide input voltage range. The ultra-low operating current at light load ensures that the iW1710 is ideal for applications targeting the newest regulatory standards for average efficiency and standby power. 3.0 Applications ●● Cable/DSL modems, WLAN access points and VOIP gateways. L N + + 1 NC 2 VSENSE VCC 8 OUTPUT 7 + VOUT RTN 3 VIN ISENSE 6 4 SD GND 5 Optional NTC Thermistor U1 iW1710 Figure 3.1 : Typical Application Circuit Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 1 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation  4.0 Pinout Description iW1710 1 NC 2 VSENSE 3 VIN 4 SD VCC 8 OUTPUT 7 ISENSE 6 GND 5 Pin # 1 Name NC Type - Pin Description No connection. 2 VSENSE Analog Input Auxiliary voltage sense (used for primary side regulation). 3 VIN Analog Input Rectified AC line average voltage sense. 4 SD Analog Input External shutdown control. Connect to ground through a resistor if not used. (see section 10.16) 5 GND Ground Ground. 6 ISENSE Analog Input Primary current sense (used for cycle-by-cycle peak current control and limit). 7 OUTPUT Output Gate drive for external MOSFET switch. 8 VCC Power Input Power supply for control logic and voltage sense for power-on reset circuitry. 5.0 Absolute Maximum Ratings Absolute maximum ratings are the parameter values or ranges which can cause permanent damage if exceeded. For maximum safe operating conditions, refer to Electrical Characteristics in Section 6.0. Parameter DC supply voltage range (pin 8, ICC = 20mA max) DC supply current at VCC pin Output (pin 7) Symbol VCC ICC Value -0.3 to 18 20 -0.3 to 18 Units V mA V VSENSE input (pin 2, IVsense ≤ 10 mA) VIN input (pin 3) ISENSE input (pin 6) SD input (pin 4) -0.7 to 4.0 V -0.3 to 18 V -0.3 to 4.0 V -0.3 to 18 V Power dissipation at TA ≤ 25°C Maximum junction temperature Storage temperature PD 526 mW TJ MAX 125 °C TSTG –65 to 150 °C Lead temperature during IR reflow for ≤ 15 seconds TLEAD 260 °C Thermal Resistance Junction-to-Ambient θJA 160 °C/W ESD rating per JEDEC JESD22-A114 2,000 V Latch-Up test per JEDEC 78 ±100 mA Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 2 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation  6.0 Electrical Characteristics VCC = 12 V, -40°C ≤ TA ≤ 85°C, unless otherwise specified (Note 1) Parameter Symbol Test Conditions VIN SECTION (Pin 3) Start-up low voltage threshold VINSTLOW TA= 25°C, positive edge Start-up current IINST VIN = 10 V, CVCC = 10 µF Shutdown low voltage threshold Input impedance VSENSE SECTION (Pin 2) Input leakage current Nominal voltage threshold Output OVP threshold - 01 (Note 2) Output OVP threshold - 21 (Note 2) VUVDC ZIN TA= 25°C, negative edge After start-up IBVS VSENSE(NOM) VSENSE(MAX) VSENSE(MAX) VSENSE = 2 V TA=25°C, negative edge TA=25°C, negative edge TA=25°C, negative edge, Load = 100 % OUTPUT SECTION (Pin 7) Output low level ON-resistance Output high level ON-resistance Rise time (Note 2) RDS(ON)LO RDS(ON)-HP tR ISINK = 5 mA ISOURCE = 5 mA TA = 25°C, CL = 330 pF 10% to 90% Fall time (Note 2) tF TA = 25°C, CL = 330 pF 90% to 10% Maximum switching frequency -01, -21 (Note 3) fSW(MAX) Any combination of line and load VCC SECTION (Pin 8) Maximum operating voltage (Note 2) Start-up threshold Undervoltage lockout threshold Operating current VCC(MAX) VCC(ST) VCC(UVL) ICCQ VCC rising VCC falling CL = 330 pF, VSENSE = 1.5 V Min 335 201 1.523 10.8 5.5 Typ 369 10 221 25 1.538 1.846 1.933 40 75 200 40 130 12 6.0 3.5 Max Unit 406 mV 15 µA 243 mV kW 1 μA 1.553 V V V W W 300 ns 60 ns 140 kHz 16 V 13.2 V 6.6 V 5 mA Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 3 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation  6.0 Electrical Characteristics (cont.) VCC = 12 V, -40°C ≤ TA ≤ 85°C, unless otherwise specified (Note 1) Parameter Symbol Test Conditions ISENSE SECTION (Pin 6) Peak limit threshold Isense short protection reference CC regulation threshold limit (Note 2) VPEAK VRSNS VREG-TH SD SECTION (Pin 4) Shutdown threshold Shutdown threshold in Startup (Note 2) Input leakage current Pull down resistance Pull up current source VSD-TH VSD-TH(ST) IBVSD RSD ISD TA = 25°C VSD = 1.0 V TA = 25°C TA = 25°C Min Typ Max Unit 1.045 1.1 1.155 V 0.127 0.15 0.173 V 1.0 V 0.95 1.0 1.05 V 1.2 V 1 µA 7916 8333 8750 W 96 107 118 µA Notes: Note 1. Note 2. Note 3. Adjust VCC above the start-up threshold before setting at 12 V. These parameters are not 100% tested, guaranteed by design and characterization. Operating frequency varies based on the line and load conditions, see Theory of Operation for more details. Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 4 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation VCC Start-up Threshold (V) VCC Supply Start-up Current (µA) % Deviation of Switching Frequency from Ideal  7.0 Typical Performance Characteristics 9.0 6.0 3.0 0.00.0 2.0 4.0 6.0 8.0 10.0 12.0 14.0 VCC (V) Figure 7.1 : VCC vs. VCC Supply Start-up Current 0.3 % -0.3 % -0.9 % -1.5 % -50 -25 0 25 50 75 100 125 Ambient Temperature (°C) Figure 7.3 : % Deviation of Switching Frequency to Ideal Switching Frequency vs. Temperature Internal Reference Voltage (V) 12.2 12.0 11.8 11.6 -50 -25 0 25 50 75 100 125 Ambient Temperature (°C) Figure 7.2 : Start-Up Threshold vs. Temperature 1.548 1.538 1.528 1.518 -50 -25 0 25 50 75 100 125 Ambient Temperature (°C) Figure 7.4 : Internal Reference vs. Temperature Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 5 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation  8.0 Functional Block Diagram VIN 3 ENABLE ZVin 25 kΩ ADC VIN_A 0.2 V ~ 2.0 V ENABLE Start-up 8 VCC VSENSE 2 ISD Signal Conditioning VVMS VFB Detection Switch Digital Logic Control Gate Driver 60 kΩ 7 OUTPUT SD 4 RSD + VSD-TH – GND 5 DAC VIPK VOCP IPEAK 0V~1V –– + – + 1.1 V 6 ISENSE Figure 8.1 : iW1710 Functional Block Diagram 9.0 Theory of Operation The iW1710 is a digital controller which uses a proprietary primary-side control technology to eliminate the optoisolated feedback and secondary regulation circuits required in traditional designs. This results in a low-cost solution for AC/DC adapters. The iW1710 uses Critical Discontinuous Conduction Mode (CDCM) or Pulse Width Modulation (PWM) mode at high output power levels and switches to Pulse Frequency Modulation (PFM) mode at light load to minimize power dissipation to meet EPA 2.0 specification. Furthermore, iWatt’s digital control technology enables fast dynamic response, tight output regulation, and full featured circuit protection with primary-side control. Referring to the block diagram in Figure 8.1, the digital logic control block generates the switching on-time and off-time information based on the line voltage and the output voltage control algorithm to reduce system design time and improve reliability. Furthermore, accurate secondary constant-current operation is achieved without the need for any secondary-side sense and control circuits. The built-in protection features include overvoltage protection (OVP), output short circuit protection (SCP) and soft-start, AC line brown out, overcurrent protection, and Isense fault protection. Also the iW1710 automatically shuts down if it detects any of its sense pins to be either open or short. iWatt’s digital control scheme is specifically designed to address the challenges and trade-offs of power conversion design. This innovative technology is ideal for balancing new feedback signal and provides commands to dynamically regulatory requirements for green mode operation with more control the external MOSFET current. The system loop is practical design considerations such as lowest possible cost, compensated internally by a digital error amplifier. Adequate smallest size and highest performance output control. system phase and gain margin are guaranteed by design and no external analog components are required for loop compensation. The iW1710 uses an advanced digital Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 6 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation  9.1 Pin Detail Pin 2 – VSENSE Sense signal input from auxiliary winding. This provides the secondary voltage feedback used for output regulation. Pin 3 – Vin Sense signal input from the rectified line voltage. VIN is used for line regulation. The input line voltage is scaled down using a resistor network. It is used for input undervoltage and overvoltage protection. This pin also provides the supply current to the IC during start-up. Pin 4 – SD If at any time the VCC voltage drops below VCC(UVL) threshold then all the digital logic is reset. At this time VIN switch turns off so that the VCC capacitor can be charged up again towards the start-up threshold. Start-up Sequencing VIN VCC(ST) VCC External shutdown control. If the shutdown control is not used, this pin should be connected to GND via a resistor. (see Section 10.16). ENABLE Pin 5 – GND Figure 9.1 : Start-up Sequencing Diagram Ground. 9.3 Understanding Primary Feedback Pin 6 – ISENSE Primary current sense. Used for cycle by cycle peak current control. Pin 7 – OUTPUT Gate drive for the external MOSFET switch. Pin 8 – VCC Power supply for the controller during normal operation. The controller will start up when VCC reaches 12 V (typical) and will shut-down when the VCC voltage is below 6 V (typical). A decoupling capacitor should be connected between the VCC pin and GND. Figure 9.2 illustrates a simplified flyback converter. When the switch Q1 conducts during tON(t), the current ig(t) is directly drawn from rectified sinusoid vg(t). The energy Eg(t) is stored in the magnetizing inductance LM. The rectifying diode D1 is reverse biased and the load current IO is supplied by the secondary capacitor CO. When Q1 turns off, D1 conducts and the stored energy Eg(t) is delivered to the output. iin(t) + ig(t) N:1 id(t) VO vin(t) vg(t) – D1 + CO IO VAUX VAUX TS(t) Q1 9.2 Start-up Prior to start-up the VIN pin charges up the VCC capacitor through the diode between VIN and VCC (see Figure 8.1). When VCC is fully charged to a voltage higher than the startup threshold VCC(ST), the ENABLE signal becomes active and enables the control logic; the VIN switch turns on, and the analog-to-digital converter begins to sense the input voltage. Once the voltage on the VIN pin is above VINSTLOW, the iW1710 commences soft start function. An adaptive soft-start control Figure 9.2 : Simplified Flyback Converter In order to tightly regulate the output voltage, the information about the output voltage and load current needs to be accurately sensed. In the DCM flyback converter, this information can be read via the auxiliary winding. During the Q1 on-time, the load current is supplied from the output filter capacitor CO. The voltage across LM is vg(t), assuming the voltage dropped across Q1 is zero. The current in Q1 ramps up linearly at a rate of: algorithm is applied at startup state, during which the initial output pulses will be small and gradually get larger until the dig (t) = vg (t) (9.1) full pulse width is achieved. The peak current is limited cycle dt LM by cycle by Ipeak comparator. Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 7 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation  At the end of on-time, the current has ramped up to: ig _ peak (tON ) = vg (t) × tON LM (9.2) This current represents a stored energy of: Eg = LM 2 × ig _ peak (tON )2 (9.3) When Q1 turns off, ig(t) in LM forces a reversal of polarities on all windings. Ignoring the communication-time caused by the leakage inductance LK at the instant of turn-off, the primary current transfers to the secondary at a peak amplitude of: id (t) = NP NS × ig _ peak (tON ) (9.4) Assuming the secondary winding is master and the auxiliary winding is slave. The real-time waveform analyzer in the iW1710 reads the auxiliary waveform information cycle by cycle. The part then generates a feedback voltage VFB. The VFB signal precisely represents the output voltage and is used to regulate the output voltage. 9.4 Constant Voltage Operation After soft-start has been completed, the digital control block measures the output conditions. It determines output power levels and adjusts the control system according to a light load or a heavy load. If this is in the normal range, the device operates in the Constant Voltage (CV) mode, and changes the pulse width (TON), and off time (TOFF) in order to meet the output voltage regulation requirements. During this mode the PWM switching frequency is between 30 kHz and 130 kHz, depending on the line and load conditions. If less than 0.2 V is detected on VSENSE it is assumed that the auxiliary winding of the transformer is either open or shorted and the iW1710 shuts down. VAUX = VO x NAUX NS 9.5 Dynamic Load Transient There are three components that compose the voltage drop during a load transient event. VAUX 0V VDROP(cable) is the drop in voltage due to the increased current going through the connector and/or the cable. VDROP(cable) = RCABLE × ∆IOUT (9.6) VAUX = -VIN x NAUX NP Figure 9.3 : Auxiliary Voltage Waveforms The auxiliary voltage is given by: VAUX = N AUX NS (VO + ∆V ) (9.5) and reflects the output voltage as shown in Figure 9.3. The voltage at the load differs from the secondary voltage by a diode drop and IR losses. The diode drop is a function of current, as are IR losses. Thus, if the secondary voltage is always read at a constant secondary current, the difference The second component which affects the voltage drop during load transient is V . DROP(sense) This voltage drop is the drop in voltage before the VSENSE signal is able to show a significant drop in output voltage. This is determined by Vmin or the reference voltage at which a load transient is detected. The smaller the Vmin is the smaller this drop in voltage is. ( ) VDROP(sense) = VSENSE(nom) − VSENSE(min) × VOUT (PCB) VSENSE ( nom) (9.7) Keep in mind that a smaller Vmin is less tolerant of noise and distortions in VSENSE than a larger one. The final drop in voltage is due to the time from when VSENSE drops Vmin to when the next VSENSE signal appears. In the worst case condition this is how much voltage drops during the longest switching period. between the output voltage and the secondary voltage will be a fixed ΔV. Furthermore, if the voltage can be read when the secondary current is small; for example, at the knee of VDROP ( IC ) = IOUT × TP(No load) COUT (9.8) the auxiliary waveform (see Figure 9.3), then ΔV will also be small. With the iW1710, ΔV can be ignored. A larger output capacitance in this case greatly reduces the V . DROP(IC) Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 8 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation Output Voltage CC mode  When the iW1710 detects the output voltage to be significantly higher than the nominal output voltage, the device begins to increase the switching period which lowers the output voltage. TPERIOD(CLAMP) refers to the maximum switching period to which the iW1710 increases to when an output voltage is detected to be significantly over the nominal. During fast load changes the output voltage may not have time to settle before the load is changed. Thus for this case when the power supply goes from light load to heavy load prior to output voltage settling TPERIOD(CLAMP) substitutes TPERIOD(PFM) in equation 9.8. To achieve this regulation the iW1710 senses the load current indirectly through the primary current. The primary current is detected by the ISENSE pin through a resistor from the MOSFET source to ground. VNOM CV mode 9.6 Valley Mode Switching In order to reduce switching losses in the MOSFET and EMI, the iW1710 employs valley mode switching when IOUT is above 50%. In valley mode switching, the MOSFET switch is turned on at the point where the resonant voltage across the drain and source of the MOSFET is at its lowest point (see Figure 9.4). By switching at the lowest VDS, the switching loss will be minimized. Output Current Figure 9.5 : Power Envelope IOUT(CC) Gate VDS Figure 9.4 : Valley Mode Switching Turning on at the lowest VDS generates lowest dV/dt, thus valley mode switching can also reduce EMI. To limit the switching frequency range, the iW1710 can skips valleys (seen in the first cycle in Figure 9.4) when the switching frequency becomes too high. iW1710 provides valley mode switching during constant output current operation. So, the EMI and switching losses are still minimized during CC mode. This feature is superior to other quasi-resonant technologies which only support valley mode switching during constant voltage operation. This is beneficial to applications, such as chargers, where the power supply mainly operates in CC mode. 9.7 Constant Current Operation 9.8 PFM Mode at Light Load The iW1710 normally operates in a fixed frequency PWM or critical discontinuous conduction mode when IOUT is greater than approximately 10% of the specified maximum load current. As the output load IOUT is reduced, the on-time tON is decreased. At the moment that the load current drops below 10% of nominal, the controller transitions to Pulse Frequency Modulation (PFM) mode. Thereafter, the ontime will be modulated by the line voltage and the off-time is modulated by the load current. The device automatically returns to PWM mode when the load current increases. 9.9 Variable Frequency Operation At each of the switching cycles, the falling edge of VSENSE will be checked. If the falling edge of VSENSE is not detected, the off-time will be extended until the falling edge of VSENSE is detected. The maximum allowed transformer reset time is 120 µs. When the transformer reset time reaches this maximum reset time, the iW1710 immediately shuts off. 9.10 Internal Loop Compensation The iW1710 incorporates an internal Digital Error Amplifier with no requirement for external loop compensation. For a The constant current mode (CC mode) is useful in battery charging applications. During this mode of operation the iW1710 will regulate the output current at a constant level typical power supply design, the loop stability is guaranteed to provide at least 45 degrees of phase margin and –20dB of gain margin. regardless of the output voltage, while avoiding continuous conduction mode. Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 9 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation  9.11 Voltage Protection Functions The iW1710 includes functions that protect against input line undervoltage (UV) and the output overvoltage (OVP). Vgate OVP Detection OTP Detection The input voltage is monitored by the VIN pin and the output voltage is monitored by the VSENSE pin. If the voltage at these pins exceed their respective undervoltage or overvoltage thresholds the iW1710 shuts down immediately. However, the IC remains biased which discharges the VCC supply. Once VCC drops below the UVLO threshold, the controller resets itself and then initiates a new soft-start cycle. The controller continues attempting start-up until the fault condition is removed. 9.12 PCL, OC and SRS Protection Peak-current limit (PCL), over-current protection (OCP) and sense-resistor short protection (SRSP) are features built-into the iW1710. With the ISENSE pin the iW1710 is able to monitor the primary peak current. This allows for cycle by cycle peak current control and limit. When the primary peak current multiplied by the ISENSE sense resistor is greater than 1.1 V over current is detected and the IC will immediately turn off the gate drive until the next cycle. The output driver will send out switching pulse in the next cycle, and the switching pulse will continue if the OCP threshold is not reached; or, the switching pulse will turn off again if the OCP threshold is still reached. Detection Switch Detection Switch: When switch is low SD pin is connected to RSD When switch is high SD pin is connected to a current source ISD Figure 9.6 : SD Pin Detection Cycles During an overvoltage monitor cycle the SD pin is connected to a resistance internal to the chip, RSD, to ground and the voltage on the SD pin is observed. SD pin iW1710 ISD VSD-TH OVP / OTP Fault Detect If the ISENSE sense resistor is shorted there is a potential danger of the over current condition not being detected. Thus the IC is designed to detect this sense-resistor-short fault after the start up, and shutdown immediately. The VCC will be discharged since the IC remains biased. Once VCC drops below the UVLO threshold, the controller resets itself and then initiates a new soft-start cycle. The controller continues attempting start-up, but does not fully start-up until the fault condition is removed. Figure 9.7 : Internal Function of SD Pin 9.13 Shutdown The shutdown (SD) pin in the iW1710 provides protection against overtemperature (OTP) and additional overvoltage (OVP) for the power supply. The iW1710 switches between monitoring overtemperature fault and overvoltage fault. In order to detect the resistance in the NTC for an overtemperature fault, the iW1710 connects a current source to the SD pin and checks the voltage on the pin. To ensure that the current source is settled before the voltage is checked both OTP and OVP are detected on the last cycle, as depicted in figure 9.6. Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 10 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation  10.0 Design Example 10.1 Design Procedure This design example gives the procedure for a flyback converter using iW1710. Refer to Figure 12.1 for the application circuit. The design objectives for this adapter are given in table 10.1. It meets UL, IEC, and CEC requirements. Determine the Design Specifications (Vout, Iout_max, Vin_max, Vin_min, ƒline, Ripple specification) Determine Part Number Determine Rvin Resistors Determine Turns Ratio Determine Operating VinTon Limit Determine Magnetizing Inductance Determine Primary Turns Determine Secondary Turns Determine Bias Turns and Vcc Capacitance Determine Vsense Resistors Parameter Input Voltage Frequency No Load Input Output Voltage Output Current Output Ripple Power Out EPA 2.0 Efficiency Symbol VIN fIN PIN VOUT(Cable) IOUT VRIPPLE POUT h Range 85 - 264 VRMS 47 - 64 Hz 100 mW 12.0 V 1.2 A < 100 mV 15 W 80% Table 10.1 : iW1710 Design Specification Table 10.2 Determine Part Number Based on design specifications, choose the most suitable part for the design. For more information on the options see section 14.0. Use equation 10.1 for VOUT in the following calculations, where VFD is the forward voltage of the output diode. VOUT = VOUT (Cable) + VCableDrop + VFD (10.1) For this example there is no cable so VCableDrop is 0 V , assuming VFD is 0.5, VOUT is: VOUT = 12.0V + 0V + 0.5V = 12.5V No Can you wind this transformer ? Yes Determine Current Sensing Resistor Determine Input Bulk Capacitance Determine Output Capacitance Determine Snubber Network Determine Ton Delay Compensation 10.3 Input Selection VIN resistors are chosen primarily to scale down the input voltage for the IC. The default scale factor for the input voltage in the IC is 0.0043 and the internal impedance of this pin is ZIN (25 kW). Therefore, the VIN resistors should equate to: RVin = Z IN 0.0043 − Z IN (10.2) From equation 10.2, ideally RVin should be 5.79 MW. A lower value of RVin can decrease the startup time of the power supply. The value of RVin affects the (VINTON) limits of the IC. Determine SD Pin Components (VIN ⋅ TON )limit = 0.0043× 720V ( Z IN RVin ⋅ ms + ZIN ) (10.3) Finish Figure 10.1 : iW1710 Design Flow Chart (VIN ) ⋅TON PFM = 0.0043× 135V ( Z IN RVin ⋅ ms + ZIN ) (10.4) Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 11 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation  For this example RVin is chosen to be 5.1 MΩ therefore, ( ) VIN ⋅TON limit = 0.0043× 25kW 720V ⋅ ms = 635V ⋅ ms (5.1M W + 25kW) (VIN ) ⋅TON PFM = 0.0043× 25kW 135V ⋅ ms = 119V ⋅ ms (5.1M W + 25kW) approximately 2 μs as a starting point and then adjusted after the power supply is tested. Gate Keep in mind, by changing RVin to be something other than 5.79 MW the minimum and maximum input voltage for startup also changes. Since the iW1710 uses the exact scaled value of VIN for its calculations, there should be a filter capacitor on the input pin to filter out any noise that may appear on the VIN signal. This is especially important for line in surge conditions. 10.4 Turns Ratio The maximum allowable turns ratio between the primary and secondary winding is determined by the minimum detectable reset time of the transformer during PFM mode. ( ) NTR(max) = VIN ⋅TON PFM TRESET (min) ×VOUT (10.5) Setting TRESET(min) at 1.5 μs, NTR(max) = 119V ⋅ms 1.5ms ×12.5V = 6.3 For this example a turns ratio of 6 is chosen. Keep in mind in valley mode switching the higher the turns ratio the lower the VDS turn-on voltage, which means less switch turn-on power loss. Also consider the voltage stress on the MOSFET (VDS) is higher with an increase in turns ratio. The voltage stress on the output diode is lower with an increase in turns ratio respectively. 10.5 Operating Maximum (VINTON) VDS TON TRESET TPERIOD TRES Figure 10.2 : VDS Timing When both criterion are met then (VINTON)MAX can be determined by equation 10.8. ( ) VIN ⋅TON max =    f SW (max op) ×   1 VINDC (min) + 1 NTR ×VOUT   −1  where, fSW (max op) = 1 TP(QRmin) (10.8) Where VINDC(min) is the minimum input voltage across the bulk capacitor. In order to avoid input undervoltage detection during normal operation, VINDC(min) should be set above the input undervoltage shutdown limit. VINDC (min) > RVin + ZIN Z IN ⋅ VUVDC Assuming TRES is 2 μs then: TP(QR min) > 10ms TP' (QR min) > 1 110kHz + 2ms = 11.1ms (10.9) VINDC (min) > 5.1M W + 25kW 25kW × 0.369V = 76V Maximum operating VINTON or (VINTON)MAX for valley mode switching is traditionally designed at full load and lowest input voltage. For the iW1710, two constraints (equation 10.6 and 10.7) need to be satisfied so that indeed (VINTON)MAX occurs at full load and lowest input voltage. To give some margin, we use 79 V for VINDC(min) in equation 10.8, Choosing, ƒSW(max op) = 72kHz and TP(QR min) = 14ms ( ) ( ) VIN ⋅TON max = 72kHz × 1 79V + 1 6×12.5V  −1  = 534V ⋅ ms TP(QR min) > 1 100kHz TP' (QR min) > 1 110kHz + TRES (10.6) (10.7) Also, to provide enough margin for component values, usually: ( ) ( ) VIN ⋅TON < max VIN ⋅TON limit × 0.85 (10.10) TRES is the VDS resonant period as shown in Figure 10.2. TRES can be estimated to be (VIN ⋅TON )max < 635V ⋅ ms × 0.85 = 540V ⋅ ms Rev. 1.9 iW1710 Page 12 5/6/10 iWatt Confidential-Protected By Traceable Watermark iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation ( )  Sincewecalculated534V·μsasourVIN·TONwehaveenough margin. 10.6 Magnetizing Inductance A feature of the iW1710 is the lack of dependence on the magnetizing inductance for the CC curve. Although the constant current limit does not depend on the magnetizing inductance, there are still restrictions on the magnetizing inductance. The maximum LM is limited by the amount of power that needs to come out of the transformer in order for the power supply to regulate. This is given by: ( ) LM (max) = VIN ⋅TON 2 max × fsw(max op) 2 × PXFMR(max) PXFMR(max) = VOUT × IOUT hX (10.11) Where ηX is the efficiency of the transformer, for this example we assume it’s 87 %. PXFMR(max) = 12.5V ×1.2A 0.87 = 17.2W (534V ⋅ms)2 × 72kHz LM (max) = 2 ×17.2W = 0.597mH The minimum LM is limited by the maximum allowable peak primary current. VREG-TH corresponds to the maximum ISENSE voltage. See section 10.11 to calculate RIsense. Therefore LM is limited by: LM (min) = 2 × PXFMR(max) fSW (max op) × VREG−TH RIsense 2 (10.12) LM (min) = 2 ×12.5V ×1.2A 72kHz × 1.0V 1.08W 2 = 0.486mH For this example, we choose LM to be 0.577 mH. If these limits do not give enough tolerance for LM, increasing (VINTON)max can raise the maximum limit on LM. Take care not to go above (VINTON)limit. Also, keep in mind that if equation 10.6 and 10.7 are not met then (VINTON)max does not occur at full load and lowest input voltage, thus some of the equations here would be invalid. ( ) NPRI ≥ VIN ⋅TON max Bmax × Ae (10.13) Where BMAX is maximum allowed flux density and Ae is the core area. From the transformer core datasheet we find that for this example BMAX is 300 mT. For an EE19 core, Ae is 22.6 mm2. N PRI ≥ 534V ⋅ms 320mT × 20.1mm2 = 83.0T For this example, we choose 90 primary turns. 10.8 Secondary Winding From the primary winding turns, we obtain the secondary winding. N SEC = N PRI NTR (10.14) Thus, in our example: N SEC = 90T 6 = 15T 10.9 Bias Winding and VCC Capacitance VCC is the supply to the iW1710 and should be below 16 V. The bias winding needs to ensure than VCC does not exceed 16 V during normal operation. ( ) NBIAS = N SEC VCC + VFD VOUT (10.15) Set VCC at around 10 V N BIAS = 15T ×10.5V 12.5V = 12T Choose a value for NBIAS to be close to this number, for this example we choose 12 turns. The VCC capacitor (CVcc) stores the VCC charge during IC operation and the controller checks this voltage and makes sure it is within range before starting and operating. The startup time is a function of how quickly this capacitor can charge up. 10.7 Primary Winding In order to keep the transformer from saturation, the tSTART −UP = CVCC ×VCC(ST ) − I VINAC × 2 RVin INST (10.16) maximum flux density must not be exceeded. Therefore the minimum primary winding must meet: 10.10 VSENSE Resistors and Winding The output voltage regulation is mainly determined by the feedback signal VSENSE. Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 13 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation VSENSE = VOUT _ PCB × KSENSE (10.17) Where: ( ) KSENSE = RBVsns RBVsns + RTVsns × NVsense N SEC (10.18) Internally, VSENSE is compared to a reference voltage V . SENSE(nom) Where, VSENSE(nom) is 1.538 V. ( )  KSENSE = VSENSE ( nom) VOUT _ PCB (10.19) K SENSE = 1.538V 12.0V = 0.128 From here we can find the ratio necessary for RBVsns and RTVsns. For this example we set RTVsns to be 24 kΩ. Assuming  we use the same winding for both VSENSE and VCC: 0.128 = RBVSNS × 12T RBVSNS + 24kW 15T → RBVSNS = 4.57kW RISNS = 6× 0.5V × 0.87 2 ×1.2 A = 1.08W We recommend using ±1% tolerance resistors for RIsense. 10.12 Input Bulk Capacitor The input bulk capacitor, CBULK is chosen to maintain enough input power to sustain constant output power even as the input voltage is dropping. In order for this to be true CBULK must be: CBULK = 2× PIN × 0.25 + 1 2π × arcsin   VINDC (min) 2 ×VINAC (min) 2 ×VI2NAC(min) − VI2NDC(min) × fline   PIN = VOUT (Cable) × IOUT hpower supply (10.24) VINAC(min) is the minimum input voltage (rms) to be inputted into the power supply and fline is the lowest line frequency for the power supply (in this case 47 Hz). VINDC(min) is calculated from equation 10.9.  At this point the transformer design is complete. This would be a good time to confirm that this transformer is feasible to ( ) build. ( ) 10.11 Current Sense Resistor  The ISENSE resistor determines the maximum current output of the power supply. The output current of the power supply is determined by: IOUT = 1 2 × NTR × I PRI ( pk ) × TRESET TPERIOD × hX (10.20) When the maximum current output is achieved the voltage seen on the ISENSE pin (VIsense) should reach its maximum. Thus, at constant current limit:  IPRI(pk) = VIsense(CC ) RIsense (10.21) Substituting this into equation 10.20 we get:  VIsense(CC) = TPERIOD TRESET × KC (10.22) PIN = 12.5V ×1.2A 0.72 = 20.8W CBULK = 2× 20.8W × 0.25 + 1 2π × arcsin 79V 2 ×85Vac 2× (85Vac )2 − (79V )2 × 47Hz   = 39mF 10.13 Output Capacitance The output capacitance affects both the steady state ripple and the dynamic response of the power supply. Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then: COUT (Steady State) = QOUT VOUT (ripple) (10.25) The output capacitor supplies the load current when the secondary current is below the output current. ( )2 QOUT = LM × ISEC( pk ) − IOUT 2 × NT2R × hX ×VOUT (10.26) For iW1710 KC is 0.5 V, therefore RIsense depends on the maximum output current by; RIsense = NTR × KC 2 × IOUT × hX (10.23) The ISEC(pk) is: ( ) ISEC( pk) = VIN ⋅TON LM MAX × NTR × hX (10.27) From table 10.1 IOUT is given to be 1.2 A, therefore RIsense is: So to keep VOUT(ripple) to be 100 mV, Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 14 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation  ISEC(pk) = 534V ⋅ ms 0.541mH × 6× 0.87 = 5.152 A COUT (Dynamic) = 0.5 A × 1.0V − 0V 461ms − 0.543V = 504mF 0.541mH (5.152A −1.2A)2 QOUT = 2 × 62 × 0.87 ×12.5V = 10.8mC COUT (Steady State) = 10.8mC 100mV = 208mF In this calculation ESR and ESL are ignored; the reason this calculation is still valid is because of the second stage LC filter on the output of the supply. These two components reduce the ESR and ESL ripple; however keep in mind that the ripple is a little higher in reality than this calculation would suggest. Assume that the load transient goes from no load to I . OUT(HIGH) Then from section 9.5, equation 9.8 we find that the relationship between output capacitance (C ) OUT(Dynamic) and VDROP(IC) is : COUT (Dynamic) = IOUT (HIGH ) × TP(No load) VDROP ( IC ) (10.28) Then solving for VDROP(IC) from Figure 9.4, where VDynamic(DROP) is the maximum allowable drop in voltage for the design during dynamic response, VDROP(Cable) is the drop in voltage due to the cable resistance, and VDROP(sense) is the drop in voltage before VSENSE signal is low enough to register a dynamic transient. COUT (Dynamic) = IOUT (HIGH ) × TP(No load) VDynamic(Drop) − VDROP(Cable) − VDROP(sense) (10.29) Where TP(No load) is the maximum period under no load condition, given by equation 10.30: ( ) TP(No load) = RPreload × VIN ⋅TON 2 × LM ×VO2UT 2 PFM × hNo load (10.30) Assume that we want no more than 1.0 V drop on VOUT(PCB) during load transient from no load to 50% load and the efficiency of the power supply at no load (ηNo ) load is 50% , then COUT(Dynamic) is: 5.6kW × (119V ⋅ms)2 TP(PFM ) = 2 × 0.541mH × (12.5V )2 = 461ms Pick the larger capacitance value between COUT(Dynamic) and C . OUT(Steady State) In this case COUT is chosen to be 680 μF. 10.14 Snubber Network The snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive. The goal is to dissipate the energy from the leakage inductance of the transformer. For simplicity and a more conservative design first assume the energy of the leakage inductance is only dissipated through the snubber. Thus: 1 2 × Llk × I 2 PRI ( pk ) = 1 2 × CSNUB × VS2nub( pk ) − VS2nub(val )   (10.31) Llk can be measured from the transformer and VDS is the voltage across the MOSFET. VSnub(pk) and VSnub(val) refer to the voltage measured across the snubber capacitor. Choose a CSNUB, keeping in mind that the larger the value of CSNUB the lower the voltage stress is on the MOSFET. However, capacitors are more expensive the larger their capacitance. Choose CSNUB based on these two criteria and select VSnub(pk) and V . Snub(val) Now a resistor needs to be selected to dissipate VSnub(pk) to VSnub(val) during the on-time of the gate driver. The dissipation of this resistor is given by: V = e Snub(val) −TP ( min op) RSNUB ⋅CSNUB VSnub( pk ) (10.32) Using equation 10.32 solve for RSNUB. This gives a conservative estimate of what CSNUB and RSNUB should be. Included in the snubber network is also a resistor in series with a diode. The diode directs current to the snubber capacitor when the MOSFET is turned off; however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on. This reverse current occurs because there is a short period of time when the diode still conducts after switching from forward biased to reverse biased. This conduction distorts the falling edge of the VSENSE signal and affects the operation of the IC. So, the resistor in series with the diode is there to diminish the reverse current that goes through the diode immediately after the MOSFET is turned on. Since there is no cable, VDROP(cable) is 0 V. VDROP ( sense ) = (1.538V −1.48V )× 12.0V 1.538V = 0.543V 10.15 TON Delay Filter iW1710 also contains a feature that allows for adjustment to match high line and low line constant current curves. The mismatch in high line and low line is due to the delay Plug everything into equation 10.19: from the IC propagation delay, driver turn-on delay, and the MOSFET turn-on delay. The driver turn-on delay maybe Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 15 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation further increased by a gate resistor to the MOSFET. To ( ) adjust for these delays the iW1710 factors these delays into its calculations and slightly over compensates for them to provide flexibility. RDly and CDly provide extra delay in the circuit to tweak the compensation. To determine values RDly and CDly follow these steps: 1.  Measure the difference between high line and low line constant current limit without filter components. 2.  Find the curve that best matches this difference from Figure 11.1.  3.  Find the LM that matches the power supply, and find the tRC. 4.  Find RDly and CDly from equation 10.33 tRC = RDly × CDly (10.33) RNTC +RSD(ext) × ISD > VSD−TH (10.34) in order not to trigger OTP fault during normal operation. OVP Only For the other four cycles, the iW1710 connects the SD pin to RSD to ground (see section 9.13). At the last cycle the iW1710 observes the voltage on the SD pin and detects an OVP fault if the voltage is higher than VSD-TH, 1 V. In order to not trigger OVP fault, assuming 0 V drop across the series diode, RSD(ext) must meet: VOUT _ PCB N SEC × N AUX × RSD RSD + RSD(ext) < VSD−TH (10.35) where, RSD = 8.333 kΩ  10.16 SD Protection The SD pin can be configured to provide three different types of protection: OTP protection, OVP protection and both OVP and OTP Protection. Figure 10.3 shows the three  configurations plus the configuration for no OTP and OVP protection. Both OTP and OVP To find RSD1(ext) so that OVP can be detected, use equation 10.35. To find RSD2(ext) in series with the NTC use equation 10.34. No OTP and OVP SD pin  RNTC RSD(ext) (optional) RSD1(ext) SD pin RNTC RSD2(ext) OUTPUT a) Overtemperature Protection only b) Overtemperature Protection and Overvoltage Protection SD pin RSD(ext)  SD pin RSD(ext) c) Overvoltage Protection only d) No Overtemperature Protection and no overvoltage protection Figure 10.3 : SD Pin Application Configurations  OTP Only If OTP and OVP from the SD pin are not needed, simply place a resistor, RSD(ext) to ground from the SD pin. Make sure RSD(ext) meets equation 10.36 so OTP protection does not trip. RSD(ext) × ISD > VSD−TH (10.36) Note that this means OVP is not detected through the SD pin; however, OVP from VSENSE pin is still active and the iW1710still shuts down if overvoltage condition is detected. Since for this example OTP and OVP are not necessary we place a resistor from SD pin to ground and calculate its value from equation 10.36. RSD(ext) > 1.2V 100mA = 12kW 10.17 PCB Layout In the iW1710, there are two signals that are important to To detect an overtemperature protection the iW1710 sends a 107 mA current (ISD) to the SD pin every four cycles (see section 9.13). On the last cycle the iW1710 observes the control the output performance; these are the ISENSE signal and the VSENSE signal. The ISENSE resistor should be close to the source of the MOSFET to avoid any trace resistance from voltage on the SD pin and detects an OTP fault if the voltage is lower than VSD-TH, 1.0 V during normal operation and 1.2 V during startup. So RSD(ext) in series with NTC must meet contaminating the ISENSE signal. Also, the ISENSE signal should be placed close to the ISENSE pin. The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal. Also for better output performance all Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 16 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation  bypass capacitors should be placed close to their respective pins. To reduce EMI, switching loops need to be minimized. These loops include: 1.  The input bulk capacitor, primary winding, MOSFET and RIsense loop. 2.  The output diode, output capacitor and secondary winding loop. 3.  VCC winding and rectifier diode loop. L N + 2) + VOUT RTN 1) + 3) Optional NTC Thermistor 1 NC VCC 8 2 VSENSE OUTPUT 7 3 VIN ISENSE 6 4 SD GND 5 U1 iW1710 Figure 10.4 : Switching Loops To improve ESD performance provide a low impedance path from the ground pin of the transformer to the ac power source and make sure this path does not go through the IC ground pin. A discharge spark gap helps to transfer ESD and EOS energy from the secondary side of the power supply directly to the external ac power source. In a switch-mode power supply there are several ground signals, namely: the power ground, the switching ground and the control logic ground. These ground signals should be connected by a star connection. Ground traces should be kept as short as possible. A thick trace on the switching ground helps to lessen switching losses. Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 17 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation (RDly x CDly), τRC (ns)  11.0 Design Example Performance Characteristics 120 100 ∆IOUT (Note1) 50 mA 40 mA 30 mA 80 20 mA 10 mA 60 40 20 00.0 0.5 1.0 1.5 2.0 2.5 3.0 Magnetizing Inductance LM (mH) Figure 11.1 : TON Compensation Chart 12.0 Application Circuit 1 A / 250 V L N 470 µH Cbulk 10 µF 10 kΩ + Cbulk + 33 µF Rvin 2.7 MΩ Rsnub 75 kΩ Csnub 1 nF Rvin 2.4 MΩ 68 Ω 470 pF + Cvcc 4.7 Ω 4.7 µF Rtvsns 24.0 kΩ Rbvsns 4.57 kΩ 22 pF 1 NC VCC 8 2 VSENSE OUTPUT 7 3 VIN ISENSE 6 4 SD GND 5 Rntc 20 kΩ U1 iW1710 Rgate 22 Ω Rdly (opt) 1000 Ω Cdly (opt) 33 pF Risense 1.08 Ω Cout + 680 µF VOUT Rpreload 5.6 kΩ RTN (opt): optional Figure 12.1 : Typical Application Circuit Note 1: ΔIOUT refers to the difference in constant current limit between 264 Vac and 90 Vac when no RDLY and CDLY are applied. Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 18 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation 13.0 Physical Dimensions 8-Lead Small Outline (SOIC) Package D 8 E 1 5 4 H  e A1  B COPLANARITY 0.10 (0.004) A2 A SEATING PLANE C h x 45° α L Symbol Inches MIN MAX A 0.053 0.069 A1 0.0040 0.010 A2 0.049 0.059 B 0.014 0.019 C 0.007 0.010 D 0.189 0.197 E 0.150 0.157 e 0.050 BSC H 0.228 0.244 h 0.10 0.020 L 0.016 0.049 α 0° 8° Compliant to JEDEC Standard MS12F Figure 13.1 : Physical dimensions, 8-lead SOIC package  Controlling dimensions are in inches; millimeter dimensions are for reference only This product is RoHS compliant and Halide free. Soldering Temperature Resistance: [a] Package is IPC/JEDEC Std 020D Moisture Sensitivity Level 1 [b] Package exceeds JEDEC Std No. 22-A111 for Solder Immersion Resistance; package can withstand 10 s immersion < 270˚C  Dimension D does not include mold flash, protrusions or gate burrs. Mold flash, protrusions or gate burrs shall not exceed 0.15 mm per end. Dimension E1 does not include interlead flash or protrusion. Interlead flash or protrusion shall not exceed 0.25 mm per side. D and E1 dimensions are determined at datum H. The package top may be smaller than the package bottom. Dimensions D and E1 are determined at the outermost extremes of the plastic bocy exclusive of mold flash, tie bar burrs, gate burrs and interlead flash, but including any mismatch between the top and bottom of the plastic body. Millimeters MIN MAX 1.35 1.75 0.10 0.25 1.25 1.50 0.35 0.49 0.19 0.25 4.80 5.00 3.80 4.00 1.27 BSC 5.80 6.20 0.25 0.50 0.4 1.25 14.0 Ordering Information  Part Number iW1710-01 iW1710-21 Options Cable Comp = 0 mV Cable Comp = 300 mV  Note 1: Tape & Reel packing quantity is 2,500/reel. Package SOIC-8 SOIC-8 Description Tape & Reel1 Tape & Reel1 Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 19 iW1710 Digital PWM Current-Mode Controller for Quasi-Resonant Operation  About iWatt iWatt Inc. is a fabless semiconductor company that develops intelligent power management ICs for computer, communication, and consumer markets. The company’s patented pulseTrain™ technology, the industry’s first truly digital approach to power system regulation, is revolutionizing power supply design. Trademark Information © 2009 iWatt, Inc. All rights reserved. iWatt, the iW light bulb, EZ-EMI and pulseTrain are trademarks of iWatt, Inc. All other trademarks and registered trademarks are the property of their respective companies. Contact Information Web: http://www.iwatt.com E-mail: info@iwatt.com Phone: 408-374-4200 Fax: 408-341-0455 iWatt Inc. 101 Albright Way Los Gatos CA 95032-1827 Disclaimer iWatt reserves the right to make changes to its products and to discontinue products without notice. The applications information, schematic diagrams, and other reference information included herein is provided as a design aid only and are therefore provided as-is. iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights. Certain applications using semiconductor products may involve potential risks of death, personal injury, or severe property or environmental damage (“Critical Applications”). iWatt SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED, INTENDED, AUTHORIZED, OR WARRANTED TO BE SUITABLE FOR USE IN LIFE‑SUPPORT APPLICATIONS, DEVICES OR SYSTEMS, OR OTHER CRITICAL APPLICATIONS. Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer. Questions concerning potential risk applications should be directed to iWatt, Inc. iWatt semiconductors are typically used in power supplies in which high voltages are present during operation. High-voltage safety precautions should be observed in design and operation to minimize the chance of injury. Rev. 1.9 iW1710 5/6/10 iWatt Confidential-Protected By Traceable Watermark Page 20

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